**Advantages of microstrip antennas:**


These features are very important for wearable communication systems. **Disadvantages of microstrip antennas**:

• Limited bandwidth (usually 1–5%). However, wider bandwidth is possible with increased antenna structure complexity.

By adjusting the location of the antenna feed point, we can achieve any impedance and match the antenna to the RF system, usually 50 Ω. The antenna shape may be rectangular, square, triangle, circle, or any arbitrary shape as presented in **Figure 15**. The antenna dimension, W, is given by Eq. (1) and is a function of the effective

*<sup>W</sup>* <sup>¼</sup> *<sup>c</sup>*

*BW* <sup>¼</sup> *<sup>H</sup>*

In the transmission line model (TLM), the patch antenna functions as two narrow slots connected by a microstrip line, as illustrated in **Figure 17**. TLM model provides a good physical understanding of the electrical characteristics of patch antennas. The electric field along and underneath the patch is given in Eq. (3) and is a function of z. In the design of a wearable patch antenna, the body electrical

The gain of patch antenna is the function of the antenna effective area and can be between 0 and 5 dBi. We may increase printed antenna gain by using antenna array configuration. In low-cost microstrip antenna arrays, the RF feed network may be integrated to the radiating elements on the same substrate. Microstrip arrays feed networks are shown in **Figure 16**. A parallel feed network is illustrated in **Figure 16(a)**. A parallel series feed network is illustrated in **Figure 16(b)**.

2*f* ffiffiffiffiffiffi

ffiffiffiffiffiffi

*<sup>ϵ</sup>eff* <sup>p</sup> (1)

*<sup>ϵ</sup>eff* <sup>p</sup> (2)

dielectric constant and resonant frequency:

**Figure 14.**

**Figure 15.**

**29**

*Fields of rectangular microstrip antenna.*

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications*

*DOI: http://dx.doi.org/10.5772/intechopen.93492*

*Microstrip antenna popular configurations.*

The antenna bandwidth is given in Eq. (2):

**4.2 Transmission line model of patch antennas**

parameter should be considered to achieve an accurate design.


The patch magnetic field is perpendicular to the E-field. There is no conductor to carry the RF current so at the edge of the strip (X/L = 0 and X/L = 1), the H-field drops to zero and is maximum in the center. The E-field is zero at the center and at maximum value (and opposite polarity) at the edges (X/L = 0 and X/L = 1), see **Figure 14**. The ratio of E- to H-field is proportional to the patch impedance. Microstrip antennas may be fed by a coaxial probe feed or by a microstrip feed line.

**Figure 13.** *Microstrip antenna electric fields, a cross-sectional view.*

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications DOI: http://dx.doi.org/10.5772/intechopen.93492*

**Figure 15.** *Microstrip antenna popular configurations.*

antennas. However, loop, PIFA, slot, and dipole-printed are widely used in RF systems. Printed antennas may be employed in communication mobile phones, IoT,

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

Microstrip antennas are etched on a low loss dielectric substrate. A crosssectional view of the microstrip antenna electric fields is presented in **Figure 13**. Microstrip antennas are thin conducting patches etched on a substrate with dielectric constant εr and thickness H. Usually, H is less than 0.1 λ. Microstrip antennas are presented in [1–7]. The wearable antenna may be attached to the human body or

**4.1 Wearable microstrip antennas for 5G and medical systems**

seekers, and in medical systems.

**Advantages of microstrip antennas:**

• Flexible, Conformal structures are possible.

**Disadvantages of microstrip antennas**:

with increased antenna structure complexity.

• Limited gain up to 30 dBi, 16 16 arrays.

*Microstrip antenna electric fields, a cross-sectional view.*

• Low cost relative to conventional wired antennas.

• Easy to fabricate a large uniform arrays and phased arrays.

These features are very important for wearable communication systems.

• Limited bandwidth (usually 1–5%). However, wider bandwidth is possible

The patch magnetic field is perpendicular to the E-field. There is no conductor to carry the RF current so at the edge of the strip (X/L = 0 and X/L = 1), the H-field drops to zero and is maximum in the center. The E-field is zero at the center and at maximum value (and opposite polarity) at the edges (X/L = 0 and X/L = 1), see **Figure 14**. The ratio of E- to H-field is proportional to the patch impedance.

Microstrip antennas may be fed by a coaxial probe feed or by a microstrip feed line.

• Low power handling less than 50 W depends on substrate thickness.

• High feed network losses at high frequencies, above 12 GHz.

• Light weight and low volume.

inserted inside a belt.

**Figure 13.**

**28**

By adjusting the location of the antenna feed point, we can achieve any impedance and match the antenna to the RF system, usually 50 Ω. The antenna shape may be rectangular, square, triangle, circle, or any arbitrary shape as presented in **Figure 15**.

The antenna dimension, W, is given by Eq. (1) and is a function of the effective dielectric constant and resonant frequency:

$$W = \frac{c}{\mathfrak{F}\sqrt{c\_{\text{eff}}}} \tag{1}$$

The antenna bandwidth is given in Eq. (2):

$$BW = \frac{H}{\sqrt{c\_{\mathcal{H}}}} \tag{2}$$

The gain of patch antenna is the function of the antenna effective area and can be between 0 and 5 dBi. We may increase printed antenna gain by using antenna array configuration. In low-cost microstrip antenna arrays, the RF feed network may be integrated to the radiating elements on the same substrate. Microstrip arrays feed networks are shown in **Figure 16**. A parallel feed network is illustrated in **Figure 16(a)**. A parallel series feed network is illustrated in **Figure 16(b)**.

#### **4.2 Transmission line model of patch antennas**

In the transmission line model (TLM), the patch antenna functions as two narrow slots connected by a microstrip line, as illustrated in **Figure 17**. TLM model provides a good physical understanding of the electrical characteristics of patch antennas. The electric field along and underneath the patch is given in Eq. (3) and is a function of z. In the design of a wearable patch antenna, the body electrical parameter should be considered to achieve an accurate design.

admittance is a function of the feed point position along the z-axis as given in Eq. (6). At the feed point, the slot admittance is transformed by the equivalent length of the transmission line. The width, *W,* of the microstrip antenna controls the input impedance. For a square patch antenna fed by a microstrip line, the input impedance is around 300 Ohms. By increasing the width, the impedance can be

> 1 þ *j ZL Z*0

> > *ZL Z*0

To prevent excitation of higher-order modes, the thickness of the substrate should be less than a tenth of the wavelength. We can calculate the cutoff frequency

*fc* <sup>¼</sup> *<sup>c</sup>*

4*H* ffiffiffiffiffiffiffiffiffiffi

As shown in **Figures 13** and **14**, the edges of microstrip line and antenna part of the fields propagate in air and the other part of the fields propagates in the dielectric substrate. The effective dielectric constant is usually higher than *<sup>ε</sup>r*þ<sup>1</sup>

and is less than the substrate's dielectric constant. The effective dielectric constant of the microstrip line may be calculated by using Eqs. (8) and (9) as

> *H W* � � � � �0*:*<sup>5</sup>

This calculation ignores the strip thickness and frequency dispersion. If the substrate thickness is less than tenth of a wavelength their effects are negligible.

A major part of losses in microstrip line are due to conductor loss. Radiation loss and dielectric losses are lower. Losses in microstrip lines and antennas are the major disadvantage of microstrip antennas and limit the gain and efficiency of microstrip antennas at high frequencies. Losses in microstrip lines and antennas increase

1 þ 12

<sup>2</sup> <sup>þ</sup> *<sup>ε</sup><sup>r</sup>* � <sup>1</sup> 2

significantly at high frequencies as presented in Eqs. (10) and (11).

tan *βl*<sup>1</sup>

¼ *Y*<sup>1</sup>

*<sup>ε</sup>* � <sup>1</sup> <sup>p</sup> (7)

<sup>þ</sup> <sup>0</sup>*:*04 1 � *<sup>W</sup>*

� � � � <sup>2</sup> " # (8)

*H W*

1 þ 12

*H*

� � � � �0*:*<sup>5</sup> " # (9)

(6)

2

þ *j*tan *βl*<sup>1</sup>

reduced. Larger widths can increase the patch bandwidth.

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications*

*DOI: http://dx.doi.org/10.5772/intechopen.93492*

*Y l*ð Þ¼ <sup>1</sup> *Z*<sup>0</sup>

*Yin* ¼ *Y*<sup>1</sup> þ *Y*<sup>2</sup>

**4.3 Excitation of higher order modes in microstrip antennas**

of the higher-order mode by using Eq. (7):

**4.4 Microstrip effective dielectric constant**

function of W/H: For *<sup>W</sup> H* � �<1,

> For *<sup>W</sup> H* � �≥1,

**31**

*<sup>ε</sup><sup>e</sup>* <sup>¼</sup> *<sup>ε</sup><sup>r</sup>* <sup>þ</sup> <sup>1</sup>

<sup>2</sup> <sup>þ</sup> *<sup>ε</sup><sup>r</sup>* � <sup>1</sup> 2

**4.5 Losses in microstrip antennas**

*<sup>ε</sup><sup>e</sup>* <sup>¼</sup> *<sup>ε</sup><sup>r</sup>* <sup>þ</sup> <sup>1</sup>

**Figure 16.**

*Configuration of integrated microstrip array and feed network. (a) Parallel feed network. (b) Parallel series feed network.*

**Figure 17.** *Patch transmission line model, two slots connected by a microstrip line.*

$$E\_{\rm x} \sim \cos\left(\frac{\pi z}{L\_{\rm eff}}\right) \tag{3}$$

At the patch edges z = 0 and z=*Leff* , the electric field is maximum. At the patch center z= *Leff* <sup>2</sup> , the electric field is equal to zero. For *<sup>H</sup> <sup>λ</sup>*<sup>0</sup> <0*:*1, the electric field distribution along the x-axis is uniform. The slot admittance is given in Eqs. (4) and (5):

$$G = \frac{W}{120\lambda\_0} \left[ 1 - \frac{1}{24} \left( \frac{2\pi H}{\lambda\_0} \right)^2 \right] for \, \frac{H}{\lambda\_0} < 0.1 \tag{4}$$

$$B = \frac{W}{120\lambda\_0} \left[ 1 - 0.636 \ln \left( \frac{2\pi H}{\lambda\_0} \right)^2 \right] \text{for } \frac{H}{\lambda\_0} < 0.1 \tag{5}$$

Here, R represents the radiation losses; G = 1/R; and B represents the capacitive nature of the slot. At resonance, for any position of the feed point along the patch, the susceptances of both slots cancel out at the feed point. However, the patch

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications DOI: http://dx.doi.org/10.5772/intechopen.93492*

admittance is a function of the feed point position along the z-axis as given in Eq. (6). At the feed point, the slot admittance is transformed by the equivalent length of the transmission line. The width, *W,* of the microstrip antenna controls the input impedance. For a square patch antenna fed by a microstrip line, the input impedance is around 300 Ohms. By increasing the width, the impedance can be reduced. Larger widths can increase the patch bandwidth.

$$Y(l\_1) = Z\_0 \frac{1 + j\frac{Z\_L}{Z\_0}\tan\beta l\_1}{\frac{Z\_L}{Z\_0} + j\tan\beta l\_1} = Y\_1$$
 
$$Y\_{in} = Y\_1 + Y\_2$$

#### **4.3 Excitation of higher order modes in microstrip antennas**

To prevent excitation of higher-order modes, the thickness of the substrate should be less than a tenth of the wavelength. We can calculate the cutoff frequency of the higher-order mode by using Eq. (7):

$$f\_{\varepsilon} = \frac{\varepsilon}{4H\sqrt{\varepsilon - 1}}\tag{7}$$

#### **4.4 Microstrip effective dielectric constant**

As shown in **Figures 13** and **14**, the edges of microstrip line and antenna part of the fields propagate in air and the other part of the fields propagates in the dielectric substrate. The effective dielectric constant is usually higher than *<sup>ε</sup>r*þ<sup>1</sup> 2 and is less than the substrate's dielectric constant. The effective dielectric constant of the microstrip line may be calculated by using Eqs. (8) and (9) as function of W/H:

For *<sup>W</sup> H* � �<1,

$$\varepsilon\_{\varepsilon} = \frac{\varepsilon\_{r} + 1}{2} + \frac{\varepsilon\_{r} - 1}{2} \left[ \left( 1 + 12 \left( \frac{H}{W} \right) \right)^{-0.5} + 0.04 \left( 1 - \left( \frac{W}{H} \right) \right)^{2} \right] \tag{8}$$

For *<sup>W</sup> H* � �≥1,

*Ex* � cos *<sup>π</sup><sup>z</sup>*

*Configuration of integrated microstrip array and feed network. (a) Parallel feed network. (b) Parallel series*

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

<sup>2</sup> , the electric field is equal to zero. For *<sup>H</sup>*

*<sup>G</sup>* <sup>¼</sup> *<sup>W</sup>* 120*λ*<sup>0</sup>

*Patch transmission line model, two slots connected by a microstrip line.*

*<sup>B</sup>* <sup>¼</sup> *<sup>W</sup>* 120*λ*<sup>0</sup>

center z=

**30**

**Figure 17.**

**Figure 16.**

*feed network.*

*Leff*

At the patch edges z = 0 and z=*Leff* , the electric field is maximum. At the patch

bution along the x-axis is uniform. The slot admittance is given in Eqs. (4) and (5):

� �<sup>2</sup> " #

<sup>1</sup> � <sup>0</sup>*:*<sup>636</sup> *ln* <sup>2</sup>*π<sup>H</sup>*

� �<sup>2</sup> " #

Here, R represents the radiation losses; G = 1/R; and B represents the capacitive nature of the slot. At resonance, for any position of the feed point along the patch, the susceptances of both slots cancel out at the feed point. However, the patch

2*πH λ*0

*λ*0

*for <sup>H</sup> λ*0

> *for <sup>H</sup> λ*0

<sup>1</sup> � <sup>1</sup> 24

*Leff* !

(3)

*<sup>λ</sup>*<sup>0</sup> <0*:*1, the electric field distri-

<0*:*1 (4)

< 0*:*1 (5)

$$\varepsilon\_{\varepsilon} = \frac{\varepsilon\_{r} + \mathbf{1}}{2} + \frac{\varepsilon\_{r} - \mathbf{1}}{2} \left[ \left( \mathbf{1} + \mathbf{12} \left( \frac{H}{W} \right) \right)^{-0.5} \right] \tag{9}$$

This calculation ignores the strip thickness and frequency dispersion. If the substrate thickness is less than tenth of a wavelength their effects are negligible.

#### **4.5 Losses in microstrip antennas**

A major part of losses in microstrip line are due to conductor loss. Radiation loss and dielectric losses are lower. Losses in microstrip lines and antennas are the major disadvantage of microstrip antennas and limit the gain and efficiency of microstrip antennas at high frequencies. Losses in microstrip lines and antennas increase significantly at high frequencies as presented in Eqs. (10) and (11).

#### *4.5.1 Conductor loss*

Conductor loss may be calculated by using Eq. (10):

$$\begin{aligned} a\_c &= 8.686 \log \left( R\_S / (2W \mathbb{Z}\_0) \right) \text{ dB/Length} \\\\ R\_S &= \sqrt{\pi f \mu \rho} \quad \text{Slein} \quad \text{Re} \, \text{si} \, \tan \alpha \end{aligned} \tag{10}$$

Conductor losses can be calculated by defining an equivalent loss tangent δc, given by δc ¼ <sup>δ</sup><sup>s</sup> *=<sup>h</sup>*, and δs = ffiffiffiffiffiffiffiffiffi 2*=*ωμσ p . The strip conductivity is σ, μ is the free space permeability, and h is the substrate height.

### *4.5.2 Dielectric loss*

The dielectric loss is given in Eq. (11):

$$\begin{aligned} \alpha\_d &= 27.3 \frac{\varepsilon\_r}{\sqrt{\varepsilon\_{eff}}} \frac{\varepsilon\_{eff} - 1}{\varepsilon\_r - 1} \frac{\text{tg}\,\delta}{\lambda\_0} \quad dB/cm\\ \text{tg}\,\delta &= \text{dielectric} \quad \text{loss} \quad coefficient \end{aligned} \tag{11}$$

**4.6 Patch radiation pattern**

*Cartesian coordinate system.*

*E<sup>θ</sup>* ¼

**Figure 18.**

*k*<sup>0</sup> ¼ <sup>2</sup>*<sup>π</sup>=λ*

> sin *kW* 2

> > *k*0*W*

*E<sup>φ</sup>* ¼

*k*<sup>0</sup> ¼ <sup>2</sup>*<sup>π</sup>=λ*

**applications**

**33**

simulated by using Eqs. (12) and (13):

sin *θ* sin *φ* � �

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications*

*DOI: http://dx.doi.org/10.5772/intechopen.93492*

<sup>2</sup> sin *<sup>θ</sup>* sin *<sup>φ</sup>*

sin *θ* sin *φ* � �

The magnitude of the fields is given by Eq. (14):

<sup>2</sup> sin *<sup>θ</sup>* sin *<sup>φ</sup>*

sin *kW* 2

*k*0*W*

The patch radiation pattern is function of the patch width, W. The coordinate

*k*0*L* 2

sin *θ* cos *φ*

sin *θ* cos *φ*

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi *E*2 *<sup>θ</sup>* <sup>þ</sup> *<sup>E</sup>*<sup>2</sup> *φ*

q

� � cos *<sup>φ</sup>* ð Þ <sup>411</sup>

� � cos *<sup>θ</sup>* sin *<sup>φ</sup>* ð Þ <sup>4</sup>*:*<sup>12</sup>

(12)

(13)

(14)

system is presented in **Figure 18**. The normalized radiation pattern may be

cos

*k*0*L* 2

*f*ð Þ¼ *θ*, *φ*

antennas' electrical parameters were calculated and optimized by.

**5. Wearable stacked microstrip antennas for 5G, IoT, and medical**

Stacked patch antennas were presented first in [1–7]. Single-layer microstrip antennas have a narrow bandwidth. This disadvantage limits the applications of microstrip antennas. By designing a double-layer microstrip antenna, we may get a wider bandwidth. Two-layer patch antennas may be the best antenna choice for wideband communication systems. On the first layer, the antenna matching network and a resonator are printed. On the second layer, the radiating element is printed. The electromagnetic field is coupled from the resonator to the radiating element. The resonator and the radiating element shapes may be rectangular, square, triangle, circle, or any other shape. The distance between the layers is optimized to get maximum bandwidth with the best antenna efficiency. The spacing between the layers may be foam or a substrate with low dielectric losses. All the

cos

Losses in microstrip lines are presented in **Tables 1** and **2**. For example, total loss of a microstrip line presented in **Table 1** at 40 GHz is 0.5 dB/cm. For example, total loss of a microstrip line presented in **Table 2** at 40 GHz is 1.4 dB/cm. We may conclude that losses in microstrip lines limit the applications of microstrip technology at high frequencies.


#### **Table 1.**

*Microstrip line losses for a substrate of 0.127 mm thickness with εr =9.9\* .*


#### **Table 2.**

*Microstrip line losses for a GaAs substrate of 0.05 mm thickness with εr =12.88\* .* *Wideband Wearable Antennas for 5G, IoT, and Medical Applications DOI: http://dx.doi.org/10.5772/intechopen.93492*

**Figure 18.** *Cartesian coordinate system.*

*4.5.1 Conductor loss*

given by δc ¼ <sup>δ</sup><sup>s</sup>

*4.5.2 Dielectric loss*

*=*

technology at high frequencies.

*\**

*\**

**32**

**Table 2.**

**Table 1.**

Conductor loss may be calculated by using Eq. (10):

ffiffiffiffiffiffiffiffiffiffi *πf μρ* q

> 2*=*ωμσ

*<sup>α</sup><sup>d</sup>* <sup>¼</sup> <sup>27</sup>*:*<sup>3</sup> *<sup>ε</sup><sup>r</sup>*

*W = 0.12 mm,Tanδ = 0.0002, 3 um gold, and conductivity = 3.5E7 mhos/meter.*

*W = 0.034 mm,Tanδ = 0.0004, 3 um gold, and conductivity = 3.5E7 mhos/meter.*

*Microstrip line losses for a GaAs substrate of 0.05 mm thickness with εr =12.88\**

*Microstrip line losses for a substrate of 0.127 mm thickness with εr =9.9\**

ffiffiffiffiffiffi *εeff* <sup>p</sup> *<sup>ε</sup>eff* � <sup>1</sup> *ε<sup>r</sup>* � 1

*tgδ* ¼ *dielectric loss coefficent*

Losses in microstrip lines are presented in **Tables 1** and **2**. For example, total loss of a microstrip line presented in **Table 1** at 40 GHz is 0.5 dB/cm. For example, total loss of a microstrip line presented in **Table 2** at 40 GHz is 1.4 dB/cm. We may conclude that losses in microstrip lines limit the applications of microstrip

**Frequency (GHz) Loss tangent loss (dB/cm) Metal loss (dB/cm) Total loss (dB/cm)**

**Frequency (GHz) Tangent loss (dB/cm) Metal loss (dB/cm) Total loss (dB/cm)**

 �0.010 �0.66 �0.67 �0.02 �0.96 �0.98 �0.03 �1.19 �1.22 �0.04 �1.38 �1.42

 �0.004 �0.23 �0.23 �0.009 �0.333 �0.34 �0.013 �0.415 �0.43 �0.018 �0.483 �0.5

*RS* ¼

*<sup>h</sup>*, and δs = ffiffiffiffiffiffiffiffiffi

permeability, and h is the substrate height.

The dielectric loss is given in Eq. (11):

*α<sup>c</sup>* ¼ 8*:*686 log ð Þ *RS=*ð Þ 2*WZ*<sup>0</sup> *dB=Length*

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

Conductor losses can be calculated by defining an equivalent loss tangent δc,

*Skin* Re *sis*tan *ce*

*tgδ λ*0

p . The strip conductivity is σ, μ is the free space

*dB=cm*

*.*

*.*

(10)

(11)

#### **4.6 Patch radiation pattern**

The patch radiation pattern is function of the patch width, W. The coordinate system is presented in **Figure 18**. The normalized radiation pattern may be simulated by using Eqs. (12) and (13):

$$E\_{\theta} = \frac{\sin\left(\frac{k\_0 W}{2}\sin\theta\sin\varphi\right)}{\frac{k\_0 W}{2}\sin\theta\sin\varphi}\cos\left(\frac{k\_0 L}{2}\sin\theta\cos\varphi\right)\cos\varphi \quad (411) \tag{12}$$

$$\begin{aligned} k\_0 &= 2\pi/\_{\dot{\lambda}} \\ E\_{\psi} &= \frac{\sin\left(\frac{k\_0 W}{2}\sin\theta\sin\varphi\right)}{\frac{k\_0 W}{2}\sin\theta\sin\varphi} \cos\left(\frac{k\_0 L}{2}\sin\theta\cos\varphi\right)\cos\theta\sin\varphi \quad (4.12) \\ k\_0 &= 2\pi/\_{\dot{\lambda}} \end{aligned} \tag{13}$$

The magnitude of the fields is given by Eq. (14):

$$f(\theta,\rho) = \sqrt{E\_{\theta}^{2} + E\_{\rho}^{2}}\tag{14}$$

## **5. Wearable stacked microstrip antennas for 5G, IoT, and medical applications**

Stacked patch antennas were presented first in [1–7]. Single-layer microstrip antennas have a narrow bandwidth. This disadvantage limits the applications of microstrip antennas. By designing a double-layer microstrip antenna, we may get a wider bandwidth. Two-layer patch antennas may be the best antenna choice for wideband communication systems. On the first layer, the antenna matching network and a resonator are printed. On the second layer, the radiating element is printed. The electromagnetic field is coupled from the resonator to the radiating element. The resonator and the radiating element shapes may be rectangular, square, triangle, circle, or any other shape. The distance between the layers is optimized to get maximum bandwidth with the best antenna efficiency. The spacing between the layers may be foam or a substrate with low dielectric losses. All the antennas' electrical parameters were calculated and optimized by.

using electromagnetic software. A 2.2 GHz square patch with circular polarization stacked antenna was designed. The resonator and the feed network were printed on a substrate with a relative dielectric constant of 2.4 and with a thickness of 0.16 cm. The dimensions of the square resonator are W = L = 4.5 cm. The radiating element was printed on a substrate with a relative dielectric constant of 2.2 and with a thickness of 0.16 cm. The radiator is a square patch with dimensions W = L = 4.8 cm. The antenna is circular polarized. A 3 dB, 90° branch coupler is connected to the antenna feed lines, as shown in **Figure 19**. The antenna bandwidth is 13% for VSWR better than 3:1. The measured antenna beamwidth is 73°. The measured antenna gain is 7.5 dBi at 2.2 GHz. This antenna may be used in wideband communication systems. Comparison of calculated and measured results of stacked patch antennas is listed in **Table 3**. The antennas listed in **Table 3** may be used in wearable communication systems. Results in **Table 3** indicate that the bandwidth of stacked patch antennas may be around 9–15% for VSWR better than 2:1. There is a good agreement between the measured and calculated results. In **Figure 20**, a stacked microstrip antenna is shown. The antenna feed and matching network is printed on FR4 with a dielectric constant of 4.2 and 1.6 mm thickness. The radiator is printed on a dielectric substrate with a dielectric constant of 2.2 and 1.6 mm thickness. The dimensions of the microstrip stacked patch antenna shown in **Figure 20** are 3.3 2 0.32 cm. The computed S11 parameters are presented in **Figure 21**. Radiation pattern of the microstrip stacked patch is shown in **Figure 22**.

**Figure 19.** *Feed network of a circular polarized stacked patch antenna.*


The antenna bandwidth is around 7% for VSWR better than 3:1. The antenna bandwidth is improved to 10% for VSWR better than 2.0:1 by adding 8 mm air

Two Ka-band, stacked patch microstrip antenna arrays, which consist of 256 radiating elements, have been designed on a substrate with εr = 2.2, 0.25 mm thick. The first Type A array with a parallel feed network, is shown in **Figure 16(a)**. The second Type B array is shown in **Figure 16(b)** has more bend discontinuities in the

spacing between the layers. The antenna beamwidth is around 72°.

*A microstrip stacked patch antenna for 5G, IoT, and medical applications.*

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications*

*DOI: http://dx.doi.org/10.5772/intechopen.93492*

The antenna gain is around 7 dBi.

*Computed S11 of the microstrip stacked patch.*

**Figure 20.**

**Figure 21.**

**35**

**5.1 Stacked microstrip 35 GHz antennas arrays**

#### **Table 3.**

*Comparison of calculated and measured results of stacked microstrip antennas.*

**Figure 20.** *A microstrip stacked patch antenna for 5G, IoT, and medical applications.*

**Figure 21.** *Computed S11 of the microstrip stacked patch.*

The antenna bandwidth is around 7% for VSWR better than 3:1. The antenna bandwidth is improved to 10% for VSWR better than 2.0:1 by adding 8 mm air spacing between the layers. The antenna beamwidth is around 72°.

The antenna gain is around 7 dBi.

#### **5.1 Stacked microstrip 35 GHz antennas arrays**

Two Ka-band, stacked patch microstrip antenna arrays, which consist of 256 radiating elements, have been designed on a substrate with εr = 2.2, 0.25 mm thick. The first Type A array with a parallel feed network, is shown in **Figure 16(a)**. The second Type B array is shown in **Figure 16(b)** has more bend discontinuities in the

using electromagnetic software. A 2.2 GHz square patch with circular polariza-

tion stacked antenna was designed. The resonator and the feed network were printed on a substrate with a relative dielectric constant of 2.4 and with a thickness of 0.16 cm. The dimensions of the square resonator are W = L = 4.5 cm. The radiating element was printed on a substrate with a relative dielectric constant of 2.2 and with a thickness of 0.16 cm. The radiator is a square patch with dimensions W = L = 4.8 cm. The antenna is circular polarized. A 3 dB, 90° branch coupler is connected to the antenna feed lines, as shown in **Figure 19**. The antenna bandwidth is 13% for VSWR better than 3:1. The measured antenna beamwidth is 73°. The measured antenna gain is 7.5 dBi at 2.2 GHz. This antenna may be used in wideband communication systems. Comparison of calculated and measured results of stacked patch antennas is listed in **Table 3**. The antennas listed in **Table 3** may be used in wearable communication systems. Results in **Table 3** indicate that the bandwidth of stacked patch antennas may be around 9–15% for VSWR better than 2:1. There is a good agreement between the measured and calculated results. In **Figure 20**, a stacked microstrip antenna is shown. The antenna feed and matching network is printed on FR4 with a dielectric constant of 4.2 and 1.6 mm thickness. The radiator is printed on a dielectric substrate with a dielectric constant of 2.2 and 1.6 mm thickness. The dimensions of the microstrip stacked patch antenna shown in **Figure 20** are 3.3 2 0.32 cm. The computed S11 parameters are presented in **Figure 21**. Radiation pattern of the microstrip stacked patch is shown in **Figure 22**.

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

**Figure 19.**

**Table 3.**

**34**

*Feed network of a circular polarized stacked patch antenna.*

**Antenna F (GHz) Bandwidth (%) Beamwidth Gain (dBi) Polarization**

Square 2.2 11 10 74 72 7.5 7.5 Circular Circular 2.2 14 15 74 72 7.5 7.9 Linear Annular disc 2.2 12 11.5 80 78 6.5 6.6 Linear Rectangular 2.0 10 9 72 72 7.5 7.4 Linear Circular 2.4 10 9 74 72 7.5 7 Linear Circular 2.4 10 10 74 72 7.5 7.5 Circular

*Comparison of calculated and measured results of stacked microstrip antennas.*

**Calc. Meas. Calc. Meas. Calc. Meas.**

modified array Type C was increased by 1.6 dB. The arrays' measured bandwidth is

A mono-pulse double-layer circular patch antenna was designed at Ku band, 15 GHz. The mono-pulse antenna consists of four circular patch antennas and a feed network as presented in **Figure 23**. The circular resonator and the branch coupler were printed on a substrate with a relative dielectric constant of 2.45 and with a thickness of 0.8 mm. The diameter of the circular microstrip resonator is 0.42 cm. The circular radiator was printed on a substrate with a relative dielectric constant of 2.25 and with a thickness of 0.8 mm. The diameter of the circular patch is 0.45 cm. The comparator consists of three 3 dB, 180° rat-race couplers that are connected to four circular patches via the antenna feed lines, as presented in **Figures 23** and **24**. The strip-line 3 dB, 180° rat-race couplers are printed on a substrate with a relative dielectric constant of 2.2 and thickness of 0.8 mm. The comparator structure and

, difference port **Δ**, an azimuth difference port **Δ**Az, and an elevation difference port **Δ**El. The antenna bandwidth is 11% for VSWR better than 2:1. The antenna beam width is around 36°. The computed and measured antenna gain is around 10.5

**Figures 23** and **24**. The comparator output ports are: a sum port

around 12% for VSWR better than 2:1.

*DOI: http://dx.doi.org/10.5772/intechopen.93492*

ports are shown in

P

**Figure 23.**

**37**

*A microstrip stacked mono-pulse antenna.*

**6. Stacked mono-pulse Ku-band patch antenna**

*Wideband Wearable Antennas for 5G, IoT, and Medical Applications*

dBi. The maximum comparator losses are 0.75 dB.

**Figure 22.** *Radiation pattern of the microstrip stacked patch.*


#### **Table 4.**

*Comparison of electrical performance of 256 stacked patch microstrip antenna arrays.*

feeding network than Type A array. In the Type C array, a 10-cm coaxial line was used to replace the same length of microstrip line in the Type A array. Comparison of measured results of the arrays, given in **Table 4**, shows that the gain of the

modified array Type C was increased by 1.6 dB. The arrays' measured bandwidth is around 12% for VSWR better than 2:1.
