**1. Introduction**

The leaky-wave antennas have attracted much attention since proposed and found wide applications in wireless communication systems. As one kind of the leaky-wave antenna, Fabry-Pérot cavity (FPC) antennas are preferred by scientists due to their broadside pattern and high-gain performance. However, the FPC antenna usually suffer some disadvantages due to its inherent limitations of the resonate structure, such as narrow band, fixing beam, and high profile. So many researches have been done on wideband and beam tilted FPC antenna. To overcome these disadvantages of the conventional FPC antenna, we proposed novel strategies that realize reconfigurable FPC antennas using the PIN diodes. In this chapter, we designed three reconfigurable FPC antennas by metasurfaces (MSs).

Through adding PIN diodes on the MS, the reflection phase of the MS can be controlled by tuning the states of the diodes. The different reflect phase distributions of the MS can make the FPC antenna present different frequency and radiation performance. The reconfigurable FPC antenna can improve the performance of antenna, so that the antenna can be widely used.

then the first transmission coefficient is

*DOI: http://dx.doi.org/10.5772/intechopen.91695*

after *n* times of reflection is

*ttotal* <sup>¼</sup> <sup>X</sup><sup>∞</sup>

power transmission coefficient is

where

tion of the antenna is

**225**

*n*¼0

*tn* ¼ *t*

After once reflected by the PRS, the transmission coefficient is

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces*

*tn* <sup>¼</sup> *trne*

So, the total transmission coefficient along *θ* direction is

X∞ *n*¼0 *r ne*

According to the energy conservation law,*T*<sup>2</sup> = 1 � *<sup>R</sup>*<sup>2</sup>

*<sup>D</sup>* <sup>¼</sup> *ttotalt* <sup>∗</sup>

maximum value of the power transmission coefficient is

*<sup>h</sup>* <sup>¼</sup> *<sup>λ</sup>*

**3.1 Design of the reconfigurable PRS**

**3. Frequency reconfigurable Fabry-Pérot antenna**

<sup>Φ</sup> <sup>¼</sup> <sup>4</sup>*<sup>π</sup>*

*<sup>D</sup>* <sup>¼</sup> <sup>1</sup> � *<sup>R</sup>*<sup>2</sup>

<sup>4</sup>*<sup>π</sup> <sup>φ</sup>*<sup>1</sup> <sup>þ</sup> *<sup>φ</sup>*<sup>2</sup> ð Þþ *<sup>λ</sup>*

*t*<sup>0</sup> ¼ *t* (2)

�2*jnkh* cos *<sup>θ</sup>*þ*inφ*<sup>2</sup> (4)

1 � *re*�2*jkh* cos *<sup>θ</sup>*þ*jφ*<sup>2</sup>

<sup>1</sup> <sup>þ</sup> *<sup>R</sup>*<sup>2</sup> � <sup>2</sup>*<sup>R</sup>* cos <sup>Φ</sup> (6)

<sup>1</sup> � *<sup>R</sup>* (8)

*N*, *N* ¼ 0, 1, 2 … (9)

*<sup>λ</sup> <sup>h</sup>* cos *<sup>θ</sup>* � *<sup>φ</sup>*<sup>1</sup> � *<sup>φ</sup>*<sup>2</sup> (7)

. Therefore, the total

(5)

*<sup>t</sup>*<sup>1</sup> <sup>¼</sup> *tre*�2*jkh<sup>=</sup>* cos *<sup>θ</sup>*þ2*jkh* tan *<sup>θ</sup>* sin *<sup>θ</sup>*þ*jφ*<sup>2</sup> <sup>¼</sup> *tre*�2*jkh* cos *<sup>θ</sup>*þ*jφ*<sup>2</sup> (3)

�2*jnkh* cos *<sup>θ</sup>*þ*jnφ*<sup>2</sup> <sup>¼</sup> *<sup>t</sup>*

where *φ*<sup>2</sup> is the reflection phase of the metal ground plane, *λ* is the wavelength in the free space, and *K* = 2π/*λ* is the propagation constant. According to the above equation, it can be deduced that the transmission coefficient of the resonate cavity

*total* <sup>¼</sup> <sup>1</sup> � *<sup>R</sup>*<sup>2</sup>

*φ*<sup>1</sup> is the reflection phase of the PRS. According to Eq. (6), when cos *Ф* = 1, the

<sup>1</sup> <sup>þ</sup> *<sup>R</sup>*<sup>2</sup> � <sup>2</sup>*<sup>R</sup>* <sup>¼</sup> <sup>1</sup> <sup>þ</sup> *<sup>R</sup>*

It can be seen that the power transmission coefficient is mainly relative to the reflection magnitude of the PRS. The power transmission coefficient increases with the increase of the reflection magnitude of the PRS, so does the gain of the antenna. The maximum radiation direction of the antenna is *θ* = 0°. And the resonate condi-

2

In order to realize the reflection phase configuration of the PRS, the unit cell of the PRS must be configurable [15]. The structure of the unit cell is shown in **Figure 2a**. The unit cell is printed on a substrate of FR4 with a thickness of 1.6 mm and

### **2. Basic theory of the Fabry-Pérot cavity antenna**

Due to its well frequency selection characteristics, the Fabry-Pérot (FP) resonate cavity are widely used in many applications, such as spectral analyzer, interference filter, and so on. The FP resonate cavity can also be used in the design of high-gain antennas [1–4]. Adding a feeder into the cavity, the electromagnetic wave emanating from the feeder experiences multiple reflections and transmissions in the cavity [5, 6]. When the resonance condition is satisfied, the wave coming out of the cavity will be in phase, and then bidirectional high-gain radiation is achieved. However, most applications require unidirectional antenna radiation. So the FP resonate cavity should be changed slightly to design high-gain FPC antenna [7–9].

In practice, the artificial partially reflection surfaces (PRS) are usually used to design FPC antenna. In order to realize unidirectional radiation, one of the PRS of the FP resonate cavity should be replaced by the metal ground plane [10–12]. The PRS and the metal ground plane form the resonate cavity of the FPC antenna. The schematic model of the FPC antenna is shown in **Figure 1**. The distance between the PRS and the ground plane is *h*. The radiator locates in the middle of the cavity above the ground plane. The electromagnetic wave emanating from the radiator is incident on the PRS with an angle of *θ*. One part of the electromagnetic wave transmits through the PRS and the other part is reflected into the cavity by the PRS. The reflected wave is totally reflected by the ground plane and is incident to the PRS again with the same angle, forming the secondary transmission and reflection. The electromagnetic wave experiences multiple reflections and transmissions between the PRS and the ground plane and finally all get through the PRS [11, 13, 14].

Assuming that the transmission coefficient and reflection coefficient of the PRS are

$$r = \operatorname{Re}^{j\rho\_1}, \quad \mathbf{t} = \operatorname{Te}^{j\theta\_1} \tag{1}$$

**Figure 1.** *The schematic model of the FPC antenna.*

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces DOI: http://dx.doi.org/10.5772/intechopen.91695*

then the first transmission coefficient is

$$\mathbf{t}\_0 = \mathbf{t} \tag{2}$$

After once reflected by the PRS, the transmission coefficient is

$$t\_1 = t \text{re}^{-2jkh/\cos\theta + 2jkh\tan\theta\sin\theta + j\rho\_2} = t \text{re}^{-2jkh\cos\theta + j\rho\_2} \tag{3}$$

where *φ*<sup>2</sup> is the reflection phase of the metal ground plane, *λ* is the wavelength in the free space, and *K* = 2π/*λ* is the propagation constant. According to the above equation, it can be deduced that the transmission coefficient of the resonate cavity after *n* times of reflection is

$$t\_n = tr^n e^{-2jnkh\cos\theta + in\rho\_2} \tag{4}$$

So, the total transmission coefficient along *θ* direction is

$$t\_{\text{total}} = \sum\_{n=0}^{\infty} t\_n = t \sum\_{n=0}^{\infty} r^n e^{-2jnkh\cos\theta + jn\rho\_2} = \frac{t}{1 - re^{-2jkh\cos\theta + j\rho\_2}}\tag{5}$$

According to the energy conservation law,*T*<sup>2</sup> = 1 � *<sup>R</sup>*<sup>2</sup> . Therefore, the total power transmission coefficient is

$$D = t\_{total} t\_{total}^\* = \frac{1 - R^2}{1 + R^2 - 2R \cos \Phi} \tag{6}$$

where

Through adding PIN diodes on the MS, the reflection phase of the MS can be controlled by tuning the states of the diodes. The different reflect phase distributions of the MS can make the FPC antenna present different frequency and radiation performance. The reconfigurable FPC antenna can improve the performance of

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

Due to its well frequency selection characteristics, the Fabry-Pérot (FP) resonate cavity are widely used in many applications, such as spectral analyzer, interference filter, and so on. The FP resonate cavity can also be used in the design of high-gain antennas [1–4]. Adding a feeder into the cavity, the electromagnetic wave emanating from the feeder experiences multiple reflections and transmissions in the cavity [5, 6]. When the resonance condition is satisfied, the wave coming out of the cavity will be in phase, and then bidirectional high-gain radiation is achieved. However, most applications require unidirectional antenna radiation. So the FP resonate cav-

In practice, the artificial partially reflection surfaces (PRS) are usually used to design FPC antenna. In order to realize unidirectional radiation, one of the PRS of the FP resonate cavity should be replaced by the metal ground plane [10–12]. The PRS and the metal ground plane form the resonate cavity of the FPC antenna. The schematic model of the FPC antenna is shown in **Figure 1**. The distance between the PRS and the ground plane is *h*. The radiator locates in the middle of the cavity above the ground plane. The electromagnetic wave emanating from the radiator is incident on the PRS with an angle of *θ*. One part of the electromagnetic wave transmits through the PRS and the other part is reflected into the cavity by the PRS. The reflected wave is totally reflected by the ground plane and is incident to the PRS again with the same angle, forming the secondary transmission and reflection. The electromagnetic wave experiences multiple reflections and transmissions between the PRS and the ground plane and finally all get through the PRS [11, 13, 14]. Assuming that the transmission coefficient and reflection coefficient of the

*<sup>r</sup>* <sup>¼</sup> *Re <sup>j</sup>φ*<sup>1</sup> , *<sup>t</sup>* <sup>¼</sup> *Te<sup>j</sup>θ*<sup>1</sup> (1)

antenna, so that the antenna can be widely used.

PRS are

**Figure 1.**

**224**

*The schematic model of the FPC antenna.*

**2. Basic theory of the Fabry-Pérot cavity antenna**

ity should be changed slightly to design high-gain FPC antenna [7–9].

$$\Phi = \frac{4\pi}{\lambda} h \cos \theta - \varphi\_1 - \varphi\_2 \tag{7}$$

*φ*<sup>1</sup> is the reflection phase of the PRS. According to Eq. (6), when cos *Ф* = 1, the maximum value of the power transmission coefficient is

$$D = \frac{\mathbf{1} - R^2}{\mathbf{1} + R^2 - 2R} = \frac{\mathbf{1} + R}{\mathbf{1} - R} \tag{8}$$

It can be seen that the power transmission coefficient is mainly relative to the reflection magnitude of the PRS. The power transmission coefficient increases with the increase of the reflection magnitude of the PRS, so does the gain of the antenna. The maximum radiation direction of the antenna is *θ* = 0°. And the resonate condition of the antenna is

$$h = \frac{\lambda}{4\pi}(\rho\_1 + \rho\_2) + \frac{\lambda}{2}N, \quad N = 0, 1, 2\dots \tag{9}$$

#### **3. Frequency reconfigurable Fabry-Pérot antenna**

#### **3.1 Design of the reconfigurable PRS**

In order to realize the reflection phase configuration of the PRS, the unit cell of the PRS must be configurable [15]. The structure of the unit cell is shown in **Figure 2a**. The unit cell is printed on a substrate of FR4 with a thickness of 1.6 mm and

**Figure 2.**

*(a) The structure of the unit cell and (b) the view of reconfigurable PRS.*

reflection phase of the unit cell can be considered as the reflection phase of the PRS. The diodes are the BAR50-03 W from Infineon. Because there is no model of the diode in the CST, it is replaced by the equivalent circuit in the simulation. When the diode is closed, they can be replaced by resistors of 1 Ω. When the diode is open, it is replaced by a circuit that consists of a 10 kΩ resistor in parallel with a 0.15 pF capacitor.

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces*

In order to realize frequency reconfiguration, the FPC antenna must adopt dualband feeder. So we design a dual-band microwave antenna as the feeder of the FPC antenna. The structure of the feeder antenna is shown in **Figure 5a**. The antenna is printed on a substrate of FR4 with the permittivity of 4.4 and thickness of 1.6 mm. It consists of two different metal patches which are connected by a microwave line. Two patches have different lengths and widths, so they can resonate at different frequencies. The bottom of the substrate is a metal ground plane. A SMA connecter is used to feed the antenna from the bottom side, and the feeding point is on the microstrip line. The antenna can realize impedance matching through tuning the position of the feeding point and the microstrip line. The feed antenna is designed

*Reflection coefficients of the unit cell. Geometry of the antenna: (a) top view of the feed antenna and (b) side*

**3.2 Antenna design**

*Effect of* w *on reflection phase of the unit cell.*

*DOI: http://dx.doi.org/10.5772/intechopen.91695*

**Figure 4.**

to work at 4.6 and 5.5 GHz.

**Figure 5.**

**227**

*view of the FPC antenna.*

**Figure 3.** *Reflection coefficients of the unit cell.*

permittivity of 4.4. It consists of a square patch and a square ring. Two slots are added on the square ring, and two PIN diodes are inserted into the slots. The two PIN diodes are controlled simultaneously. When diodes are in different states, the unit cell presents different reflection phases. **Figure 3** shows the simulation reflection coefficients of the unit cell with *w* = 7 mm. We can see that, when the diode state changes from OFF to ON, the reflection phase of the unit cell is delayed. According to Eq. (9), the *λ* increases with the decrease of *φ*1. So the antenna will operate at low frequency when the diodes are ON and high frequency when the diodes are OFF. Besides, the reflection phase of the unit cell with the variation of *w* is plotted in **Figure 4**. It can be seen that the reflection phase of the unit cell decreases with the increase of w no matter if the diodes are open or closed. So the *w* can be used in tuning the reflection phase of the unit cell to meet the requirement of the antenna.

**Figure 2b** shows the structure of the configurable PRS. It consists of 10 10 unit cells. All diodes on the PRS are set along the same orientation, so all diodes can be controlled by only one DC source. The biasing point V on the top of the PRS connects the anode of the DC source, and the biasing point Gnd on the bottom of the PRS connects the cathode of the DC source. Inductors are added between the feeding point and the metal structure on the PRS to prevent the RF signal from entering the DC source. The periodic boundary is adopted in the simulation of the unit cell, so the

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces DOI: http://dx.doi.org/10.5772/intechopen.91695*

**Figure 4.** *Effect of* w *on reflection phase of the unit cell.*

reflection phase of the unit cell can be considered as the reflection phase of the PRS. The diodes are the BAR50-03 W from Infineon. Because there is no model of the diode in the CST, it is replaced by the equivalent circuit in the simulation. When the diode is closed, they can be replaced by resistors of 1 Ω. When the diode is open, it is replaced by a circuit that consists of a 10 kΩ resistor in parallel with a 0.15 pF capacitor.

#### **3.2 Antenna design**

In order to realize frequency reconfiguration, the FPC antenna must adopt dualband feeder. So we design a dual-band microwave antenna as the feeder of the FPC antenna. The structure of the feeder antenna is shown in **Figure 5a**. The antenna is printed on a substrate of FR4 with the permittivity of 4.4 and thickness of 1.6 mm. It consists of two different metal patches which are connected by a microwave line. Two patches have different lengths and widths, so they can resonate at different frequencies. The bottom of the substrate is a metal ground plane. A SMA connecter is used to feed the antenna from the bottom side, and the feeding point is on the microstrip line. The antenna can realize impedance matching through tuning the position of the feeding point and the microstrip line. The feed antenna is designed to work at 4.6 and 5.5 GHz.

#### **Figure 5.**

*Reflection coefficients of the unit cell. Geometry of the antenna: (a) top view of the feed antenna and (b) side view of the FPC antenna.*

permittivity of 4.4. It consists of a square patch and a square ring. Two slots are added on the square ring, and two PIN diodes are inserted into the slots. The two PIN diodes are controlled simultaneously. When diodes are in different states, the unit cell presents different reflection phases. **Figure 3** shows the simulation reflection coefficients of the unit cell with *w* = 7 mm. We can see that, when the diode state changes from OFF to ON, the reflection phase of the unit cell is delayed. According to Eq. (9), the *λ* increases with the decrease of *φ*1. So the antenna will operate at low frequency when the diodes are ON and high frequency when the diodes are OFF. Besides, the reflection phase of the unit cell with the variation of *w* is plotted in **Figure 4**. It can be seen that the reflection phase of the unit cell decreases with the increase of w no matter if the diodes are open or closed. So the *w* can be used in tuning the reflection phase of

**Figure 2b** shows the structure of the configurable PRS. It consists of 10 10 unit cells. All diodes on the PRS are set along the same orientation, so all diodes can be controlled by only one DC source. The biasing point V on the top of the PRS connects the anode of the DC source, and the biasing point Gnd on the bottom of the PRS connects the cathode of the DC source. Inductors are added between the feeding point and the metal structure on the PRS to prevent the RF signal from entering the DC source. The periodic boundary is adopted in the simulation of the unit cell, so the

the unit cell to meet the requirement of the antenna.

*(a) The structure of the unit cell and (b) the view of reconfigurable PRS.*

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

**Figure 2.**

**Figure 3.**

**226**

*Reflection coefficients of the unit cell.*

#### *Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

Next, we will introduce how to realize the dual-band working of the antenna through tuning the reflection phase of the PRS. **Figure 5b** shows the geometry of the FPC antenna. We consume that the height of the cavity is *h*<sup>L</sup> when the antenna work at low frequency and *h*<sup>H</sup> when the antenna work at high frequency. According Eq. (9), they can be expressed as.

$$h\_L = \frac{(\wp\_L + \wp\_2)\lambda\_L}{4\pi} + \frac{N\_L\lambda\_L}{2} \tag{10}$$

$$h\_H = \frac{(\wp\_H + \wp\_2)\lambda\_H}{4\pi} + \frac{N\_H\lambda\_H}{2} \tag{11}$$

**Figure 8.**

**Figure 9.**

**229**

*The radiation patterns of the antenna: (a) 4.6 GHz and (b) 5 GHz.*

*Simulated and measured S11 of the antenna.*

*DOI: http://dx.doi.org/10.5772/intechopen.91695*

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces*

When *h*<sup>L</sup> = *h*H, it can be considered as the antenna operates at two different frequencies without changing the antenna structure. In Eqs. (10) and (11), the *N* is set to 1 to obtain low profile of the antenna. The antenna is designed to work at 4.6 and 5.5 GHz, so *λ*<sup>L</sup> = 65.2 mm and *λ*<sup>H</sup> = 54.5 mm. The reflection phase of the metal ground plane *φ*<sup>2</sup> is always π; thus the *h* is only related to the reflection phase of the PRS. The *φ* is related to the width of the patch in the unit cell. So we can deduce the relationship between *h* and *w*, as shown in **Figure 6**. The black line is the relationship

**Figure 6.** *The relationship between* w *and* h*.*

**Figure 7.** *The photographs of the fabricated antenna.*

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces DOI: http://dx.doi.org/10.5772/intechopen.91695*

**Figure 8.** *Simulated and measured S11 of the antenna.*

Next, we will introduce how to realize the dual-band working of the antenna through tuning the reflection phase of the PRS. **Figure 5b** shows the geometry of the FPC antenna. We consume that the height of the cavity is *h*<sup>L</sup> when the antenna work at low frequency and *h*<sup>H</sup> when the antenna work at high frequency. According

4*π* þ

4*π* þ

When *h*<sup>L</sup> = *h*H, it can be considered as the antenna operates at two different frequencies without changing the antenna structure. In Eqs. (10) and (11), the *N* is set to 1 to obtain low profile of the antenna. The antenna is designed to work at 4.6 and 5.5 GHz, so *λ*<sup>L</sup> = 65.2 mm and *λ*<sup>H</sup> = 54.5 mm. The reflection phase of the metal ground plane *φ*<sup>2</sup> is always π; thus the *h* is only related to the reflection phase of the PRS. The *φ* is related to the width of the patch in the unit cell. So we can deduce the relationship between *h* and *w*, as shown in **Figure 6**. The black line is the relationship

*NLλ<sup>L</sup>*

*NHλ<sup>H</sup>*

<sup>2</sup> (10)

<sup>2</sup> (11)

*hL* <sup>¼</sup> *<sup>φ</sup><sup>L</sup>* <sup>þ</sup> *<sup>φ</sup>*<sup>2</sup> ð Þ*λ<sup>L</sup>*

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

*hH* <sup>¼</sup> *<sup>φ</sup><sup>H</sup>* <sup>þ</sup> *<sup>φ</sup>*<sup>2</sup> ð Þ*λ<sup>H</sup>*

Eq. (9), they can be expressed as.

**Figure 6.**

**Figure 7.**

**228**

*The relationship between* w *and* h*.*

*The photographs of the fabricated antenna.*

**Figure 9.** *The radiation patterns of the antenna: (a) 4.6 GHz and (b) 5 GHz.*

when *f* = 4.6 GHz and the diodes are ON, and the red line is the relationship when *f* = 5.5 GHz and the diodes are OFF. The two lines intersect at the point *w* = 8 mm and *h* = 27.2 mm. This indicates that when *h* = 27.2 mm and *w* = 8 mm, the antenna can work at 4.6 GHz if the diodes are ON and at 5.5 GHz if the diodes are OFF. So the antenna realizes frequency reconfiguration through tuning the states of the diodes.

inserted in the slot. The diodes are divided into two groups. The diodes on the ring are named K1 and the diode in the patch is named K2. The different states of two

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces*

The unit cell is simulated by the CST. The simulation setting of the unit cell is the same as the unit cell presented in Section 2. The reflection coefficients of the unit cell under *y*-polarized wave are shown in **Figure 11**. The simulation results show that the reflection phase of the unit cell at 5 GHz ranged from 165 to 262°. This suggests that the unit cell has a large range of reflection phase variation. Meanwhile, the reflection magnitudes of the unit cell in four states are always greater than 0.7.

A reconfigurable PRS is formed by the unit cell shown in **Figure 10**. The structure of the PRS is shown in **Figure 12**. It is composed by 6 6 unit cells and divided

**Diodes S1 S2 S3 S4** K1 OFF OFF ON ON K2 OFF ON OFF ON

group diodes give the four cell states, as shown in **Table 1**.

This can ensure a high gain of the FPC antenna.

*DOI: http://dx.doi.org/10.5772/intechopen.91695*

**Table 1.**

**Figure 11.**

**Figure 12.**

**231**

*The structure of the PRS: (a) top view and (b) bottom view.*

*The reflection phase of the unit cell.*

*The four states of the unit cell.*

#### **3.3 Fabrication and measurement of the antenna**

The photographs of the fabricated antenna are shown in **Figure 7**. The red lines connect the anode of the DC source, and the black lines connect the cathode. The S11 and radiation patterns of the antenna at two frequencies are measured. Good agreement between the simulated and measured results is obtained. The simulated and measured S11 of the antenna is shown in **Figure 8**. The 10 dB impedance band of the antenna is 4.55–4.7 GHz (3.3%) when diodes are ON and 5.37–5.63 GHz (4.7%) when diodes are OFF. The measured center of the low-frequency band is higher than the simulated result. This may be caused by the fabrication errors. During the welding of SMA joints, some unnecessary metals may be introduced. This results in the deviation of the resonant frequency of the antenna.

**Figure 9** presents the simulated and measured radiation patterns of the antenna. **Figure 9a** shows the radiation patterns at 4.6 GHz and **Figure 9b** shows at 5.5 GHz. The measured maximum gain of the antenna is 13.1 dB at 4.6 GHz and 17.1 dB at 5.5 GHz. The 3 dB gain bandwidth of the antenna is 11.9 and 8.2% at two frequency bands, respectively. The gain bandwidth is wider than the impedance bandwidth, so the impedance bandwidth is the working bandwidth of the antenna. The simulated and measured results show that the antenna can realize frequency reconfiguration through tuning the states of the diodes on the PRS. Meanwhile, the antenna obtains well radiation characteristic in both frequencies.

## **4. Pattern reconfigurable Fabry-Pérot cavity antenna**

#### **4.1 Design of the configurable PRS**

Firstly, we design a reflection phase reconfigurable unit cell, as shown in **Figure 10**. The unit cell is printed on a substrate of FR4.4 with a thickness of 1.6 mm and permittivity of 4.4. It consists of a square patch and a square ring. A slot is inserted into the unit cell to separate the patch and the ring. Three PIN diodes are

**Figure 10.** *The structure of the unit cell.*

#### *Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces DOI: http://dx.doi.org/10.5772/intechopen.91695*

inserted in the slot. The diodes are divided into two groups. The diodes on the ring are named K1 and the diode in the patch is named K2. The different states of two group diodes give the four cell states, as shown in **Table 1**.

The unit cell is simulated by the CST. The simulation setting of the unit cell is the same as the unit cell presented in Section 2. The reflection coefficients of the unit cell under *y*-polarized wave are shown in **Figure 11**. The simulation results show that the reflection phase of the unit cell at 5 GHz ranged from 165 to 262°. This suggests that the unit cell has a large range of reflection phase variation. Meanwhile, the reflection magnitudes of the unit cell in four states are always greater than 0.7. This can ensure a high gain of the FPC antenna.

A reconfigurable PRS is formed by the unit cell shown in **Figure 10**. The structure of the PRS is shown in **Figure 12**. It is composed by 6 6 unit cells and divided


#### **Table 1.**

when *f* = 4.6 GHz and the diodes are ON, and the red line is the relationship when *f* = 5.5 GHz and the diodes are OFF. The two lines intersect at the point *w* = 8 mm and *h* = 27.2 mm. This indicates that when *h* = 27.2 mm and *w* = 8 mm, the antenna can work at 4.6 GHz if the diodes are ON and at 5.5 GHz if the diodes are OFF. So the antenna realizes frequency reconfiguration through tuning the states of the diodes.

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

The photographs of the fabricated antenna are shown in **Figure 7**. The red lines connect the anode of the DC source, and the black lines connect the cathode. The S11 and radiation patterns of the antenna at two frequencies are measured. Good agreement between the simulated and measured results is obtained. The simulated and measured S11 of the antenna is shown in **Figure 8**. The 10 dB impedance band of the antenna is 4.55–4.7 GHz (3.3%) when diodes are ON and 5.37–5.63 GHz (4.7%) when diodes are OFF. The measured center of the low-frequency band is higher than the simulated result. This may be caused by the fabrication errors. During the welding of SMA joints, some unnecessary metals may be introduced.

**Figure 9** presents the simulated and measured radiation patterns of the antenna. **Figure 9a** shows the radiation patterns at 4.6 GHz and **Figure 9b** shows at 5.5 GHz. The measured maximum gain of the antenna is 13.1 dB at 4.6 GHz and 17.1 dB at 5.5 GHz. The 3 dB gain bandwidth of the antenna is 11.9 and 8.2% at two frequency bands, respectively. The gain bandwidth is wider than the impedance bandwidth, so the impedance bandwidth is the working bandwidth of the antenna. The simulated and measured results show that the antenna can realize frequency reconfiguration through tuning the states of the diodes on the PRS. Meanwhile, the antenna obtains

This results in the deviation of the resonant frequency of the antenna.

**3.3 Fabrication and measurement of the antenna**

well radiation characteristic in both frequencies.

**4.1 Design of the configurable PRS**

**Figure 10.**

**230**

*The structure of the unit cell.*

**4. Pattern reconfigurable Fabry-Pérot cavity antenna**

Firstly, we design a reflection phase reconfigurable unit cell, as shown in **Figure 10**. The unit cell is printed on a substrate of FR4.4 with a thickness of 1.6 mm and permittivity of 4.4. It consists of a square patch and a square ring. A slot is inserted into the unit cell to separate the patch and the ring. Three PIN diodes are *The four states of the unit cell.*

**Figure 11.** *The reflection phase of the unit cell.*

**Figure 12.** *The structure of the PRS: (a) top view and (b) bottom view.*

into two parts. The reflection phase of two parts of the PRS can be controlled by tuning the states of the PIN diodes. Some DC-biased circuit is added on the PRS to bias the PIN diodes. Firstly, since the square rings in the unit cells form a net structure when combing the PRS and the K1 is on the net structure, so two lines of capacitors (220 pF) are added between the two parts to bias the K1 in different parts of the PRS independently. In **Figure 12a**, the points V1 and V2 are used to bias the K1 in different parts. Then, in order to bias the K2, two metalized via hole are added into each unit cell to connect two parts of the patch, and the biasing lines on the bottom side of the PRS which are shown in **Figure 12b**. The points V3 and V4 are used to bias the K2 in different parts of the PRS. The Gnd on the top and bottom side of the PRS are connected by a metalized via hole. Some inductors (20 nH) are inserted into the lines to prevent the RF signal from going into the DC sources. With the designed biasing lines, the K1 and K2 in different parts of the PRS can be controlled independently. The biasing lines are very thin (0.2 mm), and the diameter of the metalized via holes is tiny enough (0.3 mm). So the simulated results show that the DC biasing circuits have barely influence on the performance of the metasurface.

Through controlling the states of the diodes, the PRS will present different reflection phase distributions, so the beam of the antenna will tilt toward different directions. Due to the large reflection phase variation range, the antenna obtains a large steering angle. The antenna operates at 5 GHz and the reflection phase gradient of the PRS is along *x*- direction, so the beam of the antenna will tilt in *xoz* plane. We calculate the height of the cavity when the PRS is in the state of S1. After being

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces*

**States Part 1 Part 2 Beam direction (°) Gain (dB)**

 OFF OFF OFF OFF 0 11.1 OFF ON OFF OFF 6 9.8 OFF OFF OFF ON 6 9.8 ON ON OFF OFF 18 8.7 OFF OFF ON ON 18 8.7 ON OFF ON ON 24 7.6 ON ON ON OFF 24 7.6 ON OFF OFF ON 46 6.6 OFF ON ON OFF 46 6.6 ON OFF OFF OFF 54 6.3 OFF OFF ON OFF 54 6.3

**K1 K2 K1 K2**

optimized by the CST, the *h* is set to 30.5 mm finally.

*DOI: http://dx.doi.org/10.5772/intechopen.91695*

**Figure 14.**

**Table 3.**

**233**

*Detail information of the antenna at different states.*

*Photograph of the fabricated antenna.*

#### **4.2 Antenna design**

From the theoretical analysis in Section 1, when the PRS has a uniform reflection phase, the antenna radiates toward the broadside direction [16]. And the main beam of the antenna is perpendicular to the antenna surface. However, when the PRS have reflection phase gradient, the main beam of the antenna will be tilted. So when the two parts of the PRS have different reflection phases, the beam of the antenna will tilt to the direction that has lagging phase.

The structure of the FPC antenna is shown in **Figure 12**, and the values of the parameters of the antenna are shown in **Table 2**. The FPC antenna is feed by a slot coupled patch antenna, as shown in **Figure 13a**. The feed antenna is set under the PRS with a distance of *h*. The side view of the FPC antenna is shown in **Figure 13b**.


#### **Table 2.**

*Values of the parameters (mm).*

**Figure 13.** *Structure of the antenna: (a) top view of the feeding antenna and (b) side view of the FPC antenna.*

*Reconfigurable Fabry-Pérot Cavity Antenna Basing on Phase Controllable Metasurfaces DOI: http://dx.doi.org/10.5772/intechopen.91695*

Through controlling the states of the diodes, the PRS will present different reflection phase distributions, so the beam of the antenna will tilt toward different directions. Due to the large reflection phase variation range, the antenna obtains a large steering angle. The antenna operates at 5 GHz and the reflection phase gradient of the PRS is along *x*- direction, so the beam of the antenna will tilt in *xoz* plane. We calculate the height of the cavity when the PRS is in the state of S1. After being optimized by the CST, the *h* is set to 30.5 mm finally.

**Figure 14.** *Photograph of the fabricated antenna.*

into two parts. The reflection phase of two parts of the PRS can be controlled by tuning the states of the PIN diodes. Some DC-biased circuit is added on the PRS to bias the PIN diodes. Firstly, since the square rings in the unit cells form a net structure when combing the PRS and the K1 is on the net structure, so two lines of capacitors (220 pF) are added between the two parts to bias the K1 in different parts of the PRS independently. In **Figure 12a**, the points V1 and V2 are used to bias the K1 in different parts. Then, in order to bias the K2, two metalized via hole are added into each unit cell to connect two parts of the patch, and the biasing lines on the bottom side of the PRS which are shown in **Figure 12b**. The points V3 and V4 are used to bias the K2 in different parts of the PRS. The Gnd on the top and bottom side of the PRS are connected by a metalized via hole. Some inductors (20 nH) are inserted into the lines to prevent the RF signal from going into the DC sources. With the designed biasing lines, the K1 and K2 in different parts of the PRS can be controlled independently. The biasing lines are very thin (0.2 mm), and the diameter of the metalized via holes is tiny enough (0.3 mm). So the simulated results show that the DC biasing circuits have barely influence on the performance of the

*Advanced Radio Frequency Antennas for Modern Communication and Medical Systems*

From the theoretical analysis in Section 1, when the PRS has a uniform reflection phase, the antenna radiates toward the broadside direction [16]. And the main beam of the antenna is perpendicular to the antenna surface. However, when the PRS have reflection phase gradient, the main beam of the antenna will be tilted. So when the two parts of the PRS have different reflection phases, the beam of the antenna

The structure of the FPC antenna is shown in **Figure 12**, and the values of the parameters of the antenna are shown in **Table 2**. The FPC antenna is feed by a slot coupled patch antenna, as shown in **Figure 13a**. The feed antenna is set under the PRS with a distance of *h*. The side view of the FPC antenna is shown in **Figure 13b**.

*p L***s** *W***s** *g g***<sup>0</sup>** *L***p** *L h* 15 10 4.25 1.5 1.5 10.4 100 30.5

*Structure of the antenna: (a) top view of the feeding antenna and (b) side view of the FPC antenna.*

metasurface.

**Table 2.**

**Figure 13.**

**232**

*Values of the parameters (mm).*

**4.2 Antenna design**

will tilt to the direction that has lagging phase.


#### **Table 3.**

*Detail information of the antenna at different states.*
