3. Compact, planar, and wideband monopole filtennas

Two electrically small, efficient planar monopole filtennas based on capacitively loaded loop (CLL) resonators are presented. Taking advantage of the characteristics of filters that are based on a pair of electrically coupled CLL resonators, the filtenna is designed, fabricated, and measured. The experimental results demonstrate that this electrically small system had a 6.27% fractional impedance bandwidth, high out-of-band rejection, and stable omnidirectional radiation patterns. An additional

(2.989) to 11.010 GHz (10.842 GHz). Two transmission zeros are generated at

Realized gain of the arc-slot modified UWB antenna at (a) 3.0, (b) 6.5, and (c) 10 GHz.

2.3 Integration of a UWB filter into an antenna with an arc-shaped slot

the antenna with the single-wing filter is from 2.995 (2.949) to 11.047 GHz

The single-wing filter was integrated into the arc-slot antenna as shown in Figure 5. The filter was connected directly to the microstrip feedline section. As shown in Figure 5, the UWB filter-antenna design was optimized, fabricated, and measured. As depicted, the simulated (measured) 10 dB impedance bandwidth of

2.12 GHz (2.085 GHz) and at 11.5 GHz (11.449 GHz).

Electromagnetic Materials and Devices

Figure 3.

254

#### Figure 5.

The UWB antenna with both the arc-slot and the multimode resonator filter T. (a) Design model of the antenna, (b) fabricated prototype, and (c) its simulated and measured |S11|.

CLL structure, as a near-field resonant parasitic (NFRP) element, is then integrated systematically into the system to achieve a wider operational bandwidth. The resulting filtenna owns a 7.9% fractional bandwidth, together with a flat gain response, stable omnidirectional radiation patterns, and high out-of-band rejection characteristics.

### 3.1 Square CLL-based bandpass filter design

A bandpass filter with a 0° feed structure based on rectangular microstrip CLL [14] is revealed in Figure 8(a). The plane is symmetrical about the dashed lines O–O<sup>0</sup> and T–T<sup>0</sup> along the x- and y-axis, respectively. The Rogers substrate modeled 4350B, with permittivity ε<sup>r</sup> = 3.48, dielectric loss tangent tan σ = 0.0037, and permeability μ<sup>r</sup> = 1, was chosen to construct the filter. The total size of it is

29 27 1 mm. For this design, two square CLLs oriented with each other gap to gap were etched on the substrate. This geometry introduced an electrical coupling between the two components [14–16], for instance, which has been exploited previously to efficiently improve the microwave field transmission by a metallic aperture with subwavelength [17]. As depicted in Figure 8(a), the two feed ports of the

Realized gain of the arc-slot modified UWB antenna with the single-wing filter at (a) 3.0, (b) 6.5, and

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas

DOI: http://dx.doi.org/10.5772/intechopen.82305

Figure 6.

257

(c) 10 GHz.

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas DOI: http://dx.doi.org/10.5772/intechopen.82305

Figure 6.

CLL structure, as a near-field resonant parasitic (NFRP) element, is then integrated systematically into the system to achieve a wider operational bandwidth. The resulting filtenna owns a 7.9% fractional bandwidth, together with a flat gain response, stable omnidirectional radiation patterns, and high out-of-band rejection

The UWB antenna with both the arc-slot and the multimode resonator filter T. (a) Design model of the

A bandpass filter with a 0° feed structure based on rectangular microstrip CLL [14] is revealed in Figure 8(a). The plane is symmetrical about the dashed lines O–O<sup>0</sup> and T–T<sup>0</sup> along the x- and y-axis, respectively. The Rogers substrate modeled 4350B, with permittivity ε<sup>r</sup> = 3.48, dielectric loss tangent tan σ = 0.0037, and permeability μ<sup>r</sup> = 1, was chosen to construct the filter. The total size of it is

characteristics.

256

Figure 5.

3.1 Square CLL-based bandpass filter design

Electromagnetic Materials and Devices

antenna, (b) fabricated prototype, and (c) its simulated and measured |S11|.

Realized gain of the arc-slot modified UWB antenna with the single-wing filter at (a) 3.0, (b) 6.5, and (c) 10 GHz.

29 27 1 mm. For this design, two square CLLs oriented with each other gap to gap were etched on the substrate. This geometry introduced an electrical coupling between the two components [14–16], for instance, which has been exploited previously to efficiently improve the microwave field transmission by a metallic aperture with subwavelength [17]. As depicted in Figure 8(a), the two feed ports of the

Figure 7.

The maximum simulated and measured realized gain values in the broadside direction and any direction for the three UWB antennas.

filter, which were connected to 50-Ω microstrip lines (with the width 2.2 mm), are placed to be centrally symmetric about the midpoint of line O–O<sup>0</sup> , so as to create another two transmission zeros in the stopband, and the passband response remains the same. As presented in [14], 0° feed geometry is superior to having two CLLs on one side.

Figure 8(b) correspondingly reveals the equivalent circuit with lumped elements. Its L and C values represent the natural self-inductance and self-capacitance of the uncoupled resonators alone. Notation Cm denotes the two resonators' mutual capacitance. When the symmetrical line T–T<sup>0</sup> is substituted by an electric wall (a short circuit), the corresponding circuit has a lower resonant frequency fe = 1/{2π [L(C + Cm)]1/2}. In the same way, when it is replaced by a magnetic wall (an open circuit), the corresponding circuit has a higher resonant frequency fm = 1/{2π [L(<sup>C</sup> � Cm)]1/2} [14].

the symmetrical line labeled T–T<sup>0</sup>

ke ¼ �

1 2

enhanced bandwidth.

Figure 8.

259

. Furthermore, through decreasing the distance d1

continually, it produced even stronger mutual coupling, generating an even larger Cm value. As a result, both fe and fm departed each other away from f<sup>0</sup> as shown in Figure 9. Thus, with a narrower distance d1 for the filter configuration in Figure 8(a), a much stronger coupling between the CLL resonators could be observed, and this case causes the corresponding two poles to depart from each other, leading to an

Filter with electrically coupled resonators. (a) Design layout. (b) Equivalent circuit network [15].

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas

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This can be verified by calculating the electric coupling coefficient (ke) between

f 2 <sup>m</sup> þ f 2 e

� � ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi f 2 <sup>m</sup> � f 2 e

� <sup>f</sup> 2 <sup>2</sup> � f 2 1

f 2 <sup>2</sup> þ f 2 1

 !<sup>2</sup> vuut (1)

!<sup>2</sup>

two resonators. It is readily obtained using the expression [15].

f 2 f 1 þ f 1 f 2

Figure 9 demonstrates the simulated S-parameters of the filter, while varying the distance between the two resonators (d1) from 1.1 to 0.3 mm. The results show that when the distance d1 increases large enough (e.g., larger than 1.1 mm), the resonant frequency f0 remains unchanged, and also the resonant intensity (|S11| dip) at f0 presents slight variation. Because of the slight mutual coupling between the resonators, it hardly impacts the resonant frequency f0, and the mutual capacitance approaches zero, i.e., Cm ≈ 0. Therefore, in this scarcely coupled condition, fe ≈ fm ≈ f0 = 1/[2π(LC) 1/2] is achieved. For comparison, when the distance d1 decreases more enough (e.g., smaller than 0.7 mm), the resonant frequency f0 results to be completely divided into two adjacent frequencies, i.e., fe and fm, which contribute to enhance the passband. It could be attained from the simulations that the surface currents of the two CLLs are in phase at the lower resonant frequency fe. This conclusion agrees well with the selection of the electric wall substituting the plane labeled T–T<sup>0</sup> . For another, at higher resonance frequency fm, the surface currents of the split rings are out of phase. The magnetic wall agrees well to replace Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas DOI: http://dx.doi.org/10.5772/intechopen.82305

Figure 8. Filter with electrically coupled resonators. (a) Design layout. (b) Equivalent circuit network [15].

the symmetrical line labeled T–T<sup>0</sup> . Furthermore, through decreasing the distance d1 continually, it produced even stronger mutual coupling, generating an even larger Cm value. As a result, both fe and fm departed each other away from f<sup>0</sup> as shown in Figure 9. Thus, with a narrower distance d1 for the filter configuration in Figure 8(a), a much stronger coupling between the CLL resonators could be observed, and this case causes the corresponding two poles to depart from each other, leading to an enhanced bandwidth.

This can be verified by calculating the electric coupling coefficient (ke) between two resonators. It is readily obtained using the expression [15].

$$k\_{\epsilon} = \pm \frac{1}{2} \left( \frac{f\_2}{f\_1} + \frac{f\_1}{f\_2} \right) \sqrt{\left( \frac{f\_m^2 - f\_\epsilon^2}{f\_m^2 + f\_\epsilon^2} \right)^2 - \left( \frac{f\_2^2 - f\_1^2}{f\_2^2 + f\_1^2} \right)^2} \tag{1}$$

filter, which were connected to 50-Ω microstrip lines (with the width 2.2 mm), are

The maximum simulated and measured realized gain values in the broadside direction and any direction for the

another two transmission zeros in the stopband, and the passband response remains the same. As presented in [14], 0° feed geometry is superior to having two CLLs on

Figure 8(b) correspondingly reveals the equivalent circuit with lumped elements. Its L and C values represent the natural self-inductance and self-capacitance of the uncoupled resonators alone. Notation Cm denotes the two resonators' mutual capacitance. When the symmetrical line T–T<sup>0</sup> is substituted by an electric wall (a short circuit), the corresponding circuit has a lower resonant frequency fe = 1/{2π [L(C + Cm)]1/2}. In the same way, when it is replaced by a magnetic wall (an open circuit), the corresponding circuit has a higher resonant frequency fm = 1/{2π

Figure 9 demonstrates the simulated S-parameters of the filter, while varying the distance between the two resonators (d1) from 1.1 to 0.3 mm. The results show that when the distance d1 increases large enough (e.g., larger than 1.1 mm), the resonant frequency f0 remains unchanged, and also the resonant intensity (|S11| dip) at f0 presents slight variation. Because of the slight mutual coupling between the resonators, it hardly impacts the resonant frequency f0, and the mutual capacitance

1/2] is achieved. For comparison, when the distance d1

. For another, at higher resonance frequency fm, the surface

approaches zero, i.e., Cm ≈ 0. Therefore, in this scarcely coupled condition,

decreases more enough (e.g., smaller than 0.7 mm), the resonant frequency f0 results to be completely divided into two adjacent frequencies, i.e., fe and fm, which contribute to enhance the passband. It could be attained from the simulations that the surface currents of the two CLLs are in phase at the lower resonant frequency fe. This conclusion agrees well with the selection of the electric wall substituting the

currents of the split rings are out of phase. The magnetic wall agrees well to replace

, so as to create

placed to be centrally symmetric about the midpoint of line O–O<sup>0</sup>

one side.

Figure 7.

three UWB antennas.

Electromagnetic Materials and Devices

[L(<sup>C</sup> � Cm)]1/2} [14].

fe ≈ fm ≈ f0 = 1/[2π(LC)

plane labeled T–T<sup>0</sup>

258

Figure 9. S-parameters of the simulation for the electrically coupled filter while the separate distance d1 varying from 1.1 to 0.3 mm, together with the surface current distribution behaviors on the resonance at fe and fm.

where f<sup>1</sup> and f<sup>2</sup> indicate the resonance frequencies of each independent CLL resonator. Because the capacitance Cm in Figure 8 is positive, the plus sign is selected. Furthermore, because the two CLL resonators are identical, one knows that f<sup>1</sup> = f2. Thus, Eq. (1) reduces to the following:

$$k\_{\epsilon} = \frac{f\_{\;m}^{2} - f\_{\;\epsilon}^{2}}{f\_{\;m}^{2} + f\_{\;\epsilon}^{2}}.\tag{2}$$

The measured and simulated |S11| values are presented in Figure 11. The measured values confirmed that the filtenna had a 10 dB impedance bandwidth from 2.24 to 2.385 GHz (6.27% fractional impedance bandwidth) in good agreement with the simulated values 2.252–2.398 GHz (6.28% fractional impedance bandwidth). The electrical size of the measured prototype is ka 0.93, while its simulated value was 0.935. Figure 11 demonstrates that the prototype filtenna has a flat realized gain response within its passband. The measured (simulated) peak value was 1.15 (1.41) dBi. The simulated radiation efficiency was higher than 80.93% throughout the operational band. This realized filtenna prototype clearly has very good band-edge

Measured and simulated |S11| and realized gain values of the first filtenna as functions of the source frequency.

First electrically small filtenna. (a) Top and (b) side views of the HFSS simulation model. (c) Front and back

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas

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For many applications, it is desirable to have an even wider bandwidth. Consequently, the second design shown in Figure 12 was considered. In order to improve the flatness of the transmission performance within the passband while maintaining its wideband operation and steep skirts, a third resonator was introduced without

selectivity and stopband suppression.

Figure 10.

Figure 11.

261

views of the fabricated prototype.

3.3 Filtenna with enhanced bandwidth

increasing the total overall dimension of the filtenna.

Consequently, as expected for the uncoupled, weak coupling, and strong coupling cases given in Figure 9, ke is, respectively, 0, 0.14, and 0.49.

#### 3.2 Electrically small filtenna design

A filtenna having a second-order filter was co-designed and optimized. It is shown in Figure 10. Figure 10(a) indicates that one CLL element acts as the directly driven element. A fan-shaped radiator with no ground plane on the back side of it acts as a NFRP element in the presence of the monopole (CLL-based) antenna [18]. The choice of this special fan-shaped radiator establishes an even smoother impedance transition over the desired wider bandwidth. Simply starting with the resonance frequencies near to each other facilitates a straightforward numerical approach to optimize and finalize the actual antenna design. Note that the fan-shaped part of the NFRP element is placed on the opposite side of the feed port. This arrangement facilitates the creation of dual transmission zeros on the two edges of the passband. This arrangement enhances the out-of-band rejection level.

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas DOI: http://dx.doi.org/10.5772/intechopen.82305

#### Figure 10.

where f<sup>1</sup> and f<sup>2</sup> indicate the resonance frequencies of each independent CLL resonator. Because the capacitance Cm in Figure 8 is positive, the plus sign is selected. Furthermore, because the two CLL resonators are identical, one knows

S-parameters of the simulation for the electrically coupled filter while the separate distance d1 varying from 1.1

to 0.3 mm, together with the surface current distribution behaviors on the resonance at fe and fm.

ke <sup>¼</sup> <sup>f</sup> 2 <sup>m</sup> � f 2 e

pling cases given in Figure 9, ke is, respectively, 0, 0.14, and 0.49.

f 2 <sup>m</sup> þ f 2 e

Consequently, as expected for the uncoupled, weak coupling, and strong cou-

A filtenna having a second-order filter was co-designed and optimized. It is shown in Figure 10. Figure 10(a) indicates that one CLL element acts as the directly driven element. A fan-shaped radiator with no ground plane on the back side of it acts as a NFRP element in the presence of the monopole (CLL-based) antenna [18]. The choice of this special fan-shaped radiator establishes an even smoother impedance transition over the desired wider bandwidth. Simply starting with the resonance frequencies near to each other facilitates a straightforward numerical approach to optimize and finalize the actual antenna design. Note that the fan-shaped part of the NFRP element is placed on the opposite side of the feed port. This arrangement facilitates the creation of dual transmission zeros on the two edges of the passband. This arrangement enhances the out-of-band rejection level.

: (2)

that f<sup>1</sup> = f2. Thus, Eq. (1) reduces to the following:

3.2 Electrically small filtenna design

Electromagnetic Materials and Devices

Figure 9.

260

First electrically small filtenna. (a) Top and (b) side views of the HFSS simulation model. (c) Front and back views of the fabricated prototype.

Figure 11. Measured and simulated |S11| and realized gain values of the first filtenna as functions of the source frequency.

The measured and simulated |S11| values are presented in Figure 11. The measured values confirmed that the filtenna had a 10 dB impedance bandwidth from 2.24 to 2.385 GHz (6.27% fractional impedance bandwidth) in good agreement with the simulated values 2.252–2.398 GHz (6.28% fractional impedance bandwidth). The electrical size of the measured prototype is ka 0.93, while its simulated value was 0.935. Figure 11 demonstrates that the prototype filtenna has a flat realized gain response within its passband. The measured (simulated) peak value was 1.15 (1.41) dBi. The simulated radiation efficiency was higher than 80.93% throughout the operational band. This realized filtenna prototype clearly has very good band-edge selectivity and stopband suppression.

#### 3.3 Filtenna with enhanced bandwidth

For many applications, it is desirable to have an even wider bandwidth. Consequently, the second design shown in Figure 12 was considered. In order to improve the flatness of the transmission performance within the passband while maintaining its wideband operation and steep skirts, a third resonator was introduced without increasing the total overall dimension of the filtenna.

#### Figure 12.

Enhanced bandwidth filtenna with slots in its ground strip. (a) Top and (b) back views of the HFSS simulation model. (c) Front and back views of the fabricated prototype.

4. Compact filtennas with enhanced bandwidth

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas

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σ = 0.0037.

Figure 13.

[21, 22].

263

4.1 Design of compact filtennas

bandwidth filtenna with slots in its ground strip.

4.1.1 A CLL-based filtenna design

monopole and the other to the SMA.

Two filtennas are proposed by a design strategy with the merits of both a compact structure and enhanced bandwidth. The reliability of the filtennas is verified though simulations and analysis of a compact NFRP filtenna which is proposed and fabricated. The reported design employs a Rogers 4350B substrate with relative permittivity ε<sup>r</sup> = 3.48, relative permeability μ<sup>r</sup> = 1.0, and dielectric loss tangent tan

Measured and simulated |S11| and realized gain values as functions of the source frequency for the enhanced

A well-designed compact NFRP antenna is selected as the radiator [19, 20]. Then a compact NFRP antenna is designed, which consists of a traditional monopole and a rectangular microstrip capacitively loaded loop (CLL)-based band-pass filter

The elaborate geometry of the filtenna is shown in Figure 14. As depicted in Figure 14(a) and (e), the compact electrically small antenna (ESA) with NFRP was chosen as the radiating element. The NFRP element is proposed to etch upon one side of the semi-circle board, while the monopole microstrip is located on the other side, with the design principle corresponding to the reported NFRP ESAs [23–25]. The composite structure of this radiator element and filtering element, which is based on CLL resonators, is well shown in Figure 14(a)–(d). The enlarged filter is shown in Figure 14(b). This filter structure is a typical band-pass design [21, 26, 27], and is set to be symmetric about the S–S<sup>0</sup> line. One end is connected to the printed

The third resonator is an additional CLL element, shown in blue in Figure 12(a). Its gap position coincides with the driven CLL element, and it has an arm included to facilitate its coupling to the NFRP element. This collocated arrangement of the two CLLs provides a means to control the mutual coupling, further expanding the bandwidth without increasing the total overall dimensions of the filtenna. Three slots were etched in the ground strip directly beneath the two CLL elements to achieve a smoother realized gain curve. The length of the additional CLL element is set nearly equal to the driven CLL's size to make their resonance frequencies close to one another.

The simulated and measured |S11| and realized gain values of the second filtenna with the ground strip slots are given in Figure 13. The simulated (measured) realized gain values indicate that the simulated peak realized gain value is improved from 1.659 to 1.75 dBi. The corresponding measured value is 1.376 dBi, revealing more losses than expected in fabrication. For the simulated |S11| values exhibited in Figure 13, the impedance bandwidth ranges from 2.264 to 2.46 GHz (about 8.3% fractional bandwidth, i.e., a 32.2% improvement) and was from 2.261 to 2.447 GHz (7.9% fractional bandwidth, i.e., a 26% improvement) in the measurement. Similarly, the simulated ka 0.94 and measured ka 0.938 values verify that the filtenna is electrically small. Furthermore, the simulated radiation efficiency across the entire operational bandwidth is higher than 82.87%. Again, very good agreement between the simulated and measured performance characteristics was obtained.

Compact, Efficient, and Wideband Near-Field Resonant Parasitic Filtennas DOI: http://dx.doi.org/10.5772/intechopen.82305

Figure 13. Measured and simulated |S11| and realized gain values as functions of the source frequency for the enhanced bandwidth filtenna with slots in its ground strip.
