*3.2.2 RF frequency measurement*

*UWB Technology - Circuits and Systems*

schematically depicted in **Figure 16**.

consequence the Video output at its maximum values.

*Simplified schematic of an extended DYNAMIC range DLVA.*

uncommon.

range.

feasibility.

**Figure 15.**

of processing RF signals up to tenths of GHz. Diode operating at 50 GHz are not

Another way to increase the system's dynamic range is to use a SDLVA topology,

An SDLVA (Successive Detection Logarithmic Amplifier) is similar to a DLVA however, the SDLVA circuit is designed in such a way that it does not need a detector before the logarithmic video amplifier. The SDLVA uses multiple compressive stages of RF gain to emulate the exponential transfer function. The output of each stage is coupled into a linear detector. The detector operates over a narrower dynamic range, which means that more detectors are needed to cover the same dynamic

The principle of operation is the following: when the RF signal power is low, all amplifiers operate in a linear condition (i.e. the output power is proportional to the input power) and consequently the DC voltage provided by the diode detector is proportional too. As the power increases, the final stages begin to saturate and their output is capped to a saturation value. Therefore, any additional output voltage will be delivered only from the first stages until they saturate too, saturating as a

The typical dynamic range of each detector is approximately 10 dB, which are then summed in a single video amplifier so as to provide a single detected output. The overall dynamic range is 10 × *N* dB where *N* is the number of amplifiers. *N* = 7–8 represents an acceptable trade-off between high dynamic range and circuit

**32**

**Figure 16.**

*SDLVA simplified schematic.*

Microwave circuits can be profitably employed to perform an estimation of the carrier frequency of the incoming RF signal. This is all but a simple task. Nowadays, the advance of digital components has made feasible digital frequency measurement up to 10 GHz and beyond. Even in this case, some form of microwave frontend is required to prepare the signal for digital sampling. Above tenths of GHz, and considering UWB signals, pure digital frequency estimation becomes unpractical and microwave circuits have to be inserted.

UWB microwave circuits that perform frequency estimation are referred to as Instantaneous Frequency Measurement (IFM), whose very simplified schematic diagram is reported in **Figure 17**.

The RF input *vin*(*t*) = *A cos*(*ωt*), where *ω* = *2f* and *f* is the instantaneous frequency carrier, is hard-limit amplified and then split by a power divider in two equal amplitude signals, one of which is delayed with respect to the other through a delay line. Hard limitation consist in amplifying all incoming signal to a fixed power level—obviously within a feasible dynamic range. The role of the delay line is to out-phase the two signals coming from the power divider's outputs by a quantity proportional to the frequency carrier.

The higher the frequency the more the two signals applied to the mixer will be out-phased.

Considering the relationship IF = |*m\*RF + n\*LO*| already introduced in Section 3.1.1, we have that the valuable mixer's output will be a zero frequency product and therefore a DC voltage since *RF* and *LO* signals have, of course, the same carrier frequency. The amplitude of the DC voltage is proportional to cosine of the outphase between the two signals and therefore to the carrier frequency, trough the relationship:

$$\text{Vout}(t) = K \cos(\Delta \Phi),\tag{2}$$

and

$$
\Delta\Phi = 2\pi \ast \Delta L \ast f / \upsilon\_p \tag{3}
$$

where Δ*L* is the difference between the physical distance of the delay line and the direct pat; *f* is the carrier frequency, while *vp* the speed of the EM wave in the medium. Higher order mixing terms are eliminated by the low-pass filter (LPF). The BW of such circuit is limited by the BW the power divider and the mixer. As a consequence, the delay is synthesized to implement *π* shift (maximum out-phase) at the components maximum operating frequency.

Very often Vout is digitized by means of a N-BIT analog-to-digital converter (ADC). This helps the subsequent stages since the information is provided digitally, but the information accuracy is limited by the number of BITs (typically not more than four) and quantization effects. To overcome this issue the scheme in **Figure 18** is applied.

**Figure 17.** *IFM simplified schematic.*

**Figure 18.**

*Functional block diagram of a THREE-base harmonic IFM receiver.*

**Figure 19.** *Operational characteristics of a THREE-base harmonic IFM receiver.*

**35**

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications*

The input signal is split into *N* lines (three in the example depicted in **Figure 18**), the three signals are the delayed by appropriately dimensioned transmission liens. Each line has double the electrical length of the preceding section. Delay t1 is determined so that the signal measuring is unambiguous at the maximum operating frequency (worst case for ambiguity discrimination. In practice, the shortest time delay (t1) is used to resolve the frequency measurement ambiguities while the largest time delays (t2 = 2\*t1 and t3 = 4\*t1) provide a fine frequency measurement

The three output voltages (Vout 1, 2 and 3) values vs. input frequency are depicted in **Figure 19**. Vout 1 (dashed line) is the only transfer function that is unambiguous vs. the input frequency. Therefore, this value is used to obtain a coarse yet unambiguous information of the input frequency. The other two voltages (Vout 2 and 3) suffer from ambiguity, but their resolution (in terms of ΔV/Δf) is better and therefore provides an ambiguous yet accurate information. The real frequency

If a larger BW is sought, then the topology depicted in **Figure 20** becomes

branches depends on the BW of RF IN; a larger BW requires a greater number of paths. Once the signal has been divided, it is filtered into sub-bands and then translated, through mixing, at the frequency of operation of the IFM. The sum of *RFn + LOn* must be constant for each branch and equal to the operating frequency of the IFM. In order to determine the actual RF IN frequency, a detector—not reported in **Figure 20**—must be inserted in each branch so the system can discriminate between the various possibilities. The branch in which the detector reads power corresponds to the sub-band of interest. In theory, the BW can be extended by increasing the number of adjacent sub-bands. There are practical limitations,

The RF input signal is split in two or more paths. The number of parallel output

The trend to integrate several function in one MMICs is unavoidable given the low SWaP-C constraints of modern electronic systems for high-end applications. At the moment it is possible to realize a full T/R module using only two MMICs, each one realized in the appropriate semiconductor therefore leveraging its benefits and peculiarities [13], and possibly in the near future there will be an all-on-one multi-functional RF MMIC. **Figure 21** depicts a transmit/receive module diagram, very similar to the one described in Section 2.2, where the area delimited by dotted lines indicate the two multifunctional MMICs that fulfil all TRM functionalities. The two MMICs are the Single Chip Front End (SCFE) and the Core-Chip (CC). The prior is directly connected to the antenna, while the latter to the microwave back-end and the DSP sections. Red and blue arrows indicate the direction of Rx and Tx signal. In some cases, black lines, the signal

As described in Section 2.2, RF TRMs are usually realized by interconnecting several functionalities fabricated on separate MMICs. Such an approach is somehow the result of the inability of a single compound semiconductor technology to properly carry out the main features normally required by an RF TRM. In some applications, such as active electronically scanned array (AESA), this multichip approach could result in a suboptimal overall system. Indeed, the density of the radiating

estimation is performed by correlating the three output voltages values.

and the number of sub-bands is seldom greater than four.

travels in both direction according to transmit or receive mode.

**4. Multi-functional integrated circuits**

**4.1 Single-chip front-end (SCFE)**

*DOI: http://dx.doi.org/10.5772/intechopen.87095*

accuracy Δf3 = Δf1/8.

useful.

**Figure 20.** *UWB frequency measurement circuit simplified schematic.*

### *UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications DOI: http://dx.doi.org/10.5772/intechopen.87095*

The input signal is split into *N* lines (three in the example depicted in **Figure 18**), the three signals are the delayed by appropriately dimensioned transmission liens. Each line has double the electrical length of the preceding section. Delay t1 is determined so that the signal measuring is unambiguous at the maximum operating frequency (worst case for ambiguity discrimination. In practice, the shortest time delay (t1) is used to resolve the frequency measurement ambiguities while the largest time delays (t2 = 2\*t1 and t3 = 4\*t1) provide a fine frequency measurement accuracy Δf3 = Δf1/8.

The three output voltages (Vout 1, 2 and 3) values vs. input frequency are depicted in **Figure 19**. Vout 1 (dashed line) is the only transfer function that is unambiguous vs. the input frequency. Therefore, this value is used to obtain a coarse yet unambiguous information of the input frequency. The other two voltages (Vout 2 and 3) suffer from ambiguity, but their resolution (in terms of ΔV/Δf) is better and therefore provides an ambiguous yet accurate information. The real frequency estimation is performed by correlating the three output voltages values.

If a larger BW is sought, then the topology depicted in **Figure 20** becomes useful.

The RF input signal is split in two or more paths. The number of parallel output branches depends on the BW of RF IN; a larger BW requires a greater number of paths. Once the signal has been divided, it is filtered into sub-bands and then translated, through mixing, at the frequency of operation of the IFM. The sum of *RFn + LOn* must be constant for each branch and equal to the operating frequency of the IFM. In order to determine the actual RF IN frequency, a detector—not reported in **Figure 20**—must be inserted in each branch so the system can discriminate between the various possibilities. The branch in which the detector reads power corresponds to the sub-band of interest. In theory, the BW can be extended by increasing the number of adjacent sub-bands. There are practical limitations, and the number of sub-bands is seldom greater than four.
