**3. Back-end systems and their circuits**

As briefly described in Section 1, a microwave back-end system is responsible for delivering the RF signal—or better its information content—to the ADC and consequently the DSP unit or to the low-frequency (often referred to as VIDEO) analogue stages.

Typically, this is accomplished through frequency conversion, when the DSP performs A/D sampling, or by performing some manipulation on the RF signal so its power and/or frequency component can be determined by the subsequent stages.

#### **3.1 Frequency conversion**

*UWB Technology - Circuits and Systems*

tubes and solid state circuits. With the advance of semiconductor technologies, vacuum tubes are becoming a legacy product. Nonetheless, they still provide a valuable solution when the power to be transmitted is in the order of tenths of kilowatts. Typically this requirement is related with a few avionic and spaceborne applications. The advantages of solid state device in terms of ruggedness, size, reliability, performance and cost are such that, whenever a solid state alternative becomes

*Key performance of an UWB 6–18 GHz GaAs TRM output power in TX (top) and gain in RX (bottom).*

Since its first appearance in R&D labs at the beginning of the new millennium, GaN has travelled a long way, and has now become the standard semiconductor, even in ADS systems, where reliability and process repeatability is a main concern. The advantages of GaN, over other III–V semiconductor, for high-power and highfrequency systems, reside on its capability to deliver a high amount of RF power in a small footprint, with little or none thermal management issues. Especially the last feature, make GaN attractive for ADS applications, often operating in harsh

MMIC GaN HPAs are capable of delivering hundreds of Watts at low microwave frequency (<5 GHz), tenths of watts at microwave frequencies (5–20 GHz), and

accessible, it quickly becomes adopted by the System Engineering team.

**26**

**Figure 8.**

thermos-mechanical environments.

some Watts even at millimetre-wave.

UWB downconverters and up-converters usually require multi-stage conversion plan since a single frequency conversion would not be able to eliminate all over spurs or leakage of the Local Oscillator (LO) signal. In fact, a very large sweep of LO frequency would be needed to down-convert the required portion of the large input RF BW into the smaller Intermediate Frequency (IF) BW. Therefore, at some point, there will be inevitably a strong intermodulation product or harmonic of LO that would fall in the IF BW. To overcome this issue a multi-stage frequency conversion topology, depicted in **Figure 9**, is advisable when the RF BW is large.

Here we will discuss in detail the down-converter architecture, but similar assumptions and design goals hold for the up-converter.

The first stage of the schematic depicted in **Figure 9**, is the filter bank, so the UWB signal RF IN signal is split into smaller adjacent sub-bands. Typically, each sub-band is less than an octave, and consequently the number of filters depends on

**Figure 9.** *Multi-stage down-converter topology.*

the ratio of the maximum to minimum RF IN frequency. In this way, the input RF signal is preselected ensuring that only the required portion of the input RF BW is fed to the following stages.

The next section consists in a frequency translation, performed by a set of mixers. Even in a downconverter system, often this first section of mixing performs an up-conversion. This is to shift the spurs to a very high frequency, well above the IF1 range. The value of LO1 is inversely proportional to the selected RF frequency. In this way, the sum of incoming RF IN signal and LO is constant and equal to IF1. Keep in mind that, the first stage is typically an up-converter even if the overall sub-system performs down-conversion. Some variable gain is also inserted in the RF chain to level out the gain response. In fact, as the input RF frequency increases, the RF losses become more evident and need to be compensated by lowering the attenuation accordingly.

At this point, all the desired signal falls within IF1, whose BW is much smaller than the UWB signal. RF IN, and can now be more easily down-converted to IF2 which is the sub-system's output frequency. A single LO2 frequency is sufficient in most cases, while in other UWB applications a variable LO2 could be required. In some extreme cases, for example when the RF BW is larger than a decade, a triple frequency conversion could be required [10].

This topology is definitively more complex than the single down-converter case, and requires a high frequency IF1 which often could be in the millimetre wavelength. Undoubtedly, such system complexity is the price to pay for having a UWB down-converter with high spurious free dynamic range (i.e. negligible spurs or LO harmonics falling in the IF OUT BW).

A key performance indicator of a frequency-converter is the Spurious-Free Dynamic-Range (SFDR). This parameter quantifies the ratio (expressed in dB) between the fundamental (desired tone) and the intermodulation product having the highest power inside the IF band. SFDR can be seen as an indicator of the down-converter's capability to perform its characteristic function without injecting unwanted signals at the output IF. Such unwanted signals, referred to as intermodulation components (defined in the following Section 3.1.1) can be mistaken by the receiver as low power real signals, but in reality they are an unwanted by-product of the real signal's down-conversion.

Intermodulation products are unavoidable, however the important matter is that they are below a given threshold therefore becoming undetectable and will not produce 'false signals' at system level. SFDR usually has a characteristic behaviour vs. frequency since, referring to **Figure 9**, LO1 and LO2 change in order to select different input RF bands, and therefore will produce different intermodulation orders when down-converting different RF input bands. **Figure 10** depicts the SFDR of an UWB 2–18 GHz downconverter [11] where a three Local Oscillators architecture is employed.

## *3.1.1 UWB mixers*

Mixer circuits are employed to translate the information applied to the RF carrier to a different frequency, namely IF, more easily processed by other circuits, typically of a digital nature. Given the incoming RF signal, and the LO signal, the frequency components at output of the mixer are:

$$\text{Output spectrum} = |\;m\*RF + n\*LO\;\vert\tag{1}$$

**29**

**Figure 11.**

**Figure 10.**

*Multi-stage down-converter SFDR vs. frequency.*

*Example of connectorized mixer (photo courtesy of Marki Micorwave).*

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications*

vice versa. In an up converter, *IF* appears instead of *RF* in the previous relation and

*LO* frequency is selected to trade-off between feasible *LO* values and harmful intermodulation products falling close to *IF* frequency. Undesirable intermodulation components are obtained when *m* ≠ 1 or *n* ≠ 1. The order of the intermodulation product is defined as: *order = m + n*. Usually the power level decreases as the

UWB mixers are typically realized through a double balanced, or double-double balanced (sometimes referred to as a triple) topology to eliminate the most annoying intermodulation and LO harmonics. Spurs and harmonic rejection is accomplished by appropriately combining the mixing signal through hybrid quadrature couplers (BALUNs or similar circuits). Ultimately, the UWB behaviour is limited by the coupling structures since the mixing device's behaviour can be considered ideal. UWB mixers often came in a connectorized package as the example depicted in **Figure 11**. It is also possible to obtain UWB Mixers in MMIC technology, however it is more difficult to realize UWB combiner/splitters in the confined dimensions

An extensive description on the of mixers and their property can be found in

Dr. Maas's comprehensive study: Microwave Mixers [12].

*DOI: http://dx.doi.org/10.5772/intechopen.87095*

*m = n* = 1 applies.

order increases.

required by MMICs.

where *m* and *n* are positive or negative or null integers. In a down-converter the desired Intermediate Frequency (IF) value is obtained when *m* = 1 and *n* = −1 or

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications DOI: http://dx.doi.org/10.5772/intechopen.87095*

vice versa. In an up converter, *IF* appears instead of *RF* in the previous relation and *m = n* = 1 applies.

*LO* frequency is selected to trade-off between feasible *LO* values and harmful intermodulation products falling close to *IF* frequency. Undesirable intermodulation components are obtained when *m* ≠ 1 or *n* ≠ 1. The order of the intermodulation product is defined as: *order = m + n*. Usually the power level decreases as the order increases.

UWB mixers are typically realized through a double balanced, or double-double balanced (sometimes referred to as a triple) topology to eliminate the most annoying intermodulation and LO harmonics. Spurs and harmonic rejection is accomplished by appropriately combining the mixing signal through hybrid quadrature couplers (BALUNs or similar circuits). Ultimately, the UWB behaviour is limited by the coupling structures since the mixing device's behaviour can be considered ideal. UWB mixers often came in a connectorized package as the example depicted in **Figure 11**. It is also possible to obtain UWB Mixers in MMIC technology, however it is more difficult to realize UWB combiner/splitters in the confined dimensions required by MMICs.

An extensive description on the of mixers and their property can be found in Dr. Maas's comprehensive study: Microwave Mixers [12].

**Figure 10.** *Multi-stage down-converter SFDR vs. frequency.*

**Figure 11.** *Example of connectorized mixer (photo courtesy of Marki Micorwave).*

*UWB Technology - Circuits and Systems*

fed to the following stages.

attenuation accordingly.

frequency conversion could be required [10].

harmonics falling in the IF OUT BW).

the real signal's down-conversion.

architecture is employed.

frequency components at output of the mixer are:

*3.1.1 UWB mixers*

the ratio of the maximum to minimum RF IN frequency. In this way, the input RF signal is preselected ensuring that only the required portion of the input RF BW is

The next section consists in a frequency translation, performed by a set of mixers. Even in a downconverter system, often this first section of mixing performs an up-conversion. This is to shift the spurs to a very high frequency, well above the IF1 range. The value of LO1 is inversely proportional to the selected RF frequency. In this way, the sum of incoming RF IN signal and LO is constant and equal to IF1. Keep in mind that, the first stage is typically an up-converter even if the overall sub-system performs down-conversion. Some variable gain is also inserted in the RF chain to level out the gain response. In fact, as the input RF frequency increases, the RF losses become more evident and need to be compensated by lowering the

At this point, all the desired signal falls within IF1, whose BW is much smaller than the UWB signal. RF IN, and can now be more easily down-converted to IF2 which is the sub-system's output frequency. A single LO2 frequency is sufficient in most cases, while in other UWB applications a variable LO2 could be required. In some extreme cases, for example when the RF BW is larger than a decade, a triple

This topology is definitively more complex than the single down-converter case,

and requires a high frequency IF1 which often could be in the millimetre wavelength. Undoubtedly, such system complexity is the price to pay for having a UWB down-converter with high spurious free dynamic range (i.e. negligible spurs or LO

A key performance indicator of a frequency-converter is the Spurious-Free Dynamic-Range (SFDR). This parameter quantifies the ratio (expressed in dB) between the fundamental (desired tone) and the intermodulation product having the highest power inside the IF band. SFDR can be seen as an indicator of the down-converter's capability to perform its characteristic function without injecting unwanted signals at the output IF. Such unwanted signals, referred to as intermodulation components (defined in the following Section 3.1.1) can be mistaken by the receiver as low power real signals, but in reality they are an unwanted by-product of

Intermodulation products are unavoidable, however the important matter is that they are below a given threshold therefore becoming undetectable and will not produce 'false signals' at system level. SFDR usually has a characteristic behaviour vs. frequency since, referring to **Figure 9**, LO1 and LO2 change in order to select different input RF bands, and therefore will produce different intermodulation orders when down-converting different RF input bands. **Figure 10** depicts the SFDR of an UWB 2–18 GHz downconverter [11] where a three Local Oscillators

Mixer circuits are employed to translate the information applied to the RF carrier to a different frequency, namely IF, more easily processed by other circuits, typically of a digital nature. Given the incoming RF signal, and the LO signal, the

Output spectrum = ∣ *m* ∗ *RF* + *n* ∗ *LO* ∣ (1)

where *m* and *n* are positive or negative or null integers. In a down-converter the desired Intermediate Frequency (IF) value is obtained when *m* = 1 and *n* = −1 or

**28**

#### **3.2 Microwave measurements**

In some applications, the incoming RF signal is unknown and therefore the system requires to quickly determine some of its key features, for example, carrier frequency and envelope power. Performing these measurements on UWB RF signals using only Digital HW can be complicated, or in some cases even impossible considering the power and frequency limitations of Silicon based technology. For this reason, microwave circuits have to be inserted in the receiving system to perform preliminary yet fast signal characterization.

### *3.2.1 RF power measurement*

The power measurement of an RF signal is commonly performed by appropriately feeding the RF signal to a diode—or to an array of diodes.

The diode has a well-known input-to-output square law characteristic, around the origin, when the diode is slightly forward biased. The signal coming out from the diode is composed by the even order harmonics of the incident RF signal and, most of all, its zero order term that is a DC voltage. The latter is captured by the following stages to perform the power measurement. There are several circuits that are capable of performing such functions, and their topology depends on the overall BW of the RF signal and also its dynamic range, i.e. the maximum to minimum power ratio to be analysed. A simplified block diagram of the diode detector is shown in **Figure 12**, together with the waveforms at each section of the circuit.

A detector logarithmic video amplifier (DLVA) consists of a diode detector followed by a logarithmic video amplifier (LVA), and its simplified schematic is given in **Figure 13**.

As explained before, the diode (detector circuit) converts RF signal into a DC voltage which is then fed to an amplifier with a logarithmic transfer function. Let us analyse the role of the two circuits.

The diode's quadratic law provides the DC voltage component of the incoming RF signal. The level of such DC voltage can vary significantly: from tenths of μV to some Volts. The ratio between the maximum and minimum voltage could be unacceptable for the subsequent processing stages. A LVA is therefore inserted to compress the dynamic and make it more usable for the following stages. In fact, the log amplifier will greatly amplify the weak signals leaving the strong signal practically unaffected. It is worthwhile noting LVA's operating BW is a few tenths of MHz

**31**

**Figure 14.**

**Figure 13.**

*DLVA simplified schematic.*

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications*

as opposed to the incoming signal which may be a few tenths of GHz. The dotted line in **Figure 14** depicts the linear relationship between input power in dBm and output power in Volts when using a LVA after the diode. The same **Figure 14** reports

It appears that the log relationship is very useful for performing power measurements, since its gradient is constant (approximately 25 mV/dB in **Figure 14**). The response without LVA is practically null up to −5 dBm and then rises instantaneously to +1.0 V after −5 dBm, making it unpractical. When dynamic range is not critical, a DLVA with a 30- or 40-dB dynamic range may provide sufficient perfor-

When the dynamic range of the incoming RF signal is greater than the one accepted by the diode, then the schematic depicted in **Figure 15** can be used to

Basically, the input signal is split in two paths: one path having high RF gain and one path having low RF gain hence an RF attenuated path. Then, the VIDEO signal

The principle of operation is the following: when the signal is low, only the amplified DLVA detects thanks to the RF gain (G) before the DLVA. When the signal power level becomes high, the gain path is practically saturated, delivering a constant voltage value, while the attenuated (Att) path performs the additional power measurement whose voltage is increased by the constant term coming from the saturated gain path. Finally, it is worthwhile noting that the BW of these circuits is practically limited by the BW of the RF circuits preceding the diode. In fact, the latter is often capable

*DOI: http://dx.doi.org/10.5772/intechopen.87095*

also the I/O characteristic without LVA, solid line.

mance to help capture and process all signals present.

coming out from both DLVAs is summed at the output.

*Output voltage (with and without LVA) as function of input power expressed in dBm.*

increase the system's dynamic range.

**Figure 12.** *Simplified schematic of a diode detector and waveforms at each node of the circuit.*

## *UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications DOI: http://dx.doi.org/10.5772/intechopen.87095*

as opposed to the incoming signal which may be a few tenths of GHz. The dotted line in **Figure 14** depicts the linear relationship between input power in dBm and output power in Volts when using a LVA after the diode. The same **Figure 14** reports also the I/O characteristic without LVA, solid line.

It appears that the log relationship is very useful for performing power measurements, since its gradient is constant (approximately 25 mV/dB in **Figure 14**). The response without LVA is practically null up to −5 dBm and then rises instantaneously to +1.0 V after −5 dBm, making it unpractical. When dynamic range is not critical, a DLVA with a 30- or 40-dB dynamic range may provide sufficient performance to help capture and process all signals present.

When the dynamic range of the incoming RF signal is greater than the one accepted by the diode, then the schematic depicted in **Figure 15** can be used to increase the system's dynamic range.

Basically, the input signal is split in two paths: one path having high RF gain and one path having low RF gain hence an RF attenuated path. Then, the VIDEO signal coming out from both DLVAs is summed at the output.

The principle of operation is the following: when the signal is low, only the amplified DLVA detects thanks to the RF gain (G) before the DLVA. When the signal power level becomes high, the gain path is practically saturated, delivering a constant voltage value, while the attenuated (Att) path performs the additional power measurement whose voltage is increased by the constant term coming from the saturated gain path.

Finally, it is worthwhile noting that the BW of these circuits is practically limited by the BW of the RF circuits preceding the diode. In fact, the latter is often capable

**Figure 13.** *DLVA simplified schematic.*

*UWB Technology - Circuits and Systems*

**3.2 Microwave measurements**

*3.2.1 RF power measurement*

in **Figure 13**.

analyse the role of the two circuits.

perform preliminary yet fast signal characterization.

ately feeding the RF signal to a diode—or to an array of diodes.

together with the waveforms at each section of the circuit.

*Simplified schematic of a diode detector and waveforms at each node of the circuit.*

In some applications, the incoming RF signal is unknown and therefore the system requires to quickly determine some of its key features, for example, carrier frequency and envelope power. Performing these measurements on UWB RF signals using only Digital HW can be complicated, or in some cases even impossible considering the power and frequency limitations of Silicon based technology. For this reason, microwave circuits have to be inserted in the receiving system to

The power measurement of an RF signal is commonly performed by appropri-

The diode has a well-known input-to-output square law characteristic, around the origin, when the diode is slightly forward biased. The signal coming out from the diode is composed by the even order harmonics of the incident RF signal and, most of all, its zero order term that is a DC voltage. The latter is captured by the following stages to perform the power measurement. There are several circuits that are capable of performing such functions, and their topology depends on the overall BW of the RF signal and also its dynamic range, i.e. the maximum to minimum power ratio to be analysed. A simplified block diagram of the diode detector is shown in **Figure 12**,

A detector logarithmic video amplifier (DLVA) consists of a diode detector followed by a logarithmic video amplifier (LVA), and its simplified schematic is given

As explained before, the diode (detector circuit) converts RF signal into a DC voltage which is then fed to an amplifier with a logarithmic transfer function. Let us

The diode's quadratic law provides the DC voltage component of the incoming RF signal. The level of such DC voltage can vary significantly: from tenths of μV to some Volts. The ratio between the maximum and minimum voltage could be unacceptable for the subsequent processing stages. A LVA is therefore inserted to compress the dynamic and make it more usable for the following stages. In fact, the log amplifier will greatly amplify the weak signals leaving the strong signal practically unaffected. It is worthwhile noting LVA's operating BW is a few tenths of MHz

**30**

**Figure 12.**

**Figure 14.** *Output voltage (with and without LVA) as function of input power expressed in dBm.*

of processing RF signals up to tenths of GHz. Diode operating at 50 GHz are not uncommon.

Another way to increase the system's dynamic range is to use a SDLVA topology, schematically depicted in **Figure 16**.

An SDLVA (Successive Detection Logarithmic Amplifier) is similar to a DLVA however, the SDLVA circuit is designed in such a way that it does not need a detector before the logarithmic video amplifier. The SDLVA uses multiple compressive stages of RF gain to emulate the exponential transfer function. The output of each stage is coupled into a linear detector. The detector operates over a narrower dynamic range, which means that more detectors are needed to cover the same dynamic range.

The principle of operation is the following: when the RF signal power is low, all amplifiers operate in a linear condition (i.e. the output power is proportional to the input power) and consequently the DC voltage provided by the diode detector is proportional too. As the power increases, the final stages begin to saturate and their output is capped to a saturation value. Therefore, any additional output voltage will be delivered only from the first stages until they saturate too, saturating as a consequence the Video output at its maximum values.

The typical dynamic range of each detector is approximately 10 dB, which are then summed in a single video amplifier so as to provide a single detected output. The overall dynamic range is 10 × *N* dB where *N* is the number of amplifiers. *N* = 7–8 represents an acceptable trade-off between high dynamic range and circuit feasibility.

#### **Figure 15.**

*Simplified schematic of an extended DYNAMIC range DLVA.*

**33**

**Figure 17.**

*IFM simplified schematic.*

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications*

Microwave circuits can be profitably employed to perform an estimation of the carrier frequency of the incoming RF signal. This is all but a simple task. Nowadays, the advance of digital components has made feasible digital frequency measurement up to 10 GHz and beyond. Even in this case, some form of microwave frontend is required to prepare the signal for digital sampling. Above tenths of GHz, and considering UWB signals, pure digital frequency estimation becomes unpractical

UWB microwave circuits that perform frequency estimation are referred to as Instantaneous Frequency Measurement (IFM), whose very simplified schematic

The RF input *vin*(*t*) = *A cos*(*ωt*), where *ω* = *2f* and *f* is the instantaneous frequency carrier, is hard-limit amplified and then split by a power divider in two equal amplitude signals, one of which is delayed with respect to the other through a delay line. Hard limitation consist in amplifying all incoming signal to a fixed power level—obviously within a feasible dynamic range. The role of the delay line is to out-phase the two signals coming from the power divider's outputs by a quantity

The higher the frequency the more the two signals applied to the mixer will be

*Vout*(*t*) = *K cos*(ΔΦ), (2)

ΔΦ = *2* ∗ Δ*L* ∗ *f*/*vp* (3)

where Δ*L* is the difference between the physical distance of the delay line and the direct pat; *f* is the carrier frequency, while *vp* the speed of the EM wave in the medium. Higher order mixing terms are eliminated by the low-pass filter (LPF). The BW of such circuit is limited by the BW the power divider and the mixer. As a consequence, the delay is synthesized to implement *π* shift (maximum out-phase)

Very often Vout is digitized by means of a N-BIT analog-to-digital converter (ADC).

This helps the subsequent stages since the information is provided digitally, but the information accuracy is limited by the number of BITs (typically not more than four) and quantization effects. To overcome this issue the scheme in **Figure 18** is applied.

we have that the valuable mixer's output will be a zero frequency product and therefore a DC voltage since *RF* and *LO* signals have, of course, the same carrier frequency. The amplitude of the DC voltage is proportional to cosine of the outphase between the two signals and therefore to the carrier frequency, trough the

Considering the relationship IF = |*m\*RF + n\*LO*| already introduced in Section 3.1.1,

*DOI: http://dx.doi.org/10.5772/intechopen.87095*

and microwave circuits have to be inserted.

diagram is reported in **Figure 17**.

proportional to the frequency carrier.

at the components maximum operating frequency.

out-phased.

relationship:

and

*3.2.2 RF frequency measurement*

**Figure 16.** *SDLVA simplified schematic.*

*UWB Circuits and Sub-Systems for Aerospace, Defence and Security Applications DOI: http://dx.doi.org/10.5772/intechopen.87095*
