**3. FSSs in antenna engineering**

the electric field should have a component parallel to the direction of the dipoles. It is well known that at the resonant frequency of the array, the latter performs as a fully metalized screen and the incident waves are fully reflected with a phase reversal. Moreover, at resonance, the current is in phase with the incident field, i.e., the impedance seen by the incident wave is purely real, since the capacitive and inductive parts cancel each other. Also, a maxi-

For periodic arrays in proximity to a ground plane, some differences emerge. Due to the ground plane, incident waves are fully reflected at all frequencies. However, in this type of structures, careful investigation reveals that two distinct resonant phenomena occur for a normally incident wave. By assuming a free-standing array in proximity to an all-metal ground plane illuminated by a normally incident wave, the array resonance can be defined at the frequency where the currents excited on the array are in phase with the incident wave. At this frequency, the incident wave is reflected from the periodic array with a phase reverse, as in the case of the free-standing array resonance. However, it can be found that there also occurs a Fabry-Perot type of resonance at the cavity formed between the ground plane and the array. The Fabry-Perot resonance occurs at frequencies different from the array resonance. This strong cavity-type resonance excites maximum currents on the elements (which in general are out of phase with the incident wave), and the incident wave is reflected with a zero

The presence of vias in a mushroom-type structure imposes an EBG at the same frequency range as the AMC property. In other words, the mushroom structure exhibits high surface impedance for both normally incident and surface waves at the same frequency band. Hence, at the same frequency, it reflects a normally incident plane wave with zero phase shift behaving, therefore, as an AMC and supports no surface waves behaving, therefore, as an EBG. It was demonstrated that the AMC operation is not directly related to the resonance of the FSS array. In fact, it is noticed that with varying array periodicity, the AMC band moves oppositely to the FSS resonance frequency. On the contrary, it is shown that the EBG frequency

It can be shown that the specular reflection coefficient for one array equals the transmission coefficient for its complementary array. This is a simple case of the general "Babinet's principle." Based on this observation, it is often expected that the investigation of one of the two cases is enough. However, this is in general not the case. First of all, the conducting screen must be a perfect conductor and "infinitely" thin, typically less than 1/1000 wavelength. If the screen is thick, the bandwidth of the dipole array will be larger while the bandwidth of the slot array will be smaller. Furthermore, if a thin layer of dielectric is added, the resonant frequency will be lowered somewhat for both the dipole and the slot arrays, but for a dielectric

thickness of the order of λ/4 or more, the two cases behave vastly different [3].

mum current magnitude is excited on the elements.

phase shift [2].

**2.2. AMC resonance cavity**

18 UWB Technology and its Applications

**2.3. Complementary arrays**

follows the trend of the FSS resonance [2].

FSSs' valuable features emphasized through the analysis above have encouraged their use in antenna engineering to improve antenna performance and create further properties that would not be achievable otherwise. They have been used, to widen the operating band of backing reflectors and to enhance the performance of broadband reconfigurable antennas, as superstrates and as reflectors.

#### **3.1. FSSs as reflectors and ground planes**

Extending the bandwidth of backing reflectors is among the rich utilizations of FSSs. In [4], an FSS is sandwiched between a tightly coupled array and a metallic plane, providing an additional reflecting plane for a higher frequency band. In this way, the metallic ground plane will operate at lower frequencies and the FSS will cover higher frequencies, which leads to an extended bandwidth, while the location of the metallic plane without an FSS would be suitable only for a relatively limited frequency range.

Placement of the metallic plane at a quarter wavelength distance from the antenna allows obtaining a good matching with only modest degradation of the achievable gain, but the improvement of the front-to-back ratio will come at the expense of the antenna bandwidth. The targeted application in [4] forces the integration of two frequency bands: one corresponding to the typical radar X-band, 8.50–10.50 GHz, and the other corresponding to a Tactical Common Data Link (TCDL) system, 14.40–15.35 GHz. Therefore, the used FSS was designed to be reflective at the higher frequency range and to be practically transparent for the lower band where the metallic ground plane is in charge of the reflection. More importantly, the FSS should separate the two frequency bands. Therefore, a special FSS has been chosen to serve the design purposes. The chosen element exhibits a good performance against angular variation and allows a packed lattice, with a further gain in angular independence.

In [5], a novel FSS design aimed at enhancing the performance of a broadband reconfigurable antenna has been presented. Designing FSSs' subject to phase requirements was also elaborated, revealing that some compromise, in the response magnitude, should be made to achieve the desired phase requirements. The broadband requirements also presented the need for noncommensurate FSS designs, contrary to previous FSSs that were primarily designed on the basis of the reflection coefficient amplitude and were intended for radome applications rather than substrates. When traditional broadband antennas such as log-periodic are printed on substrates, their bandwidth characteristics are altered, and one approach to regain the broadband behavior of the antenna element is to employ frequency-dependent substrates or ground planes (GPs). From here comes the suggestion of using FSSs to create substrates on which broadband antennas can be printed without affecting their broadband behavior. This can be achieved by using multiple layer FSSs as part of the substrate in a similar manner to that used for designing broadband microwave filters. Each screen is resonant at a given frequency and is placed at a distance, of a quarter of the wavelength at the screen's resonance frequency, away from the antenna's surface.

#### **3.2. FSSs as UWB reflectors**

In [6], a reflector consisting of two layers separated by an air gap of a width of 9.5 mm has been proposed. The upper layer was designed to be reflective over high frequencies of the UWB band and the second layer (lower layer) was used to reflect the transmitted waves through the upper layer. In other words, the upper layer operates as a band-stop filter for higher frequencies and a band-pass filter for lower frequencies, and the lower layer has an opposite operation.

from the antenna in the opposite direction to the FSS reflector. It is expected that the gain of the antenna in the presence of the FSS reflector will be maximum when the two wave compo-

The evaluation of the phase at the reference plane T is described by the following equations:

*φ<sup>T</sup>* = *φ<sup>R</sup>* + *φ<sup>S</sup>* (1)

Since the phase delay is frequency dependent and increases with frequency, the ideal FSS reflection phase should decrease with frequency at the same rate, which is associated with the slope of the curve (lower plot in **Figure 2**) that is controlled by the spacing between the

Several UWB antennas have been located above this reflector to verify its functionality. In [7], the antenna was located at 10 mm, which is approximately λ/4 at the center frequency of 6 GHz, above the reflector and a maximum gain of 9.5 dBi was achieved at 4.2 GHz. The gain variation, over the frequency band from 3 to 10 GHz, was ±1.5 dB. In [8], a rectangular slot antenna, fed through microstrip rectangular patch, was employed as a radiator. An optimized height of 10 mm was chosen to separate the antenna structure from the FSS. It was revealed that the optimized UWB FSS reflector has a very small effect on the impedance bandwidth of the radiator, which is 145% with reflector and 149% without it, while the gain was significantly improved due to the reflector. The average peak gain achieved by the slot antenna

alone is 5.7 dBi, while with the FSS reflector, the average peak gain is 10.9 dBi.

**Figure 3.** Multilayer FSS for constant gain UWB antenna [9].

2*f* \_\_\_

*<sup>C</sup>* L is the round-trip free-space propaga-

Ultra-Wideband FSS-Based Antennas http://dx.doi.org/10.5772/intechopen.79888 21

should be zero (or an integral multiple of *π*) at all frequencies.

nents are added in phase, giving rise to constructive interference.

) at reference plane R and *φ<sup>S</sup>* <sup>=</sup> <sup>2</sup> <sup>×</sup>

tion phase delay between the antenna and the top of the FSS reflector.

where *<sup>φ</sup> <sup>R</sup>* <sup>=</sup> *<sup>f</sup>*(*<sup>φ</sup>* <sup>1</sup>

antenna and reflector.

; *φ* <sup>2</sup>

Note that, for phase coherence, *φ<sup>T</sup>*

In order to gain insights into the operation mechanism of multilayer FSS/antenna combination, a schematic describing the operating principle is presented in **Figure 2**. Two reference planes have been defined, namely, plane R and plane T. To obtain a prescribed phase variation, the dual-layer FSS has been optimized over the ultra-wide band. The upper layer, which is responsible for providing reflection at higher frequencies, is formed by a set of cross dipoles and square loops. The reflection phase from upper layer is noted by *φ*<sup>1</sup> , *φ*<sup>2</sup> is the reflection phase provided by the lower layer, and *φ<sup>R</sup>* is the overall phase reflected from the multilayer FSS at the reference plane R.

When an antenna is placed at a distance L (mm) above the FSS, the wave radiated toward the FSS is reflected. This reflected radiation would be added to the direct outgoing wave radiated

**Figure 2.** Operation mechanism of the dual-layer UWB reflector in [6].

from the antenna in the opposite direction to the FSS reflector. It is expected that the gain of the antenna in the presence of the FSS reflector will be maximum when the two wave components are added in phase, giving rise to constructive interference.

**3.2. FSSs as UWB reflectors**

20 UWB Technology and its Applications

phase provided by the lower layer, and *φ<sup>R</sup>*

**Figure 2.** Operation mechanism of the dual-layer UWB reflector in [6].

FSS at the reference plane R.

In [6], a reflector consisting of two layers separated by an air gap of a width of 9.5 mm has been proposed. The upper layer was designed to be reflective over high frequencies of the UWB band and the second layer (lower layer) was used to reflect the transmitted waves through the upper layer. In other words, the upper layer operates as a band-stop filter for higher frequencies and a band-pass filter for lower frequencies, and the lower layer has an opposite operation. In order to gain insights into the operation mechanism of multilayer FSS/antenna combination, a schematic describing the operating principle is presented in **Figure 2**. Two reference planes have been defined, namely, plane R and plane T. To obtain a prescribed phase variation, the dual-layer FSS has been optimized over the ultra-wide band. The upper layer, which is responsible for providing reflection at higher frequencies, is formed by a set of cross dipoles

When an antenna is placed at a distance L (mm) above the FSS, the wave radiated toward the FSS is reflected. This reflected radiation would be added to the direct outgoing wave radiated

, *φ*<sup>2</sup> is the reflection

is the overall phase reflected from the multilayer

and square loops. The reflection phase from upper layer is noted by *φ*<sup>1</sup>

The evaluation of the phase at the reference plane T is described by the following equations:

$$
\phi\_r = \phi\_k + \phi\_s \tag{1}
$$

where *<sup>φ</sup> <sup>R</sup>* <sup>=</sup> *<sup>f</sup>*(*<sup>φ</sup>* <sup>1</sup> ; *φ* <sup>2</sup> ) at reference plane R and *φ<sup>S</sup>* <sup>=</sup> <sup>2</sup> <sup>×</sup> 2*f* \_\_\_ *<sup>C</sup>* L is the round-trip free-space propagation phase delay between the antenna and the top of the FSS reflector.

Note that, for phase coherence, *φ<sup>T</sup>* should be zero (or an integral multiple of *π*) at all frequencies.

Since the phase delay is frequency dependent and increases with frequency, the ideal FSS reflection phase should decrease with frequency at the same rate, which is associated with the slope of the curve (lower plot in **Figure 2**) that is controlled by the spacing between the antenna and reflector.

Several UWB antennas have been located above this reflector to verify its functionality. In [7], the antenna was located at 10 mm, which is approximately λ/4 at the center frequency of 6 GHz, above the reflector and a maximum gain of 9.5 dBi was achieved at 4.2 GHz. The gain variation, over the frequency band from 3 to 10 GHz, was ±1.5 dB. In [8], a rectangular slot antenna, fed through microstrip rectangular patch, was employed as a radiator. An optimized height of 10 mm was chosen to separate the antenna structure from the FSS. It was revealed that the optimized UWB FSS reflector has a very small effect on the impedance bandwidth of the radiator, which is 145% with reflector and 149% without it, while the gain was significantly improved due to the reflector. The average peak gain achieved by the slot antenna alone is 5.7 dBi, while with the FSS reflector, the average peak gain is 10.9 dBi.

**Figure 3.** Multilayer FSS for constant gain UWB antenna [9].

In [9], a four-layer FSS has been used to form a UWB reflector. The four layers are separated by dielectric layers of thickness of 1.58 mm, and a UWB microstrip slot antenna is placed at a distance of 19 mm above the FSS reflector. With this structure, an average peak gain of 9.3 dB was achieved with an oscillation of ±0.5 dBi, versus an average peak gain of 4 dBi and a variation of ±2 dB of the UWB microstrip slot antenna without the reflector. **Figure 3** illustrates the structure in [9].
