Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter

*Thaiyal Naayagi Ramasamy* 

## **Abstract**

The insulated-gate bipolar transistor (IGBT) offers low conduction loss and improved performance and, hence, is a potential candidate for high-current and high-voltage power electronic applications. This chapter presents the power loss estimation of IGBTs as employed in a high-voltage high-power dual active bridge (DAB) DC-DC converter. The mathematical models of the device currents are derived, and the power loss prediction is clearly explained using the mathematical models. There are many parameters to consider when selecting an appropriate power device for a given application. This chapter highlights the step-by-step procedure for selecting suitable IGBTs for a 20 kW, 540/125 V, 20 kHz DAB converter designed for aerospace energy storage systems. Experimental results are given to demonstrate the device performance at 540 V, 80 A operation of high-voltage IGBTs and 125 V, 300 A operation of low-voltage IGBTs and thus validate the selection procedure presented.

 **Keywords:** power loss prediction, HV-side IGBTs, LV-side IGBTs, zero voltage switching, zero current switching, dual active bridge, DC-DC converter

## **1. Introduction**

The dual active bridge (DAB) [1, 2] is a bidirectional DC-DC converter that has a small transformer between the two active bridges. This topology has received significant attention from researchers for high-power applications for decades. Compared with a single-phase DAB, a three-phase DAB of similar rating offers the benefits of smaller passive components and improved magnetic utilization [3]. Acronyms used in this chapter are listed in **Table 1**. A high-voltage SiC IGBT is used to achieve optimum design of a DAB converter based on the consideration of the device and transformer characteristics in [4]. However, the high *dv/dt* of the SiC IGBT switches [5–7] causes a huge spike and ringing in the currents during hard switching at high voltage levels.

An optimization algorithm for non-active power loss minimization in the DAB DC-DC converter is discussed in [8] to improve the converter efficiency. Ref. [9] presents an advanced control strategy to reduce DAB converter losses. A survey of material-level developments in the key components of PWM power converters and the potential for improving system power density using advanced components is presented in [10]. Ref. [11, 12] indicate that a reduction of the switching loss by tenfold


#### **Table 1.**

*List of acronyms and their abbreviations.* 

is achievable with the use of such advanced devices. Although wide bandgap devices have merits over Si devices, most have not been commercialized due to various reasons, such as the difficulty of mass producing large-diameter SiC wafers, the difficulty of controlling the impurity level of these devices, and high cost. Several parameters must be considered to select the correct power semiconductor device for any power electronic circuit. Device output characteristics, thermal characteristics, switching characteristics, blocking voltage, leakage current, and safe operating area (SOA) are all important factors with respect to reliability. It is important to note that at any point of time during device operation, the maximum ratings specified in the device datasheet should not be exceeded [13–15]. The gate charge characteristics and gate driver requirements of a power device and the associated driver circuit loss are also important factors to be considered when selecting an appropriate power device. The focus of this chapter is the estimation of the conduction power loss and the switching power loss. The key parameters and the device characteristics portrayed in the datasheets are discussed in [16].

The DAB DC-DC converter is shown in **Figure 1**. The converter has two H-bridge circuits interfaced via a high-frequency transformer. To correctly choose

**Figure 1.**  *Dual active bridge DC-DC converter.* 

#### *Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696*

the power device for the DAB, power loss calculation is performed for all power devices under the worst-case operating condition of the converter.

The performances of various metal-oxide-semiconductor field-effect transistors (MOSFETs) and IGBTs produced by commercial manufacturers are compared to determine the appropriate power semiconductor device for the high-voltage (HV) and low-voltage (LV) sides of the converter [17–21]. For mission-critical applications such as aerospace systems, voltage transients are inevitable and are taken into consideration when selecting the voltage ratings of devices: 1200 V IGBTs are chosen for the HV side, and 600 V IGBTs are chosen for the LV side [22]. In addition to the device voltage and current handling capabilities and the power density, the anti-parallel diode's on-state voltage and soft recovery characteristics should be taken into consideration. An analysis of MOSFET usage for the low-voltage side of the converter is presented in [23]. Due to the low voltage on the secondary side, the devices should be able to handle high currents. Hence, three or more MOSFETs should be connected in parallel, which makes the circuit very complex. Consequently, IGBT technology [24–26] has been chosen due to its high-current and high-voltage handling capabilities. The IGBT power modules have the benefit of reduced internal inductance and improved heat dissipation and are easy to connect. This chapter discusses the selection procedure of appropriate power devices for a high-power DAB DC-DC converter design. The mathematical model of the device current equations is presented. Researchers at an early stage of their research will find this information valuable as a reference for the design and development of converter prototypes.

 This chapter significantly extends the experimental results given in [16]. Specifically, the experimental results for high-voltage IGBTs with the snubber capacitor and for low-voltage IGBTs with and without the snubber capacitors are included. In addition, the experimental measurements of power losses of the HV- and LV-side IGBTs are presented. The remainder of this chapter is organized as follows. The power loss estimation of HV- IGBTs of the converter from various manufacturers is presented in Section 2. Section 3 discusses the power loss calculation of LV-side IGBTs and the loss comparison of IGBTs with similar ratings from various leading manufacturers. The experimental results are presented in Section 4, and the conclusions are given in Section 5.

## **2. Power loss estimation of HV-side devices of the DAB DC-DC converter**

 To select the correct device, a power loss calculation should be performed for the worst-case scenario of the converter for the chosen application. In this work, we consider the design of a DAB converter for an aerospace energy storage system whose HV side is connected to the DC link of an aircraft and its LV side is connected to an ultracapacitor-based energy storage system. The working range for the ultracapacitor is assumed to be 2:1. The worst-case operating condition of the converter is *VHV* = 540 V, *VLV* = 62.5 V, inductor RMS current *IRMS* = 427 A, peak inductor current *IP* = 640 A, average ultracapacitor current *I0* = 320 A, power throughput *P0* = 20 kW, and *d* = 0.5. The main component values of the converter are determined by applying the worst-case operating condition to the equations derived based on the mathematical model of the DAB converter, which are presented in [16, 27].

The loss calculation is performed for the devices provided by various leading manufacturers for a 20 kHz switching frequency. The theoretical waveforms of the converter and the device currents are illustrated in **Figure 2**. The terms and symbols used in the analysis are listed in **Table 2**. Piecewise linearity is assumed for the

#### **Figure 2.**

*Theoretical waveforms of devices on the HV side and LV side of the DAB DC-DC converter. (Device conduction intervals are marked in pink).* 

device current waveforms. Hence, the average current flow through the IGBT on the HV side of the converter during the on-time is given by

$$\begin{array}{l}\text{vertex during the on-time is given by} \\\\ I\_T = \frac{\frac{1}{2} \times I\_{L1} \times \left(\frac{dT\_S}{2} - t\_B\right) + \frac{1}{2} \times \left(I\_{L1} + I\_P\right) \times \left(\frac{T\_S}{2} - \frac{dT\_S}{2}\right)}{\frac{T\_S}{2} - t\_B} \\\end{array} \tag{1}$$

The IGBT has a constant voltage drop during the on-state. Hence, the conduction loss can be calculated using the average IGBT current and the duty cycle, which is given as

$$P\_{\text{Con}IT} = \; V\_{\text{CE(sat)}} \times I\_T \times d \tag{2}$$

The duty cycle is the ratio of the on-period of the transistor to the switching period. The datasheet specifies the forward voltage drop of the transistor (*VCE(sat)*) and the anti-parallel diode (*VF*) with respect to the main terminals of the modules, which includes the voltage drops across the terminals. Due to the high power densities of the devices, terminal losses cannot be neglected compared with the semiconductor losses. Hence, it is important to specify the voltage drop at the chip level and across the terminals (*rCE'*) separately. The voltage drop across the terminals is given by

*Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696* 

$$\mathbf{V}\_{\rm CE(sat)} = \mathbf{V}\_{\rm CEO} + \left(\mathbf{r}\_{\rm CE} \times I\_T\right) \tag{3}$$

The average diode current on the HV side of the converter during the on-state, as shown in **Figure 2**, is given as

$$I\_D = \frac{1}{2} \times I\_P \tag{4}$$

In Eq. (4), the diode conduction interval and the base interval are the same on the HV side; hence, the diode current is computed regardless of time. Assuming a constant voltage drop, the conduction losses of the diode are estimated as


**Table 2.** 

*List of symbols/terms used in analysis and their description.* 

$$P\_{CombD} = \; V\_F \times I\_D \times d \tag{5}$$

The DAB converter has four transistors and four antiparallel diodes on the HV side. Hence, the total conduction loss *PC* of the semiconductor devices on the HV side of the converter is given as

$$P\_{\mathcal{C}} = \text{ } \nwarrow \text{ } \nwarrow \text{ } \mathcal{P}\_{\text{Cond}T} + P\_{\text{Cond}D}\text{ }\text{ }\tag{6}$$

Switching losses are calculated from the turn-on (*EON*) and turn-off (*EOFF*) energy loss curves, which are usually given as a function of the collector current at the switching instants of an IGBT, as given in the manufacturer datasheet. In the DAB converter configuration under zero voltage switching (ZVS), the diode always turns off with zero current, thus eliminating the diode reverse recovery losses. Therefore, the power loss equations for the turn-on and turn-off instants are

$$P\_{ON} = E\_{ON} \times f \tag{7}$$

and

$$P\_{\rm OFF} = \; E\_{\rm OFF} \times f \tag{8}$$

 *PON* is the power loss during the turn-on instant, *POFF* is the power loss during the turn-off instant, and *f* is the switching frequency. In some manufacturers' datasheets, the energy loss curves for turn-on and turn-off may not be available. For those cases, it is necessary to consider the simplified voltage and current waveforms during the switching process. Switching losses are predominant during the risetime and fall-time periods. Hence, the loss equations are approximated during the turn-on and turn-off instants, which are given as

$$P\_{\rm ON} = \frac{1}{2} V\_{\rm CE} \times I\_{\rm C} \times t\_r \times f \tag{9}$$

and

$$P\_{\rm OFF} = \frac{1}{2} V\_{\rm CE} \times I\_C \times t\_f \times f \tag{10}$$

In DAB converter operation during ZVS, the anti-parallel diode current is always transferred to the IGBTs. Hence, the turn-on switching losses can be neglected. Therefore, the total switching losses (PSW) are approximated as

$$P\_{SW} = \pounds \times \left[ E\_{OFF} \times f \right] \tag{11}$$

The junction temperature of the devices is calculated as

$$T\_{jIGBT} = T\_s + \left(P\_T \times R\_{th\text{(c-s)}}\right) + \left(P\_{IGBT} \times R\_{th\text{(j-c)}}\right) \tag{12}$$

and

$$T\_{jDide} = T\_s + \left(P\_T \times R\_{th\text{(c-s)}}\right) + \left(P\_{Dide} \times R\_{th\text{(j-c)}}\right) \tag{13}$$

where *TS* is the heat sink temperature and *PT* is the total power loss of the IGBT module. The case to sink *Rth( −s)* and junction to case *Rth(j− )* thermal resistances can *Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696* 

**Figure 3.**  *HV-side IGBT and anti-parallel diode power loss comparison for various manufacturers.* 

be obtained from the device datasheet. The total power losses of the IGBTs and the antiparallel diode of the HV side of the DAB converter are compared for devices with ratings similar to 1200 V, 300 A from five different manufacturers:


The power losses of IGBTs and anti-parallel diodes are predicted for the worstcase operating condition and are depicted in **Figure 3**. Based on a comparison of the losses of all devices, the ultra-fast SKM300GB125D 1200 V, 300 A phase leg IGBT modules from Semikron [28] are selected for the HV side of the converter due to their low loss. As shown in **Figure 3**, the switching power losses of the IGBTs are significant and are a major contributor to the overall power loss due to the high switching frequency and high turn-off current.

## **3. Power loss estimation of LV-side devices of the DAB DC-DC converter**

The power loss of LV-side devices is predicted using a procedure similar to that discussed in Section 2. The main difference is that the shape of the LV-side device currents differs from those of the HV-side devices due to the phase shift introduced between the HV- and LV-side bridges as shown in **Figure 2**. The transistor and diode current equations during the on-state are

$$I\_T = \frac{\frac{1}{2} \times I\_{L1} \times \left(\frac{dT\_S}{2} - t\_B\right)}{\left(\frac{dT\_S}{2} - t\_B\right)}\tag{14}$$

and

$$I\_D = \frac{\frac{1}{2} \times (I\_P \times t\_B) + \frac{1}{2} \times (I\_{L1} + I\_P) \times \left(\frac{T\_S}{2} - \frac{dT\_S}{2}\right)}{\left(\frac{T\_S}{2} - \frac{dT\_S}{2} \star t\_B\right)}\tag{15}$$

Similar to the HV side, the on-state and switching power losses of the IGBTs and the antiparallel diode of the LV side of the DAB converter are compared for devices with ratings similar to 600 V, 700 A at Tcase = 25°C from four leading manufacturers:


**Figure 4** portrays the power loss comparison of an LV IGBT and antiparallel diode for each module. As shown in **Figure 4**, all IGBT modules exhibit similar performance. During the forward conduction mode (charging mode), the diodes perform the rectification function. Hence, the conduction losses of the diodes are predominant. When the power flow reverses, the loss values are exchanged between the diode and IGBT. The phase leg modules from Semikron have lower losses than the other devices. In addition, the Semikron module has a maximum junction temperature of 175°C, whereas the other modules have a maximum junction temperature of 150°C. Hence, the Semikron SKM600GB066D high-temperature phase leg IGBT modules [29], which have a rating of 600 V, 760 A, were chosen for the LV side of the converter.

**Figure 4.**  *LV-side IGBT and antiparallel diode power loss comparison from various manufacturers.* 

*Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696* 

## **4. Experimental results**

Experiments were performed for the HV- and LV-side devices of the converter using a suitable reactive load. A 39 μH air-core inductor was used for HV bridge testing by phase shifting the two half-bridge legs of the IGBTs, and the devices are subjected to maximum currents. Because the load was purely reactive, the current drawn from the source was used to meet the device conduction losses, switching losses, and the losses due to the passive components. A photograph of the DAB converter prototype is shown in **Figure 5**. Experimental results are shown in **Figures 6** and **7** for the HV-side IGBTs of the DAB converter for 540 V, 80 A operation with and without the snubber capacitor.

 A 47 nF snubber capacitor was used on the HV side to minimize the turn-off losses and improve the switching transient of the HV-side IGBTs. A direct mount snubber capacitor type was used with a reduced effective series resistance (ESR) and with a low effective series inductance (ESL). As evident from the waveforms shown in **Figures 6** and **7**, the diode begins conduction before the transistor, which ensures ZVS turn-on at peak current and ZVS/ZCS during turn-off. Similarly, the IGBT has ZVS/ZCS turn-on and ZVS turn-off at peak current.

The power loss measured during the experiments for the HV bridge converter at 16.6 kVA operation is plotted in **Figure 8**. Two cases are considered: in the first, a 47 nF

#### **Figure 6.**

*Experimental results for the HV-side converter. Vin = 540 V, L = 39 μH, VLrms = 363 V, ILrms = 65.6 A, fs = 20 kHz, IOFF = 80 A. Channel 1 (yellow)—gate pulse of transistor A2, 10 V/div. Channel 2 (pink)—gate pulse of transistor B2, 10 V/div. Channel 3 (blue)—IGBT A1 voltage, 350 V/div. Channel 4 (green)—IGBT A1 current, 50A/div. Time scale: 20 μs/div.* 

#### **Figure 7.**

*Experimental results for the HV-side converter with a 47 nF snubber. Vin = 540 V, Iin = 1.318 A, L = 39 μH, fs = 20 kHz, VLrms = 363 V, ILrms = 65.6 A, IOFF = 80 A, RG = 3 Ω, Cs = 47 nF. Time scale: 10 μs/div. Channel 1 (yellow)—gate pulse of transistor A2, 10 V/div. Channel 2 (pink)—gate pulse of transistor B2, 10 V/div. Channel 3 (blue)—IGBT A1 voltage, 200 V/div. Channel 4 (green)—IGBT A1 current, 50A/div.* 

snubber capacitor is connected across each of the four IGBTs of the HV-side bridge; in the second, no snubber capacitor is used. The power losses of the leading-phase and lagging-phase leg devices are displayed separately in **Figure 8** due to different conduction intervals of the devices. Power losses in the cables, busbar, and connections are neglected. Due to the ZCS turn-off of the diodes, their reverse recovery losses are neglected; due to ZVS/ZCS turn-on of the IGBTs, their turn-on losses are omitted. As shown in **Figure 8**, the air-core inductor copper losses are high although it has only 11 turns. Diode and IGBT conduction losses, however, remain approximately constant with or without the snubber capacitors. As evident in **Figure 8**, there is a considerable reduction in turn-off losses when the snubber capacitor is used.

*Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696* 

**Figure 8.** 

*Experimental power loss breakdown of the HV-side converter with and without the snubber capacitor. Vin = 540 V, L = 39 μH, fs = 20 kHz, IOFF = 80 A, VLrms = 363 V, ILrms = 65.6 A, RG = 3 Ω.* 

 The LV-side devices of the converter were tested, and the LV-side IGBTs were subjected to a peak current of 300 A at 125 V. The two legs of the LV-side bridge were connected through an air core inductor with a value of 4.17 μH. An external electrolytic capacitor bank of 19.8 mF capacitance (comprising three 6.6 mF capacitor banks) was added to the DC supply to smooth the input ripple current. **Figure 9** shows the current and voltage waveforms of the leading-leg IGBT and the driving signals of the IGBTs, D2 and C2. A CWT15 Rogowski current probe with a sensitivity of 2 mV/A was used

#### **Figure 9.**

*Experimental results for the LV-side converter without the snubber. Vin = 125 V, Iin = 10.51 A, L = 4.17 μH, VLrms = 106 V, ILrms = 202 A, fs = 20 kHz, IOFF = 300 A, RG = 2.5 Ω. Time scale: 20 μs/div. Channel 1 (yellow) gate pulse of transistor C2, 10 V/div. Channel 2 (pink)—gate pulse of transistor D2, 10 V/div. Channel 3 (blue)—IGBT C1 voltage, 100 V/div. Channel 4 (green)—IGBT C1 current, 100A/div.* 

#### **Figure 10.**

*Experimental waveforms of the LV-side converter with a 100 nF snubber. Vin = 125 V, Iin = 9.29 A, L = 4.17 μH, VLrms = 106 V, ILrms = 202 A, fs = 20 kHz, IOFF = 300 A, RG = 2.5 Ω. Time scale: 10 μs/div. Channel 1 (yellow) gate pulse of transistor C2, 10 V/div. Channel 2 (pink)—gate pulse of transistor D2, 10 V/div. Channel 3 (blue)—IGBT C1 voltage, 100 V/div. Channel 4 (green)—IGBT C1 current, 100A/div.* 

#### **Figure 11.**

*Experimental measurement of power loss on the LV-side converter with and without the snubber. Vin = 125 V, L = 4.17 μH, fs = 20 kHz, IOFF = 300 A, VLrms = 106 V, ILrms = 202 A, RG = 2.5 Ω.* 

 to measure the device currents. The measured current fall-time is 205 ns for the initial current fall time and 620 ns for the tail current fall duration. High-frequency ringing is observed in the device waveforms when the snubbers are introduced across the devices. This may be resonance due to parasitic inductances of the module (with values of 15–20 nH), the busbars, and the snubber capacitor connections.

The input power drawn from the source using 100 nF snubber capacitors is lower than those observed for other snubbers. Moreover, a significant reduction

### *Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696*

in *dv/dt* during the switching transients is observed. Hence, 100 nF snubbers are chosen for the LV-side IGBTs. **Figure 10** shows the experimental waveforms for the LV-side IGBT C1 with a 100 nF snubber capacitor. **Figure 11** illustrates the power loss breakdown for the LV-side bridge converter at 19.2 kVA operation with and without the snubber capacitors, as determined from the measurements. The IGBT switching losses are dominant. Power losses due to cables, busbars, and connections are not included. When 100 nF snubber capacitors are connected across the IGBTs, no significant reduction in switching losses occurs. The experimental power loss using the 100 nF snubber is 135 W, and *dv/dt* during the switching transient is reduced significantly, thereby minimizing the device stresses.

Comparisons of the theoretical and practical power losses of the HV- and LV-side devices are shown in **Figures 12** and **13**, respectively, for the leading leg devices to validate the mathematical models describing the device current equations. A close correlation between the calculated and experimental power loss values is observed, which demonstrates the effectiveness of the mathematical models presented.

#### **Figure 12.**

*Power loss comparison of HV-side devices—calculated (Calc) and experimental (Exp).* 

**Figure 13.** 

*Power loss comparison of LV-side devices—calculated (Calc) and experimental (Exp).* 

## **5. Conclusion**

Power loss analysis of IGBTs in a high-voltage high-power DAB DC-DC converter intended for use in an aerospace energy storage system was presented. The guidelines for selecting the appropriate IGBTs for a 20 kW, 540/125 V, 20 kHz DAB DC-DC converter prototype suitable for aerospace applications were given. The important parameters provided in the device datasheet for calculating the device power losses were discussed. Power loss analysis was performed for HV IGBTs corresponding to five leading manufacturers and LV IGBTs corresponding to four leading manufacturers based on the DAB converter prototype design. Apart from the guidelines given in this chapter, some IGBT manufacturers offer customer support for device loss estimation through a software package. Such software packages may be used for preliminary studies to estimate the device losses and the junction temperature for various operating conditions. In this work, experimental results were presented for 540 V, 80 A peak current operation on the HV-side IGBT, and 125 V, 300 A peak current on the LV-side IGBT. Using snubber capacitors on the HV side of the converter resulted in a nearly 45% reduction of IGBT switching losses. Introducing snubber capacitors on the LV-side devices created parasitic ringing, with no significant reduction of switching power losses. However, snubbers on the LV side did reduce the device stresses by limiting *dv/dt*. The experimental results demonstrate that the use of snubber capacitors across IGBTs reduces switching losses and device stresses and thus improves converter performance.

## **Acknowledgements**

The author thanks Rolls-Royce plc and the Engineering and Physical Sciences Research Council (EPSRC), UK, for the DHPA scholarship at the University of Manchester.

## **Author details**

Thaiyal Naayagi Ramasamy School of Electrical and Electronic Engineering, Newcastle University International Singapore, Singapore

\*Address all correspondence to: naayagi.ramasamy@ncl.ac.uk

© 2018 The Author(s). Licensee IntechOpen. This chapter is distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/ by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

*Power Device Loss Analysis of a High-Voltage High-Power Dual Active Bridge DC-DC Converter DOI: http://dx.doi.org/10.5772/intechopen.80696* 

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## Chapter 6

## Resonant Power Converters

Mohammed Salem and Khalid Yahya

## Abstract

Recently, DC/DC resonant converters have received much research interest as a result of the advancements in their applications. This increase in their industrial application has given rise to more efforts in enhancing the soft-switching, smooth waveforms, high-power density, and high efficiency features of the resonant converters. Their suitability to high frequency usage and capacity to minimize switching losses have endeared them to industrial applications compared to the hard-switching conventional converters. However, studies have continued to suggest improvements in certain areas of these converters, including high-power density, wide load variations, reliability, high efficiency, minimal number of components, and low cost. In this chapter, the resonant power converters (RPCs), their principles, and their classifications based on the DC-DC family of converters are presented. The recent advancements in the constructions, operational principles, advantages, and disadvantages were also reviewed. From the review of different topologies of the resonant DC-DC converters, it has been suggested that more studies are necessary to produce power circuits, which can address the drawbacks of the existing one.

Keywords: soft-switching, LLC resonant converter, control strategies, resonant power converters (RPCs),

### 1. Introduction

Several studies have been conducted on the switching mode DC-DC converters to ensure that they satisfy the most demanding criteria for power electronics application. The possibility of minimizing the switching and conduction losses in the switch-mode through increasing the switching frequency makes them more attractive. However, several switching topologies can attain a high-power transfer [1, 2] but the problem is the power switches (transistors or MOSFET), diodes, and energy storage passive elements (capacitors and inductors) contained in the structure of the power converters, which affects their efficiency. Efficient circuits have been developed for power converters along with the developments in materials, control systems, and devices technology. Such topologies can minimize the overall converter size and the switching losses, thereby, providing a higher efficiency [3–5]. The DC-DC converters can be categorized into three major groups (linear, hardswitching, and soft-switching mode resonant converters) based on their modes of operation as depicted in Figure 1 [6–8].

The advantages of the linear regulator technologies include low noise, fast response, simplicity, and excellent regulation. However, they have some disadvantages such as power dissipation in any working condition, which can result in low efficiency. The switching-mode topologies can be categorized based on the isolation

#### Figure 1.

The classification of the DC-DC converters.

#### Figure 2.

Current waveform and voltage waveform of hard-switching and soft-switching at both turn-on and turn-off transitions.

feature to galvanic isolation converter, which also called chopper, and non-isolated converter. Galvanic isolation is required in most applications during the process of converting the power from the utility grid (voltages ranging from 100 to 600 V) for safety reason [9–11]. Since a higher power transformer is required in these converters, the single-switch converter may not be a proper solution for higher power applications, and as such, the other DC-DC isolated converters with more than one switch, such as the push-pull and full or half-bridge converters, are more appropriate for such high-power applications. Hard-switching transitions in devices operating with switched-mode converters will result in high power dissipation and a gradual reduction of the efficiency of the converter; it can damage the switching devices as well. Snubbers are used to reduce the stress of the switches and solve this problem of power dissipation. Furthermore, the hard-switching converters have other disadvantages such as limited frequency, high EMI, high switching losses, large size, and heavy weight. Two more problems are encountered during the control of the transferred power; the first one is the generated noise during the switching process while the second one is the energy lost in the switches.

Regardless of these drawbacks, this study implemented the hard-switching transitions [12]. The voltage and the active switches' current were modified to overcome or minimize their effects [13]. These modified methods work by either forcing the voltage across the switch or current through the switch to zero, and such a transition technique can only be achieved with a soft-switching technique (Figure 2). The current waveform is forced to reduce to 0 in the zero current switching (ZCS) circuit while the voltage waveform is treated as such by the zero voltage switching (ZVS) circuit. Power circuits with these types of transition techniques are called soft-switching converters (SSCs) [14]. Meanwhile, there are

Figure 3. Categories of the SSCs.

several advantages of the SSCs over the linear mode regulators. The advantages include (i) the possibility of using a small ferrite transformer core due to the highswitching frequency, making operation in a wider DC input voltage range possible compared to the linear regulators; and (ii) it offers a higher efficiency. However, the complexity of the control circuit is associated with several drawbacks compared to the linear control circuit; and the power switching technique may increase the supply noise.

The SSCs have been significantly improved in the areas of switching losses, the EMIs, and device stresses, allowing the converters to perfectly work even at higher frequencies and a consequent reduction in the magnetic components. Generally, the SSC families are categorized into resonant transition converters (RTCs), resonant power converters (RPCs), quasi-resonant converters (QRCs), and multi-resonant converters (MRCs) based on their modes of operation, as shown in Figure 3.

## 2. Resonant power converters (RPCs)

Figure 4 shows the structure of the RPCs and each stage represents a specific job to be carried out. The controlled switching network (CSN) is powered by the DC source; it rapidly switches on and off depending on the working frequency to generate the output voltage or current, which feeds the next stage. The sinusoidal voltage and current signals are generated at a stage of the high-frequency resonant tank network (RTN), where there are two or more reactive components. This is to ensure a reduced electromagnetic interference and harmonic distortion [15]. Being deployed as an energy cushioning stage between the load and the CSN, a frequencyselective network can identify this stage. The impedances of both capacitance and inductance at resonance condition are equivalent and will produce the resonant

Figure 4. The structure of the RPC.

frequency. A rectifier network and a pass filter are then used for rectifying and filtering the output signal to generate the anticipated DC output voltage [16].

#### 2.1 Control switching network (CSN)

The full and half bridge are the commonest switching networks, whose usage is power-dependent. For high-power applications, the full bridge inverter is often used as opposed to the single ended or half bridge inverters, which can only supply the active switch with half of the input voltage. This indicates that full bridge inverters have a low rate of voltage switch, making them ideal for application in high input voltage conditions [17]. The RPCs and either a half or full-bridge inverter are often deployed together along with each of center-tapped or full-bridge rectifiers [18].

The CSN depicted in Figure 5 generates a square waveform voltage Vs(t) of the switching frequency fs (ɷ<sup>s</sup> = 2π fs) as represented in Eq. 1 by the Fourier series. Considering the response of the resonant tank, which is dominant to the basic constituent fs of the voltage waveform Vs(t), then, the infinitesimal response clearly demonstrated the harmonic frequencies nfs, n = 3, 5, 7, …. As a result, the power that correlates to the basic voltage waveform constituent Vs(t) is moved to the resonant tank as represented in Eq. (2). The basic constituent is a sinusoidal waveform with a peak amplitude of (4/π) times the DC source voltage. This basic constituent and the original waveform are in the same phase. However, when the S1 is on, there is a positive output sinusoidal switched current (t) but negative when S2 is off. This is due to the alternate working principle of the two switches, and its peak amplitude Is1 with phase equal to φs. The input current (DC) to the CSN can be computed by dividing the sinusoidal switched current with half the switching period [6, 16].

$$\mathbf{V\_s(t)} = \frac{4\mathbf{V\_g}}{\pi} \sum\_{\mathbf{n}=1,\ 3,\ 5,\ \dots} \frac{1}{\mathbf{n}} \sin\left(\mathbf{n} \mathbf{o\_s t}\right) \tag{1}$$

$$\mathbf{V\_{s1}(t)} = \frac{4\mathbf{V\_{g}}}{\pi} \sin\left(\text{nos}\,\mathbf{t}\right) \tag{2}$$

$$\mathbf{i}\_s(\mathbf{t}) = \mathbf{I}\_{s1} \sin \left(\alpha\_s \mathbf{t} - \mathbf{q}\_s\right) \tag{3}$$

$$\mathbf{I}\_{\rm in} = \frac{2}{\mathbf{T}\_{\rm s}} \int\_{0}^{\frac{\mathbf{T}\_{\rm s}}{2}} \mathbf{i}\_{\rm s}(\mathbf{t}) \mathbf{d}\_{\rm t} = \frac{2}{\pi} \mathbf{I}\_{\rm s1} \cos \left( \boldsymbol{\uprho}\_{\rm s} \right) \tag{4}$$

#### 2.2 Resonant tank network (RTN)

Resonant tank networks (RTNs) comprise of LC circuit (reactive elements) that stock oscillating energy with the frequency of circuit resonant. The LC circuit's resonance h attains the electromagnetic frequency useful in several applications, including the telecommunication technology. The tank can possibly be charged to a

Figure 5. The equivalent circuit of CSN.

#### Resonant Power Converters DOI: http://dx.doi.org/10.5772/intechopen.81629

certain resonant frequency through the adjustment of reactive element data. In addition, the essential phase of the resonant power converter is the resonant tank network. There are different kinds of RTN; it is mainly categorized into three parameters. Firstly, the resonant power converter can be sectioned through the connection technique used in tank element. The main common three resonant circuits include a series-parallel resonant converter (SPRC), a series resonant converter (SRC), and parallel resonant converter (PRC) [19]. The second factor lies in a quantity of the reactive elements (amount of transfer function order). However, the third one depends on the elements and multi-elements resonant tank [16]. Topographies of the three elements RTN (third order resonant tank) are controlled in overpowering the inadequacies in the two elements RTN. Most especially, the third element is put in the two elements RTN with a certain rumination to generate the three elements RTN. Thus, these can be taken as an integration of the advantages of mostly used two elements resonant converters PRC and SRC and enhance their inadequacies. The third order RTN contains 36 various tanks, the most common used tanks are LCC, LCL, and CLL [14, 20]. Multi-element resonant converters are RTN that contains four and beyond the number of elements. It should be noted that an increase in the number of reactive elements cause the network to be more complicated based on its size, analysis, and cost. For instance, Figure 6 illustrates the fourth order RTN, which is referred to as the LCLC tank system [21]; this topology includes the characteristics of two main famous three-element systems such as LLC and LCC, and then reflects on their setbacks.

## 2.3 Diode rectifier network with low pass filter (DR-LPF)

The RTN generates voltage waveforms and sinusoidal current based on the output voltage and resonant frequency, which are taken to be the input to the last phase DR-LPF is the pulse waveform. This implies that the purpose of utilizing DR-LPF is to filter and correct the AC waveform to achieve the entailed DC output waveform. In the previous studies on resonant power converter, the DR-LPS had been illustrated as a full-bridge or center-tapped rectifiers. While the center-tapped rectifier is inappropriate due to an elevated voltage stress from the diodes, the low pass filter had been investigated for the entire occurrences of inductance or capacitive [16, 22].

## 2.3.1 Diode rectifier with capacitive low pass filter (DR-CLPF)

In the DR-CLPF, the input voltage VR(t) is appraised as the square wave of a resonant frequency as illustrated in Figure 7. The input voltage VR(t) can be estimated based on the resonant tank filtering with its basic component VR1(t) as shown in Eqs. (6) and (7), respectively. In addition, the basic component and current are in the same phase, any drop in current to zero alters the basic compartment because of the variations in the conducting diodes.

Figure 6. An illustration of four-element RTN (LCLC).

#### Figure 7.

DR-CLPF containing a capacitive pass filter and its variables waveforms.

$$\dot{a}\_{\mathbb{R}}(t) = I\_{\mathbb{P}} \sin \left( a\_{\mathbb{s}} t - \mathbf{q}\_{\mathbb{s}} \right) \tag{5}$$

$$V\_R(t) = \frac{4V\_o}{\pi} \sum\_{n=1,\ 3,\ 5,\ \dots} \frac{1}{n} \sin\left(n\phi\_s t - \phi\_s\right) \tag{6}$$

$$V\_{R1}(t) = \frac{4V\_o}{\pi} \sin\left(\alpha\_\circ t - \varphi\_\circ\right) \tag{7}$$

$$I\_o = \frac{2}{T\_s} \int\_0^{\frac{T\_s}{2}} i\_R(t) d\_t = \frac{2}{\pi} I\_R \tag{8}$$

#### 2.3.2 Diode rectifier coupled with inductive low pass filter (DR-LLPF)

The diode rectifier is coupled with the input voltage VR(t) (sinusoidal waveform) and an inductive filter jacket. The current inputted is appraised as the square waveform iR(t) as illustrated in Figure 8.

Figure 8. DR-LLPF coupled with an inductive pass filter and its variables waveforms.

Resonant Power Converters DOI: http://dx.doi.org/10.5772/intechopen.81629

$$V\_R(t) = V\_\text{p} \sin\left(a\_\text{s}t - \text{q}\_\text{s}\right) \tag{9}$$

$$i\_R(t) = \frac{4I\_o}{\pi} \sum\_{n=1,\ 3,\ 5,\ \dots} \frac{1}{n} \sin\left(n\phi\_s t - \eta\_s\right) \tag{10}$$

$$i\_{R1}(t) = \frac{4I\_o}{\pi} \sin\left(a\_\circ t - \mathbf{q}\_\circ\right) \tag{11}$$

$$V\_o = \frac{2}{T\_s} \int\_0^{\frac{T\_s}{2}} V\_R(t)d\_t = \frac{2}{\pi} V\_R \tag{12}$$

#### 3. Properties of resonant power converters

#### 3.1 Parallel resonant converter (PRC)

The PRC is categorized into two elements tank converter. The resonant capacitor Cr needs to be parallel to the diode rectifier network DR and load. For the effective load resistance, Rac is much more enormous relative to resonant capacitor reactance Cr; this implies that the resonant current is unproportionable to the load. Moreover, in addition, the voltage over the resonant capacitor and parallel resistance Rac can be improved by declining the load [14]. Figure 9 shows the relationship between the load quality factor and PRC voltage gain depending on switching frequency based on Eq. (13). It can be observed that the high voltage gain is attained from the switching frequencies and light load conditions, which are nearly equivalent to resonant frequency fs = fr. Therefore, PRC can either step the output voltage down or up depending on the variation in the control switching system frequency. The voltage output can be adjusted with load states, whereas, the resonant current is restricted to resonant inductor data, this causes the PRC to be appropriate for open and short circuit applications [23].

$$\mathbf{M}\_{V} = \frac{1}{\sqrt{\left[\mathbf{1} - \left(\frac{f\_r}{f\_r}\right)^2\right]^2 + \left[\frac{f\_r}{f\_r}\left(\frac{1}{Q}\right)\right]^2}}\tag{13}$$

#### 3.2 LLC resonant converter

LLC resonant converter in the RTN comprises three reactive elements, whereby it is appraised as a conventional SRC and addition of inductor Lr equidistant to the

Figure 9. Voltage gain characteristic of PRC.

load that is referred to as parallel inductor Lrp. The parallel inductor can be replaced by using the magnetizing inductance when using a transformer [14, 16]. The topology of LLC generates two resonant frequencies: firstly, the series resonant frequency fr1 depending on the series elements Lr Cr and secondly, the parallel resonant frequency fr2 depending on the entire three tank elements (Lrp, Cr, and Lr) as illustrated in Eqs. (14) and (15), where fr1 > fr2.

$$f\_{r1} = \frac{1}{2\pi\sqrt{(L\_r + L\_{rp})C\_r}}\tag{14}$$

$$f\_{r2} = \frac{1}{2\pi\sqrt{L\_rC\_r}}\tag{15}$$

Figure 10 illustrates the LLC voltage gain depending on the unity inductance ration (AL = 1), switching frequency, and load quality factor as shown in the Eq. (16).

$$\mathbf{M}\_V = \frac{\mathbf{1}}{\sqrt{\left[\mathbf{1} - \left(\frac{f\_s}{f\_r}\right)^2\right]^2 + \left[\frac{f\_s}{f\_r}\left(\frac{1}{Q}\right)\right]^2}}\tag{16}$$

#### 3.3 LCC resonant converter

The topology of LCC in the RTN comprises three reactive elements, the capacitor Crp that is connected to the load in parallel represents the third element. Thus, the topology contains two resonant frequencies: firstly, the series resonant frequency fr1 depending on the series elements Lr Cr and secondly, the parallel resonant frequency fr2 depending on the entire three tank elements (Lrp, Cr, and Lr) as illustrated in as they are shown in Eqs. (17) and (18), where fr1 < fr2.

$$f\_{r1} = \frac{1}{2\pi\sqrt{L\_rC\_r}}\tag{17}$$

$$f\_{r2} = \frac{1}{2\pi\sqrt{L\_r\left(\frac{C\_rC\_p}{C\_r+C\_p}\right)}}\tag{18}$$

Figure 10. Voltage gain property of LLC converter.

Figure 11. Voltage gain property of LCC converter.

In the LCC converters, the proportion of resonant capacitors AC should be chosen prudently to be equal to the targeted peak gain. Figure 11 explains the LCC voltage gain as depending on the load quality parameter and switching frequency with single inductance ratio (AC = 1) as described in Eq. (19). It is seen that the light load voltage gain advances across the converter properties and parallel resonant frequency fr2 acts as a parallel resonant converter PRC.

$$\mathcal{M}\_V = \frac{1}{\sqrt{\left(\mathbf{1} + \mathbf{A}\_C\right)^2 \left[\mathbf{1} - \left(\frac{f\_s}{f\_n}\right)^2\right]^2 + \left[\frac{1}{Q}\left(\left(\frac{f\_s}{f\_n}\right) - \frac{A\_C}{A\_C}\frac{f\_r}{f\_s}\right)\right]^2}}\tag{19}$$

#### 3.4 LCLC resonant converter

Similar to LCC, the topology of LCLC in the RTN comprises four reactive elements as illustrated in Figure 6, where this topology homogenizes the characteristics of LLC and LCC. The topology structure comprises parallel resonant capacitor Crp, parallel resonant inductance Lrp, and a series elements Lr Cr, which implies that the topology contains two proportions whereby the inductance AL and capacitance AC must be appraised in the design. Moreover, the topology comprises three frequencies: two parallel frequencies frp1, frp2, and one series resonant frequency frs. Figure 12 reflected the relationship between the load quality factor, AC = AL = 1 and LCLC voltage gain depending on the switching frequency as shown in Eq. (20).

$$M\_V = \frac{1}{\sqrt{\left[\mathbf{1} + A\_C + A\_L - A\_C \left(\frac{f\_r}{f\_r}\right)^2 - A\_L \left(\frac{f\_r}{f\_r}\right)^2\right]^2 + \left[\frac{1}{Q}\left(\frac{f\_r}{f\_r} - \frac{f\_r}{f\_r}\right)\right]^2}}\tag{20}$$

### 4. Controlled strategies of resonant converters

The controlled strategies of the resonant converter is a bit disparate from the pulse width modulated (PWM) converters. A lot of parameters should be considered to attain a soft switching at a certain segment to fabricate the precise controller

Figure 12. Voltage gain property of LCLC converter.

that can achieve the desired results like load condition, energy storage elements, frequency range, and among others. There were several controlled topologies, which had been applied in the previous studies to manage the series resonant converters. For example, the pulse density modulation, voltage and current control, diode conduction control, and frequency control [24]. The full bridge resonant converter voltage had been regulated through the phase shift control; this phenomenon can be referred to as the switching signal primary control. In addition, to enhance the outcome of a control system, several improved techniques through adaptive controls had been reported [25], which include the passivity-based control and auto disturbance-rejection control (ADRC). A phase shift control had been used to control the current of resonant [26]. Because of this, the control outcome was increased in relative to the traditional PSRC control system. From the previous studies, the controlled techniques can be categorized into their implementation technique either through analog or digital. The digital controls are used because of their flexibility features in compact, programming, and light in comparison to analog controllers, they are also more resistant to inferences and noise. A three element DC-DC resonant converter type LLC has been discussed in this part, in order to compare the performance of frequency (duty-cycle) with variable frequency control, in terms of wide load variation.

Moreover, to ensure the expansion range of ZVS for entire inverter switches (S1–S4), and to improve the converter voltage gain. Then, the magnetizing inductance and resonant tank are used to generate a second resonant frequency <sup>1</sup> fr1 <sup>¼</sup> <sup>p</sup>ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi. <sup>ð</sup>LrþLmÞ:Cr

From Figure 14, the LLC resonant converter gain M is evaluated using voltage divider law by considering the load quality factor Q and transformer step-up ratio as shown in Eq. (21).

$$M = \frac{V\_o}{V\_{in}} = \frac{Z\_i}{Z\_o} = \frac{F^2 (A\_L - 1)}{n\sqrt{\left(\frac{f\_r^2}{f\_{r2}} - 1\right)^2 + \left(Q\left(F^3 - F\right)^2 (A\_L - 1)\right)^2}}\tag{21}$$

#### 4.1 Fixed frequency control

The purpose of using this technique is that the voltage output is being influenced by changing the duty-cycle to attain a targeted voltage output. However, the error

that exists between the constant desired voltage (reference voltage) Verr ¼ Vref � Vmeas and measured voltage is used for the PI controller. Thus, the controlled signal pertained to the PWM generator through the utilization of the switching frequency in generating a gate signal for the entire switches. From Eq. (21), the switching frequency that is equal to 42:5 kHz is the highest gain that can be obtained if the switching frequency and resonant frequency are closer. Therefore, a duty cycle is the only factor that measures the output voltage. Because the voltage ripple can vary by changing the load, then, the duty cycle will react to the variation of load variation relative to the estimated output voltage ripple.

## 4.2 Variable-frequency control

This control varies the switching frequency in adjusting the output voltage so as to reach the targeted load stage and save the output voltage of converter stable in any situation as illustrated in Figure 5. Moreover, the error that exists between the stable targeted voltage (reference voltage) Verr ¼ Vref � Vmeas and measured is used in the PI controller. Therefore, the controller increases or decreases the switching frequency based on the targeted output voltage if any variation occurs from the required output or error sign. Thus, a signal will be sent to the entire device switches. Depending on the magnetizing inductance and resonant impedance, the tank response needs to be saved inductively to appraise the attainment of the ZVS in the entire switches. However, the ranges of controlled switching frequency need to be restrained by higher and lower frequencies that affirm the fulfillment of ZVS (45 � 37:5 kHzÞ.

## 4.3 Simulation results

The model simulation of resonant converter is carried out and implemented by utilizing the MATLAB/SIMULINK software using the factors enlisted in Table 1; this is to verify the LLC series resonant DC-DC converter design circuit as shown in Figure 13 and control techniques analysis.

In this frequency control, the controlled signal is used for the PWM generator in generating the gate signals for the entire switches. Then, the switches S1 and S4 are enhanced concurrently and replace the S2 and S3 switches to generate the input voltage for the resonant tank VAB. Therefore, the resonant elements generate the voltage VCr and sinusoidal current iLr as illustrated in Figure 15.

Figure 16 illustrates an output voltage of the duty-cycle controller dynamic response, the load is stepped up and down within half load of ð200 ΩÞ and a full


Table 1. The parameters of the LLC converter.

Figure 13. A simplified illustration of full-bridge LLC resonant converter.

Figure 14.

The AC equivalent circuit between the rectifier and inverter.

#### Figure 15.

Simulation waveforms of the resonant tank input voltage VAB, resonant inductor current iL, resonant capacitor voltage Vcr, and transformer primary voltage Vpri:

#### Figure 16.

The dynamic response of the output voltage, output current, and controlled duty-cycle signal with respect to the load changes.

load of ð100 ΩÞ. It was observed that the output voltage ripple is enormous at full load condition in relative to the half load state that reflects a direct duty cycle changes within the entire load conditions. Although, the system generates a favorable outcome by controlling the output voltage equivalent to 300 V, nevertheless, the resonant tank parameters mislaying the resonant concept during the changes in load changes as illustrated in Figure 15. At simulation time t ¼ 0:1 s, the load varies the AC tank parameters hold-up by the 1:5e � 3 s, thereafter, the resonant parameters reproduce the targeted AC parameters depending on a value of the load, which can be appraised as a disadvantage of the duty-cycle control technique.

In the variable frequency control technique, the measured output voltage is used to detect frequency. Thereafter, the controlled signal is implemented on PWM to produce gate signals to the entire switches by considering the switching duration depending on the converter nature to generate enormous output voltage gain. Moreover, the variation in load is being tested and applied to affirm the controller dynamic responses. Figure 18 illustrates the parameter frequency controller dynamic response of the output voltage, and the load is stepped up and down in the

#### Figure 17.

Simulation waveforms of the resonant tank input voltage VAB, resonant inductor current iL, resonant capacitor voltage Vcr, and transformer primary voltage Vpri.

similar way of duty-cycle control. It can also be observed that in this control technique, the full load condition results in enormous output voltage ripple as compared to half load state, and this affirms the significant variation in frequency with the entire load condition. Moreover, the parameter frequency control gives a significant response to the tank AC parameters as illustrated in Figure 17. As the load varies, the AC parameters temporarily react by increasing or decreasing the voltage and resonant current values depending on the varied frequency and keeping a shape of the sinusoidal waveform.

## 5. Applications of resonant power converters

Based on the sufficient demerit of RPCs as earlier stated in the above segments, they have uncommon application in modern industries. The summary of the noticeable implementations is discussed in this segment. The main areas of RPCs application are household applications like induction cookers, portable power supplies, network connection of renewable energy mains, and hybrid and electric vehicles. In a case of the portable power supply, requirements of the converter

#### Figure 18.

The dynamic response of output voltage, output current, and controlled frequency signal as the function of load change from full to half load.

include a low price tag, light-weight and small size, high efficiency, high reliability, and low electromagnetic interference (EMI). Soft switching is the way of ensuring higher efficiency; it can be implemented by utilizing RPCs. Based on the area of application, the topology can be chosen to ensure maximum efficiency, ideal cost, and size. For example, the supply of power to an electron beam welding compartment uses a full bridge LLC resonant converter [16]. The soft switching technique and topology solves the problem associated with power utilization within the filament supply by staying away from the inverter heating challenge and ensures higher efficiencies. RPCs are used in the electrostatic precipitator. This is a highpower appliance industrially utilized for removing smoke and dust from a flowing gas. The series-parallel RPC coupled with phase control suggested by [27] is negligible in size; it gives a faster temporary response and possesses a higher efficiency as compared to the traditional line frequency power supplies. RPCs are known for charging hybrid vehicles whereby the batteries need to be charged either by wireless or wired. Due to being smaller and demonstrating higher effectiveness, the charging of EVs through wire [28] and plug-in EVs [29] RPCs are used. For example, the high-performance LLC converters are suggested in the two-stage smart battery chargers. The converter evacuates the low and frequencies ripples from the current output and increases the life of a battery without the size increment of the charger. Previous studies had proposed several wired charging topologies. Apart from wired chargers, wireless power transfer (WPT) has been a modern charging process used in the hybrid and electric vehicles. The accessible WPT technologies include electromagnetic, magnetic, and electric power transfer. Among all of these, the magnetic coupling method utilizes RPCs, which portray higher power

transmission and higher efficiency within a closer distance. Application of RPC in WPT for hybrid and electric vehicles had been reported [30].

The grid integration of renewable energy mains such as fuel cell, wind, and solar PV needs converters of minimal current ripple and higher efficiency. DC-DC converters are the main prerequisite in processing power from renewable energy mains. Out of several options, RPCs can be the main competitor because of its low EMI, higher efficiency, robustness, and low output ripple current. RPCs are used in FC networks [31, 32], PVs [33], grid connection, and electrolyser interfaces [34]. Moreover, RPCs is applied to home induction cookers. The resonant inverter is the induction cooker key element; it produces an AC current that heats up the inductorvessel compartment. The resonant inverters utilize in induction cooker are multilevel, half bridge, single-switch, and full bridge inverters [35].

## 6. Conclusion

This chapter has vividly explained the resonant power converters, which include the effective generation of RPCs as a high-switching converter that can serve as a solution to electromagnetic interference (EMI) and switching losses difficulties that surface by using the PWM converters. Furthermore, the resonant converter classifications depending on several points of views as well as controlled techniques (either varied or fixed frequency techniques) were elucidated. The controlled techniques and several application areas of resonant converters have been explained. The variable and fixed frequency controls are used to cross-check the LLC converter output voltage. The load is changed when using 50% of the full load in the entire control techniques; the obtained results affirmed the significant stable response of both controllers with little overshoot voltage. Nevertheless, the variable frequency control gives a significant outcome based on the resonant tank waveforms in relative to the stable frequency control when changing the load.

## Author details

Mohammed Salem<sup>1</sup> \* and Khalid Yahya<sup>2</sup>

1 School of Electrical and Electronic Engineering, Universiti Sains Malaysia, Nibong Tebal, Malaysia

2 Kocaeli University, İzmit, Kocaeli, Turkey

\*Address all correspondence to: salemm@usm.my

© 2019 The Author(s). Licensee IntechOpen. This chapteris distributed underthe terms oftheCreative Commons Attribution License (http://creativecommons.org/licenses/ by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

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Section 4

Electric Power Conversion

Applications

Section 4
