5. RF MEMS-based reconfigurable component design

cb ¼

The bridge model can be presented by the next ABCD matrix:

3 5 ¼ 2 4

tools. The scattering parameters can be expressed by the following form:

A þ B=Z<sup>0</sup> � CZ<sup>0</sup> � D A þ B=Z<sup>0</sup> þ CZ<sup>0</sup> þ D

2 A þ B=Z<sup>0</sup> þ CZ<sup>0</sup> þ D

> Zc <sup>2</sup> � <sup>Z</sup><sup>2</sup> 0 � � <sup>þ</sup> 2RbZc

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �<sup>2</sup>

> 0 @

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �

> 0 @

The simulation results of the capacitive RF MEMS switch at the two states OFF (the bridge at downstate) and ON (bridge position g = 3 μm) are shown, respectively, in Figures 7 and 8.

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi ð Þ 2RbZc <sup>2</sup> <sup>þ</sup> ð Þ <sup>2</sup>ZcXb <sup>2</sup>

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi

� tg �<sup>1</sup> 2Xbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc

q

2RbZc � �

2RbZc � �

<sup>S</sup><sup>12</sup> <sup>¼</sup> <sup>S</sup><sup>21</sup> <sup>¼</sup> <sup>∣</sup>S12∣e<sup>j</sup>Φ<sup>12</sup> <sup>¼</sup> 2RbZc <sup>þ</sup> j2ZcXb

q

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �<sup>2</sup>

Ab Bb Cb Db � �

106 MEMS Sensors - Design and Application

4.2. The scattering parameters model

S<sup>11</sup> S<sup>12</sup> S<sup>21</sup> S<sup>22</sup> � �

¼>

¼>

8

>>>>>>>>>><

>>>>>>>>>>:

8 >>>>>>>>><

>>>>>>>>>:

∣S11∣ ¼

∣S12∣ ¼

¼

<sup>S</sup><sup>11</sup> <sup>¼</sup> <sup>S</sup><sup>22</sup> <sup>¼</sup> <sup>∣</sup>S11∣e<sup>j</sup>Φ<sup>11</sup> <sup>¼</sup> Zc

r

8 >><

>>:

<sup>Φ</sup><sup>12</sup> <sup>¼</sup> tg �<sup>1</sup> <sup>2</sup>ZcXb

r

<sup>Φ</sup><sup>12</sup> <sup>¼</sup> tg �<sup>1</sup> <sup>2</sup>ZcXb

¼

2 4

cdown <sup>¼</sup> <sup>ε</sup>0εrA th

(20)

(22)

(23)

(24)

1 0

1

3

5 (21)

1 Rb þ j Lð Þ <sup>b</sup>ω þ 1=Cbω

2ð Þ AD � BC A þ B=Z<sup>0</sup> þ CZ<sup>0</sup> þ D

�A þ B=Z<sup>0</sup> � CZ<sup>0</sup> þ D A þ B=Z<sup>0</sup> þ CZ<sup>0</sup> þ D

þ j2Xbð Þ Z<sup>0</sup> þ Zc

þ j2Xbð Þ Z<sup>0</sup> þ Zc

<sup>þ</sup> ð Þ 2Xbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup>

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �

9 >>=

>>;

1 A

> 1 A

<sup>þ</sup> ð Þ 2Xbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup>

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �

cup <sup>¼</sup> <sup>ε</sup>0<sup>A</sup> g<sup>0</sup> þ th εr

1 0 1 Rb þ jXb

The standard output of simulation tools is the S-parameters. In this way, we need to transform the previous developed model to present it in a convenient way to allow comparison with such

where the reflection coefficient and its phase are given by Eq. (23) and the insertion loss and its phase are given by Eq. (24). The scattering parameters are written in the following form:

> <sup>2</sup> � <sup>Z</sup><sup>2</sup> 0

ð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc <sup>2</sup> <sup>þ</sup> 2Rbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc � �

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi

� �<sup>2</sup> <sup>þ</sup> ð Þ <sup>2</sup>ZcXb <sup>2</sup>

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi

� tg �<sup>1</sup> 2Xbð Þ <sup>Z</sup><sup>0</sup> <sup>þ</sup> Zc

� � <sup>þ</sup> 2RbZcÞ þ j2ZcXb

1

3 5 ¼ 2 4

This section will present the use of RF MEMS switches to obtain different tunable RF devices, namely, a phase shifter, a resonator, and a tunable antenna.

#### 5.1. Reconfigurable phase shifter at 18 GHz based on RF MEMS

The reflection-type phase shifter as shown in Figure 9 is constituted with hybrid coupler and RF MEMS capacitive switches (i.e., metal-dielectric-metal) [20]. The first RF MEMS is connected between a through-port and ground. The second is linked between the coupled port and ground.

In this design, the tunability is achieved by the use of a capacitive RF MEMS switch acting as a reflection load. The capacitor value, which is controlled by a DC voltage, operates from downstate to upstate. This variable capacitance is used to tune the variable phase shifter.

The reflection-type phase shifter using switch RF MEMS capacitive was implemented in ADS simulation software. The reflection coefficient Γ is given by Eq. (25), where (Xb ¼ Lbω� ð Þ 1=cbω ).

$$\Gamma = |\Gamma| \mathbf{e}^{j\phi\_{21}} = \frac{\mathbf{Z\_s} - \mathbf{Z\_0}}{\mathbf{Z\_s} + \mathbf{Z\_0}} = \frac{\left[\mathbf{Z\_c^2} + 2\mathbf{Z\_c}\mathbf{R\_b} - \mathbf{Z\_c}\mathbf{Z\_0} - \mathbf{Z\_0}\mathbf{R\_b}\right] + j\mathbf{X\_b}[2\mathbf{Z\_c} - \mathbf{Z\_0}]}{\left[\mathbf{Z\_c^2} + 2\mathbf{Z\_c}\mathbf{R\_b} + \mathbf{Z\_c}\mathbf{Z\_0} + \mathbf{Z\_0}\mathbf{R\_b}\right] + j\mathbf{X\_b}[2\mathbf{Z\_c} + \mathbf{Z\_0}]} \tag{25}$$

Figure 10 shows the RF phase shifter performance around 18 GHz, in terms of the return and insertion loss, and the phase shift dependency on the applied voltage. Despite not fully linear, it is possible to observe an almost linear characteristic of the phase shifter in different frequency ranges.

Figure 9. Simulation results of the scattering parameters versus frequency at ON state.

effect is due to the presence of two meander inductors which are integrated in line with the RF waveguide. The combination of these two effects leads to a resonant phenomenon, introducing different resonant frequencies. If the applied voltage Vp is equal to 0 V, the bridge is in the UP state; therefore, the device is at a normally ON state. Moreover, the spacing g between the membrane bridge and the RF line affects the resonance frequency. The spacing g among bridge

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The proposed tunable resonator was simulated with HFSS and CST-MWS tools. Figure 12 presents the scattering parameters for different bridge positions, in order to achieve a tunabil-

Figure 12a and b shows, respectively, the return loss (S11) and the insertion loss (S12) for g = 2, 2.5, and 3 μm. It is possible to observe that controlling the bridge position level allows to

The S12 parameter presents almost constant value equal to 1 dB for all simulated spacing g factor when the S11 parameter is down to 10 dB. There is a good correspondence between the

Table 4 summarizes the spacing g factor versus the applied voltage. Moreover, the resonance frequency and the frequency range in different states of the bridge are shown. The RF MEMS is

> Resonance frequency (GHz) Frequency range (GHz) HFSS CST HFSS CST

and CPW line varies between g = 2 μm at OFF state and g = 3 μm at ON state.

ity on the frequency band between 10 GHz and 40 GHz.

obtain three resonant frequencies: 21.9, 24, and 25.1 GHz.

simulation results on HFSS and CST-MWS simulators.

Figure 12. Scattering parameters at g = 2, 2.5, and 3 μm: (a) return loss (b) insertion loss.

2 25 V 21.9 21 15.6–25.7 10–26.1 2.5 19 V 24 23.1 17.8–27.6 14.4–27.8 3 0 V 25.1 24.6 19.5–29 16.8–29

Space g (μm) Applied voltage (V) Cover band

Table 4. Simulation results of the proposed resonator.

Figure 10. Simulation results of the phase shifter versus the applied voltage at 18 GHz: (a) scattering parameters and (b) phase.

#### 5.2. MEMS-based reconfigurable resonator

There is an important claim that reconfigurable radio-frequency components on a single chip with high performances and multiband characteristics may be a solution for wireless communication [21, 22]. In this study, an improvement of the capacitive RF MEMS structure is proposed in order to obtain a reconfigurable resonator. Figure 11 shows the suggested RF MEMS resonator structure [23].

The tunable RF MEMS characteristic was designed based on capacitive and inductive effects. The capacitive effect is due to the space between the bridge and the RF line, while the inductive

Figure 11. Proposed resonator RF MEMS.

effect is due to the presence of two meander inductors which are integrated in line with the RF waveguide. The combination of these two effects leads to a resonant phenomenon, introducing different resonant frequencies. If the applied voltage Vp is equal to 0 V, the bridge is in the UP state; therefore, the device is at a normally ON state. Moreover, the spacing g between the membrane bridge and the RF line affects the resonance frequency. The spacing g among bridge and CPW line varies between g = 2 μm at OFF state and g = 3 μm at ON state.

The proposed tunable resonator was simulated with HFSS and CST-MWS tools. Figure 12 presents the scattering parameters for different bridge positions, in order to achieve a tunability on the frequency band between 10 GHz and 40 GHz.

Figure 12a and b shows, respectively, the return loss (S11) and the insertion loss (S12) for g = 2, 2.5, and 3 μm. It is possible to observe that controlling the bridge position level allows to obtain three resonant frequencies: 21.9, 24, and 25.1 GHz.

The S12 parameter presents almost constant value equal to 1 dB for all simulated spacing g factor when the S11 parameter is down to 10 dB. There is a good correspondence between the simulation results on HFSS and CST-MWS simulators.

Table 4 summarizes the spacing g factor versus the applied voltage. Moreover, the resonance frequency and the frequency range in different states of the bridge are shown. The RF MEMS is

Figure 12. Scattering parameters at g = 2, 2.5, and 3 μm: (a) return loss (b) insertion loss.


Table 4. Simulation results of the proposed resonator.

5.2. MEMS-based reconfigurable resonator

MEMS resonator structure [23].

108 MEMS Sensors - Design and Application

Figure 11. Proposed resonator RF MEMS.

phase.

There is an important claim that reconfigurable radio-frequency components on a single chip with high performances and multiband characteristics may be a solution for wireless communication [21, 22]. In this study, an improvement of the capacitive RF MEMS structure is proposed in order to obtain a reconfigurable resonator. Figure 11 shows the suggested RF

Figure 10. Simulation results of the phase shifter versus the applied voltage at 18 GHz: (a) scattering parameters and (b)

The tunable RF MEMS characteristic was designed based on capacitive and inductive effects. The capacitive effect is due to the space between the bridge and the RF line, while the inductive normally ON component, i.e., at g = 3 μm, the applied voltage equal to 0 V. This table presents a comparison study of the simulation results of the resonator between HFSS and CST-MWS. The proposed resonator covers three bands.

the RF MEMS resonator. The antenna considered was a modified patch antenna with a printed

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Figure 14 shows the reflection coefficient simulation results. The resonant frequencies can be observed at three states of the bridge. For g = 2 μm, the device has a single resonant frequency of 26.3 GHz, and the return loss will be 15.1 dB; for g = 2.5 μm, it shows two resonant frequencies: first at 27 GHz with a return loss of 23 dB and, second at 29.8 GHz with a return loss of 18 dB; and for g = 3 μm with also two resonant frequencies: 27.5 and 30.6 GHz, with

Figure 15 shows the radiation pattern for different resonance frequencies at three different antenna configuration states, considering phi = 90. Firstly, the three states bridge given three resonance frequencies and the main lobe at teta = 310. Secondly, only for g = 2.5 and g = 3 μm

Table 5 summarizes the simulation results of the reconfigurable antenna in terms of the spacing g factor versus the applied voltage, the resonant frequencies, the return loss, bandwidths, and

Figure 15. Realized gain of the reconfigurable antenna at different states: (a) at 2 μm, (b) at 2.5 μm, and (c) at 3 μm.

inverted U-shaped ring resonator.

gains.

return loss of 19.84 dB and 26.62 dB, respectively.

given the resonance frequency and the main lobe at teta = 0.

#### 5.3. Reconfigurable antenna based on a RF MEMS resonator

Figure 13 presents the structure of the proposed reconfigurable CPW antenna [24], which is based on the integration of a resonating RF MEMS device with the CPW antenna on the same substrate. The reconfigurability of this antenna depends on the load provided by the state of

Figure 13. Reconfigurable antenna based on RF MEMS.

Figure 14. Return loss results versus frequencies.

the RF MEMS resonator. The antenna considered was a modified patch antenna with a printed inverted U-shaped ring resonator.

normally ON component, i.e., at g = 3 μm, the applied voltage equal to 0 V. This table presents a comparison study of the simulation results of the resonator between HFSS and CST-MWS.

Figure 13 presents the structure of the proposed reconfigurable CPW antenna [24], which is based on the integration of a resonating RF MEMS device with the CPW antenna on the same substrate. The reconfigurability of this antenna depends on the load provided by the state of

The proposed resonator covers three bands.

110 MEMS Sensors - Design and Application

Figure 13. Reconfigurable antenna based on RF MEMS.

Figure 14. Return loss results versus frequencies.

5.3. Reconfigurable antenna based on a RF MEMS resonator

Figure 14 shows the reflection coefficient simulation results. The resonant frequencies can be observed at three states of the bridge. For g = 2 μm, the device has a single resonant frequency of 26.3 GHz, and the return loss will be 15.1 dB; for g = 2.5 μm, it shows two resonant frequencies: first at 27 GHz with a return loss of 23 dB and, second at 29.8 GHz with a return loss of 18 dB; and for g = 3 μm with also two resonant frequencies: 27.5 and 30.6 GHz, with return loss of 19.84 dB and 26.62 dB, respectively.

Figure 15 shows the radiation pattern for different resonance frequencies at three different antenna configuration states, considering phi = 90. Firstly, the three states bridge given three resonance frequencies and the main lobe at teta = 310. Secondly, only for g = 2.5 and g = 3 μm given the resonance frequency and the main lobe at teta = 0.

Table 5 summarizes the simulation results of the reconfigurable antenna in terms of the spacing g factor versus the applied voltage, the resonant frequencies, the return loss, bandwidths, and gains.

Figure 15. Realized gain of the reconfigurable antenna at different states: (a) at 2 μm, (b) at 2.5 μm, and (c) at 3 μm.


Acknowledgements

Temáticas.

Author details

Bassem Jmai<sup>1</sup>

Tunis, Tunisia

References

9781848825062

This work was supported by the Laboratory of Circuit and Electronic System in High Frequency of University of Tunis El Manar and Research Center for Microelectromechanical Systems (CMEMS) of the University of Minho Braga-Portugal. Foundation for Science and Technology (FCT) project PTDC/EEI-TEL/5250/2014, by FEDER funds through POCI-01-145-FEDER-16695 and Projecto 3599-Promover a Produção Científica e Desenvolvimento Tecnológico e a Constituição de Redes

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\*, Adnen Rajhi2,3, Paulo Mendes4 and Ali Gharsallah<sup>1</sup>

1 Department of Physics, FST, Unit of Research in High Frequency Electronic Circuit and

4 Department of Industrial Electronics, Microelectromechanical Systems Research Center,

[1] Gevorgian S. Ferroelectrics in Microwave Devices, Circuits and Systems: Physics, Modeling, Fabrication and Measurements. Springer Science & Business Media; 2009. ISBN:

[2] Sharma M, Kuanr BK, Sharma M, Basu A. Tunable coplanar waveguide microwave devices on ferromagnetic nanowires. International Journal of Materials, Mechanics and Manufactur-

[3] El Cafsi MA, Nedil M, Osman L, Gharsallah A. The design of a 360-switched-beam-base station antenna. In: Antenna Arrays and Beam-formation. Rijeka: InTech; May 2017. pp.

[4] Mabrouki M, Jmai B, Ghyoula R, Gharsallah A. Miniaturisation of a 2-bits reflection phase shifter for phased array antenna based on experimental realisation. International Journal of Advanced Computer Science and Applications. May 2017;8(5):438-454. DOI: 10.14569/

[5] Rijks TG, Steeneken PG, Beek JTM, Ulenaers MJE, Jourdain A, Tilmans HAC, Coster JD, Puers R. Microelectromechanical tunable capacitors for reconfigurable RF architectures.

ing. February 2014;2(1):9-13. DOI: 10.7763/IJMMM.2014.V2.88

2 Department Electrical Engineering, National School of Engineering Carthage, Tunis, Tunisia 3 Laboratory of Physics Soft Materials and EM Modelisation, FST, University Tunis El Manar,

\*Address all correspondence to: bassem.jmaiesti@gmail.com

System, University Tunis El Manar, Tunis, Tunisia

University of Minho, Guimarães, Portugal

13-30. ISBN: 978-953-51-3146-5

IJACSA.2017.080553

Table 5. The reconfigurable antenna results.
