3. Multi-port (six-port) circuit-based front-end receivers

The six-port (multi-port) quadrature down-conversion is an innovative approach in millimeter-wave technology. A comprehensive theory, validated by various simulations and measurements of 60 GHz (V-band) direct conversion receivers, has been presented in literature over the recent years [11, 12].

The block diagram in Figure 1 highlights the operation principle of a front-end receiver based on a six-port circuit to demodulate RF signals. This structure is composed of three 90 hybrid

Figure 1. Block diagram of a six-port circuit-based front-end receiver.

couplers and a Wilkinson power divider. The proposed architecture makes possible to obtain, from output power measurements, the phase difference and the amplitude ratio between an unknown signal from the antenna (a6) and the reference signal coming from a local oscillator (a5).

The output signals bi can be expressed as a function of the signals ai and the S-parameter Sij by the following linear relationship:

$$b\_i = \sum\_{j=1}^{6} S\_{i\bar{\eta}} a\_{\bar{\eta}} \qquad \text{i} = 1, \ldots, 6 \tag{1}$$

Using Eq. (4), we may obtain the formulas of the four waveforms, b1, b2, b3, and b4, as a

Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation

In order to simplify calculations, we assume that the RF signals resulting from the antenna a<sup>6</sup> and the local oscillator (LO) a<sup>5</sup> have an amplitude ratio α, a phase difference Δφ (t) = φ6(t) � φ5, and a frequency difference Δω = ω � ω<sup>0</sup> (ω = 2πf). Therefore, these signals can be expressed by

<sup>j</sup>ð Þ <sup>ω</sup>0∙tþφ<sup>5</sup> (6)

h i (8)

h i (9)

<sup>j</sup>ð Þ <sup>Δ</sup>ω∙tþΔφð Þ<sup>t</sup> h i (10)

h i (11)

<sup>i</sup> , i ¼ 1, …, 4 (12)

<sup>4</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup><sup>2</sup> � <sup>2</sup>∙α<sup>∙</sup> cos ½ � <sup>Δ</sup>ω∙<sup>t</sup> <sup>þ</sup> <sup>Δ</sup>φð Þ<sup>t</sup> � � (13)

jð Þ Δω∙tþΔφð Þþt π

<sup>j</sup> <sup>Δ</sup>ω∙tþΔφð Þþ<sup>t</sup> <sup>π</sup> ð Þ<sup>2</sup>

<sup>j</sup> <sup>Δ</sup>ω∙tþΔφð Þ�<sup>t</sup> <sup>π</sup> ð Þ<sup>2</sup>

<sup>j</sup>ð Þ <sup>Δ</sup>ω∙tþΔφð Þ<sup>t</sup> : (7)

http://dx.doi.org/10.5772/intechopen.72715

a<sup>5</sup> ¼ a∙e

By replacing the expressions of signals a<sup>5</sup> and a<sup>6</sup> in the system of Eq. (5), we obtain

a 2 ∙e

a 2 ∙e

> a 2 ∙e

a 2 ∙e

<sup>j</sup> <sup>ω</sup>0∙tþφ<sup>6</sup> ð Þ ð Þ<sup>t</sup> <sup>¼</sup> <sup>α</sup>∙a5∙<sup>e</sup>

<sup>j</sup>ð Þ <sup>ω</sup>0tþφ<sup>5</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup>∙<sup>e</sup>

<sup>j</sup>ð Þ <sup>ω</sup>0tþφ<sup>5</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup>∙<sup>e</sup>

<sup>j</sup>ð Þ <sup>ω</sup>0tþφ<sup>5</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup>∙<sup>e</sup>

<sup>j</sup>ð Þ <sup>ω</sup>0tþφ<sup>5</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup>∙<sup>e</sup>

It should be noted that the signals at the intermediate frequency (IF) band are the results of connecting the four six-port circuit outputs to the power detectors (see Figure 1). We consider that the power delivered at the output of each ideal power detector is proportional to the

a<sup>6</sup> ¼ α∙a∙e

b1ðÞ¼� t j

b2ðÞ¼ t

b3ðÞ¼ t

b4ðÞ¼� t j

square of the RF signal magnitude [11, 13]. Under these conditions

Given that the power detectors are identical (Ki = K), then

<sup>v</sup>1ðÞ¼ <sup>t</sup> <sup>K</sup> <sup>a</sup><sup>2</sup>

vi <sup>¼</sup> Ki<sup>∙</sup> bi j j<sup>2</sup> <sup>¼</sup> Ki∙bi∙b<sup>∗</sup>

ð5Þ

89

function of the two incident waves a<sup>5</sup> and a6, as described in Eq. (5):

the following equations [11]:

The Sij parameters of the six-port circuit can be obtained directly from Figure 1. For that purpose, the S-parameter matrices [S] of the 90� hybrid coupler and the Wilkinson power divider are employed. The corresponding matrices are given in Eqs. (2) and (3):

$$[S] = \frac{1}{\sqrt{2}} \begin{bmatrix} 0 & j & 1 & 0 \\ j & 0 & 0 & 1 \\ 1 & 0 & 0 & j \\ 0 & 1 & j & 0 \end{bmatrix} \tag{2}$$
 
$$[S]\_{ij} = \begin{bmatrix} 0 & 1 & 1 \\ 1 & 1 & 1 \end{bmatrix}$$

$$[S] = -j\frac{1}{\sqrt{2}} \begin{bmatrix} 1 & 0 & 0 \\ 1 & 0 & 0 \end{bmatrix} \tag{3}$$

Thus, the global S-parameter matrix [S] of the six-port circuit in Figure 1 is obtained by Eq. (4):

$$[\mathbf{S}] = \frac{1}{2} \begin{bmatrix} 0 & 0 & 0 & 0 & -j & j \\ 0 & 0 & 0 & 0 & 1 & j \\ 0 & 0 & 0 & 0 & 1 & 1 \\ 0 & 0 & 0 & 0 & -j & -1 \\ -j & 1 & 1 & -j & 0 & 0 \\ j & j & 1 & -1 & 0 & 0 \end{bmatrix} \tag{4}$$

Using Eq. (4), we may obtain the formulas of the four waveforms, b1, b2, b3, and b4, as a function of the two incident waves a<sup>5</sup> and a6, as described in Eq. (5):

$$\begin{cases} \begin{aligned} b\_1 &= -j\frac{a\_6}{2} + j\frac{a\_6}{2} \\\\ b\_2 &= \frac{a\_5}{2} + j\frac{a\_6}{2} \\\\ b\_3 &= \frac{a\_5}{2} + \frac{a\_6}{2} \\\\ b\_4 &= -j\frac{a\_5}{2} - \frac{a\_6}{2} \end{aligned} \tag{5}$$

In order to simplify calculations, we assume that the RF signals resulting from the antenna a<sup>6</sup> and the local oscillator (LO) a<sup>5</sup> have an amplitude ratio α, a phase difference Δφ (t) = φ6(t) � φ5, and a frequency difference Δω = ω � ω<sup>0</sup> (ω = 2πf). Therefore, these signals can be expressed by the following equations [11]:

$$a\_5 = a \cdot e^{j\left(a\_0 \cdot t + \phi\_5\right)} \tag{6}$$

$$a\_6 = \alpha \cdot a \cdot \mathbf{e}^{j\left(\omega\_0 \cdot t + \varphi\_b(t)\right)} = \alpha \cdot a\_5 \cdot \mathbf{e}^{j\left(\Lambda a \cdot t + \Lambda \varphi(t)\right)}.\tag{7}$$

By replacing the expressions of signals a<sup>5</sup> and a<sup>6</sup> in the system of Eq. (5), we obtain

$$b\_1(t) = -j\frac{a}{2} \cdot e^{j\left(a\_0 t + \varphi\_\S\right)} \cdot \left[1 + \alpha \cdot e^{j\left(\Lambda a \cdot t + \Lambda \varphi(t) + \pi\right)}\right] \tag{8}$$

$$b\_2(t) = \frac{a}{2} \cdot e^{j\left(a\omega t + \varphi\_5\right)} \cdot \left[1 + \alpha \cdot e^{j\left(A\omega t + A\varphi(t) + \frac{\pi}{2}\right)}\right] \tag{9}$$

$$b\_3(t) = \frac{a}{2} \cdot e^{j\left(a\circ t + \varphi\_5\right)} \cdot \left[1 + a \cdot e^{j\left(A\circ t + A\varphi(t)\right)}\right] \tag{10}$$

$$b\_4(t) = -j\frac{a}{2} \cdot e^{j\left(a\_0 t + \varphi\_5\right)} \cdot \left[1 + \alpha \cdot e^{j\left(\Delta\alpha \cdot t + \Delta\psi(t) - \frac{\pi}{2}\right)}\right] \tag{11}$$

It should be noted that the signals at the intermediate frequency (IF) band are the results of connecting the four six-port circuit outputs to the power detectors (see Figure 1). We consider that the power delivered at the output of each ideal power detector is proportional to the square of the RF signal magnitude [11, 13]. Under these conditions

$$\left| \upsilon\_{i} = K\_{i} \cdot \left| b\_{i} \right|^{2} = K\_{i} \cdot b\_{i} \cdot b\_{i}^{\*}, \qquad i = 1, \ldots, 4 \tag{12}$$

Given that the power detectors are identical (Ki = K), then

couplers and a Wilkinson power divider. The proposed architecture makes possible to obtain, from output power measurements, the phase difference and the amplitude ratio between an unknown signal from the antenna (a6) and the reference signal coming from a local oscillator (a5). The output signals bi can be expressed as a function of the signals ai and the S-parameter Sij by

The Sij parameters of the six-port circuit can be obtained directly from Figure 1. For that purpose, the S-parameter matrices [S] of the 90� hybrid coupler and the Wilkinson power

ffiffiffi 2 p

2 6 4

Thus, the global S-parameter matrix [S] of the six-port circuit in Figure 1 is obtained by Eq. (4):

0 00 0 �j j 0 00 0 1 j 0 00 0 1 1 0 00 0 �j �1 �j 1 1 �j 0 0 j j 1 �10 0

> 011 100 100

3 7

Sijaj, i ¼ 1,…, 6 (1)

<sup>5</sup> (3)

(2)

(4)

bi <sup>¼</sup> <sup>X</sup> 6

88 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

j¼1

divider are employed. The corresponding matrices are given in Eqs. (2) and (3):

½ �¼ <sup>S</sup> <sup>1</sup> ffiffiffi 2 p

½ �¼� <sup>S</sup> <sup>j</sup> <sup>1</sup>

½ �¼ <sup>S</sup> <sup>1</sup> 2

the following linear relationship:

Figure 1. Block diagram of a six-port circuit-based front-end receiver.

$$w\_1(t) = K \frac{a^2}{4} \cdot \left\{ 1 + a^2 - 2 \cdot a \cdot \cos\left[\Delta\omega \cdot t + \Delta\varphi(t)\right] \right\} \tag{13}$$

$$w\_2(t) = K \frac{a^2}{4} \cdot \left\{ 1 + a^2 - 2 \cdot a \cdot \sin\left[\Delta \omega \cdot t + \Delta \varphi(t)\right] \right\} \tag{14}$$

millimeter-wave extension of the VNA). However, extrapolation of measurements and comparison with simulations allow assessing the circuit behavior from 57 to 60 GHz. The microphotograph in Figure 2(a) shows the fabricated six-port circuit ready for port 4 to port 5 measurements using a coplanar line to microstrip transition and a precise on-wafer measurement structure equipped with ground-signal-ground (GSG)-150 μm coplanar probes, as

Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation

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91

As can be seen in Figure 2(a), all remaining ports are terminated by matched loads (50 Ω), integrated on the same substrate using a 100 Ω per square titanium oxide thin layer. In order to avoid the metalized via holes that are complicated to achieve with accuracy and repeatability at millimeter-wave frequencies, the 50 Ω loads use a quarter wavelength open stubs as millimeter-wave RF short circuits (see ports 1, 2, 3, and 4). The outer six-port dimensions are

In order to ensure the accuracy of S-parameter measurements, the on-wafer through-reflectline (TRL) calibration technique is employed, using the calibration kits on the same ceramic alumina substrate as the devices under test (DUT) [9]. Its standards are shown in Figure 3. It consists of a thru line (T), two open circuits as reflect (R), and a short line (L). Due to the fragility of the very thin gold layer metallization (1 μm), multiple identical standards are designed on the same ceramic substrate to ensure repeatability and success of on-wafer cali-

Figure 4 shows the typical return loss measurement at port 6 (RF input) and port 5 (LO input), as well as the isolation between them. As can be seen, at the center operating frequency, around 60 GHz, all values are better than 20 dB. Moreover, the measured values are better than 15 dB at the highest frequency (64 GHz) of the unlicensed frequency band. In fact, the achieved high isolation level is a result of employing a highly isolated round-shaped Wilkinson power divider with a high-precision integrated resistor, which has been implemented with accuracy on the 100 Ω per square titanium oxide thin layer, using MHMIC technology [14], as

Figure 2. Microphotograph of the fabricated millimeter-wave six-port circuit: (a) the on-wafer S-parameter measurement

configuration and (b) the on-wafer S-parameter measurement process using GSG-150 μm coplanar probes.

shown in Figure 2(b).

described earlier.

more or less 6.5 mm by 6.5 mm.

brations and S-parameter measurements.

$$w\_3(t) = K \frac{a^2}{4} \cdot \left\{ 1 + a^2 + 2 \cdot a \cdot \cos\left[\Delta\omega \cdot t + \Delta\varphi(t)\right] \right\} \tag{15}$$

$$w\_4(t) = K \frac{a^2}{4} \cdot \left\{ 1 + a^2 + 2 \cdot a \cdot \sin\left[\Delta a \cdot t + \Delta \varphi(t)\right] \right\} \tag{16}$$

In order to generate quadrature signals IF/IQ, we use differential amplifiers in the intermediate frequency band, at outputs 1, 3, and 2, 4 (see Figure 1):

$$\boldsymbol{\sigma}\_{\rm IF}^{l}(t) = \boldsymbol{A}\_{\rm IF} \cdot \left[\boldsymbol{\upsilon}\_{3}(t) - \boldsymbol{\upsilon}\_{1}(t)\right] = \boldsymbol{a} \cdot \boldsymbol{\mathsf{K}} \cdot \boldsymbol{a}^{2} \cdot \boldsymbol{A}\_{\rm IF} \cdot \cos\left[\boldsymbol{\Delta}\boldsymbol{a} \cdot \boldsymbol{t} + \boldsymbol{\Delta}\boldsymbol{\varphi}(t)\right] \tag{17}$$

$$\boldsymbol{\sigma}\_{\rm IF}^{Q}(t) = \boldsymbol{A}\_{\rm IF} \cdot \left[\boldsymbol{v}\_{4}(t) - \boldsymbol{v}\_{2}(t)\right] = \boldsymbol{\alpha} \cdot \boldsymbol{\text{K}} \cdot \boldsymbol{a}^{2} \cdot \boldsymbol{A}\_{\rm IF} \cdot \sin\left[\boldsymbol{\Delta}\boldsymbol{a} \cdot \boldsymbol{t} + \boldsymbol{\Delta}\boldsymbol{\varphi}(t)\right] \tag{18}$$

A second frequency conversion followed by low-frequency filtering is performed. The I/Q baseband signal formulas are thus obtained:

$$I(t) = \frac{1}{2} \cdot a \cdot K \cdot a^2 \cdot A\_{IF} \cdot A\_{BB} \cdot \cos\left[\varDelta q(t)\right] \tag{19}$$

$$Q(t) = \frac{1}{2} \cdot a \cdot K \cdot a^2 \cdot A\_{IF} \cdot A\_{BB} \cdot \sin\left[\Delta\varphi(t)\right] \tag{20}$$

In fact, the baseband I/Q signal can be expressed in the complex plane by the following equation:

$$
\Gamma(t) = I(t) + Q(t) = \frac{1}{2} \cdot \alpha \cdot K \cdot a^2 \cdot A\_{IF} \cdot A\_{BB} \cdot \mathbf{e}^{i\Delta\varphi(t)} \tag{21}
$$

This expression shows that the terms AIF and ABB are related to the intermediate frequency (IF) band and baseband (BB) amplification. The receiver works in both architectures: heterodyne, according to Eqs. (17) and (18), and homodyne, according to Eqs. (19) and (20). The amplitude ratio α and the phase difference, Δφ(t) = φ6(t) � φ5, can be obtained in baseband. This relationship between the RF domains and the intermediate (IF) band as well as the baseband has highlighted the role of the six-port circuit (reflectometer) as a phase, frequency, and amplitude discriminator [11].

#### 3.1. Multi-port (six-port) circuit design and characterization

A broadband six-port circuit with an improved symmetry and rounded shapes has been designed on a thin alumina substrate (ε<sup>r</sup> = 9.9 and h = 127 μm) using the novel Wilkinson power divider/combiner and the three rounded 90� hybrid couplers [9]. The central design frequency is 60.5 GHz, in the center of an unlicensed frequency band (57 to 64 GHz).

It should be noted that all measurements are performed from 60 GHz, due to the available measurement setup capabilities (WR-12 rectangular waveguide modules for the 60–90 GHz millimeter-wave extension of the VNA). However, extrapolation of measurements and comparison with simulations allow assessing the circuit behavior from 57 to 60 GHz. The microphotograph in Figure 2(a) shows the fabricated six-port circuit ready for port 4 to port 5 measurements using a coplanar line to microstrip transition and a precise on-wafer measurement structure equipped with ground-signal-ground (GSG)-150 μm coplanar probes, as shown in Figure 2(b).

<sup>v</sup>2ðÞ¼ <sup>t</sup> <sup>K</sup> <sup>a</sup><sup>2</sup>

90 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

<sup>v</sup>3ðÞ¼ <sup>t</sup> <sup>K</sup> <sup>a</sup><sup>2</sup>

<sup>v</sup>4ðÞ¼ <sup>t</sup> <sup>K</sup> <sup>a</sup><sup>2</sup>

IFðÞ¼ <sup>t</sup> AIF∙½ �¼ <sup>v</sup>3ð Þ� <sup>t</sup> <sup>v</sup>1ð Þ<sup>t</sup> <sup>α</sup>∙K∙a<sup>2</sup>

IFðÞ¼ <sup>t</sup> AIF∙½ �¼ <sup>v</sup>4ðÞ� <sup>t</sup> <sup>v</sup>2ð Þ<sup>t</sup> <sup>α</sup>∙K∙a<sup>2</sup>

I tðÞ¼ <sup>1</sup> 2 ∙α∙K∙a<sup>2</sup>

Q tðÞ¼ <sup>1</sup> 2 ∙α∙K∙a<sup>2</sup>

<sup>Γ</sup>ðÞ¼ <sup>t</sup> I tð Þþ Q tðÞ¼ <sup>1</sup>

3.1. Multi-port (six-port) circuit design and characterization

frequency band, at outputs 1, 3, and 2, 4 (see Figure 1):

vI

vQ

equation:

amplitude discriminator [11].

baseband signal formulas are thus obtained:

<sup>4</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup><sup>2</sup> � <sup>2</sup>∙α<sup>∙</sup> sin ½ � <sup>Δ</sup>ω∙<sup>t</sup> <sup>þ</sup> <sup>Δ</sup>φð Þ<sup>t</sup> (14)

<sup>4</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup><sup>2</sup> <sup>þ</sup> <sup>2</sup>∙α<sup>∙</sup> cos ½ � <sup>Δ</sup>ω∙<sup>t</sup> <sup>þ</sup> <sup>Δ</sup>φð Þ<sup>t</sup> (15)

<sup>4</sup> <sup>∙</sup> <sup>1</sup> <sup>þ</sup> <sup>α</sup><sup>2</sup> <sup>þ</sup> <sup>2</sup>∙α<sup>∙</sup> sin ½ � <sup>Δ</sup>ω∙<sup>t</sup> <sup>þ</sup> <sup>Δ</sup>φð Þ<sup>t</sup> (16)

∙AIF∙ cos ½ � Δω∙t þ Δφð Þt (17)

∙AIF∙ sin ½ � Δω∙t þ Δφð Þt (18)

∙AIF∙ABB∙ cos ½ � Δφð Þt (19)

∙AIF∙ABB∙ sin ½ � Δφð Þt (20)

<sup>j</sup>Δφð Þ<sup>t</sup> (21)

In order to generate quadrature signals IF/IQ, we use differential amplifiers in the intermediate

A second frequency conversion followed by low-frequency filtering is performed. The I/Q

In fact, the baseband I/Q signal can be expressed in the complex plane by the following

2 ∙α∙K∙a<sup>2</sup>

This expression shows that the terms AIF and ABB are related to the intermediate frequency (IF) band and baseband (BB) amplification. The receiver works in both architectures: heterodyne, according to Eqs. (17) and (18), and homodyne, according to Eqs. (19) and (20). The amplitude ratio α and the phase difference, Δφ(t) = φ6(t) � φ5, can be obtained in baseband. This relationship between the RF domains and the intermediate (IF) band as well as the baseband has highlighted the role of the six-port circuit (reflectometer) as a phase, frequency, and

A broadband six-port circuit with an improved symmetry and rounded shapes has been designed on a thin alumina substrate (ε<sup>r</sup> = 9.9 and h = 127 μm) using the novel Wilkinson power divider/combiner and the three rounded 90� hybrid couplers [9]. The central design

It should be noted that all measurements are performed from 60 GHz, due to the available measurement setup capabilities (WR-12 rectangular waveguide modules for the 60–90 GHz

frequency is 60.5 GHz, in the center of an unlicensed frequency band (57 to 64 GHz).

∙AIF∙ABB∙e

As can be seen in Figure 2(a), all remaining ports are terminated by matched loads (50 Ω), integrated on the same substrate using a 100 Ω per square titanium oxide thin layer. In order to avoid the metalized via holes that are complicated to achieve with accuracy and repeatability at millimeter-wave frequencies, the 50 Ω loads use a quarter wavelength open stubs as millimeter-wave RF short circuits (see ports 1, 2, 3, and 4). The outer six-port dimensions are more or less 6.5 mm by 6.5 mm.

In order to ensure the accuracy of S-parameter measurements, the on-wafer through-reflectline (TRL) calibration technique is employed, using the calibration kits on the same ceramic alumina substrate as the devices under test (DUT) [9]. Its standards are shown in Figure 3. It consists of a thru line (T), two open circuits as reflect (R), and a short line (L). Due to the fragility of the very thin gold layer metallization (1 μm), multiple identical standards are designed on the same ceramic substrate to ensure repeatability and success of on-wafer calibrations and S-parameter measurements.

Figure 4 shows the typical return loss measurement at port 6 (RF input) and port 5 (LO input), as well as the isolation between them. As can be seen, at the center operating frequency, around 60 GHz, all values are better than 20 dB. Moreover, the measured values are better than 15 dB at the highest frequency (64 GHz) of the unlicensed frequency band. In fact, the achieved high isolation level is a result of employing a highly isolated round-shaped Wilkinson power divider with a high-precision integrated resistor, which has been implemented with accuracy on the 100 Ω per square titanium oxide thin layer, using MHMIC technology [14], as described earlier.

Figure 2. Microphotograph of the fabricated millimeter-wave six-port circuit: (a) the on-wafer S-parameter measurement configuration and (b) the on-wafer S-parameter measurement process using GSG-150 μm coplanar probes.

Figure 3. Microphotograph of TRL calibration kit.

Figure 4. Measured RF inputs return loss and isolation for the fabricated six-port circuit.

The measured return losses at output ports (1, 2, 3, and 4) are illustrated in Figure 5. These results exhibit a good matching at all ports, with better than 25 dB around the operating frequency of 60 GHz while keeping a good matching level for the rest of the band (better than 15 dB at the highest frequency (64 GHz)).

maximum supplementary insertion loss does not exceed 0.6 dB, reaching approximately 1.2 dB at the upper edge. The magnitude unbalance between two pairs of outputs is close to 0 dB at 62.5 GHz and is less than 0.5 dB over the frequency band of interest. It should be noted that due to the high symmetry of the designed circuit, similar results are obtained between port 5 and the other two six-port outputs (1, 3). The same is valid for the transmission coefficient

Figure 6. Typical transmission magnitude measurements for the fabricated six-port circuits.

Figure 5. Measured RF outputs return loss (ports 1, 2, 3, and 4) for the fabricated six-port circuit.

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Figure 7 shows the phase difference between the two typical transmission S-parameters, S52 and S54, as well as S61 and S63. The obtained results show two quasi-parallel characteristics. The measured phase difference between each two adjacent ports is close to the quadratic reference of 90. However, the observed phase difference error is less than 2 up to 64 GHz.

results between RF port 6 and outputs (2, 4).

Figure 6 shows the measured transmission coefficient results, commonly known as the power splitting between the LO port 5 and two adjacent outputs (ports 2 and 4), as well as between the RF port 6 and two other adjacent outputs (ports 1 and 3). As can be seen, the measured transmission coefficients are close to the value of 6 dB over the considered frequency band, according to the six-port theory. However, around the center frequency of 60 GHz, the

Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation http://dx.doi.org/10.5772/intechopen.72715 93

Figure 5. Measured RF outputs return loss (ports 1, 2, 3, and 4) for the fabricated six-port circuit.

Figure 6. Typical transmission magnitude measurements for the fabricated six-port circuits.

The measured return losses at output ports (1, 2, 3, and 4) are illustrated in Figure 5. These results exhibit a good matching at all ports, with better than 25 dB around the operating frequency of 60 GHz while keeping a good matching level for the rest of the band (better than

Figure 4. Measured RF inputs return loss and isolation for the fabricated six-port circuit.

92 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

Figure 6 shows the measured transmission coefficient results, commonly known as the power splitting between the LO port 5 and two adjacent outputs (ports 2 and 4), as well as between the RF port 6 and two other adjacent outputs (ports 1 and 3). As can be seen, the measured transmission coefficients are close to the value of 6 dB over the considered frequency band, according to the six-port theory. However, around the center frequency of 60 GHz, the

15 dB at the highest frequency (64 GHz)).

Figure 3. Microphotograph of TRL calibration kit.

maximum supplementary insertion loss does not exceed 0.6 dB, reaching approximately 1.2 dB at the upper edge. The magnitude unbalance between two pairs of outputs is close to 0 dB at 62.5 GHz and is less than 0.5 dB over the frequency band of interest. It should be noted that due to the high symmetry of the designed circuit, similar results are obtained between port 5 and the other two six-port outputs (1, 3). The same is valid for the transmission coefficient results between RF port 6 and outputs (2, 4).

Figure 7 shows the phase difference between the two typical transmission S-parameters, S52 and S54, as well as S61 and S63. The obtained results show two quasi-parallel characteristics. The measured phase difference between each two adjacent ports is close to the quadratic reference of 90. However, the observed phase difference error is less than 2 up to 64 GHz.

3.2. Millimeter-wave power detector design and characterization

resistance of the diode, this characteristic will be expressed by

K is the Boltzmann constant, and T is the temperature.

Knowing that the voltage VRF (t) can be expressed by

i tðÞ¼ Is

Figure 9. Typical configuration of a zero-bias Schottky diode-based power detector.

exponential function to obtain

presented as follows (Figure 10) [16]:

As we have seen in the previous section, to recover the low IF or the baseband signals, the implementation of power detectors at the four outputs of the six-port circuit is required. For that purpose, the HSCH-9161 millimeter-wave zero-bias GaAs Schottky diode of Keysight Technologies is selected for power detection, due to its broadband and high-speed properties [15, 16]. The typical configuration of the power detector usually includes the Schottky diode

Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation

The nonlinear characteristic between the current i(t) that passes through the diode and the input RF voltage VRF(t) is generally described by the Schottky law. By neglecting the parasitic

where Is is the saturation current, qo is the charge of the electron, n is the coefficient of ideality,

On the other hand, considering that the input signal VRF (t) has a low power and that it satisfies the condition, A < VT, then we can reexpress Eq. (22) by using the limited development of the

> þ 1 2

Moreover, the low-frequency equivalent circuit at the output of the power detector may be

The dynamic resistance of the diode RV represents the video resistance [16]. The latter with the resistor R and the capacitor C forms a first-order low-pass filter having a cutoff frequency fc:

vRFð Þt nVT � �<sup>2</sup>

" #

vRFð Þt nVT � �

nKT � � � <sup>1</sup>

� � (22)

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vRFðÞ¼ t A∙ cos ð Þ ωRFt , ωRF ¼ 2πf RF (23)

þ …

(24)

followed by a low-pass filter to extract the DC component, as illustrated in Figure 9.

i tðÞ¼ Is exp qovRFð Þ<sup>t</sup>

Figure 7. Typical transmission phase measurements for the fabricated six-port circuits.

The plot of the q<sup>i</sup> points using the S-parameter measurement results of the proposed six-port circuits is shown in Figure 8. As can be seen, the q<sup>i</sup> points are positioned equidistantly from the origin and angularly spaced by 360 divided their number (i = 4 in this case). These results underline the performance of the fabricated six-port circuit and prove the high accuracy location of the q<sup>i</sup> points over the considered 60 GHz frequency band. Consequently, the magnitudes of the q<sup>i</sup> points are equal and closer to 1, while the argument difference is closer to 90 between two corresponding q<sup>i</sup> points.

Figure 8. The q<sup>i</sup> points of the fabricated six-port circuit for the considered frequency band 60–65 GHz.

#### 3.2. Millimeter-wave power detector design and characterization

As we have seen in the previous section, to recover the low IF or the baseband signals, the implementation of power detectors at the four outputs of the six-port circuit is required. For that purpose, the HSCH-9161 millimeter-wave zero-bias GaAs Schottky diode of Keysight Technologies is selected for power detection, due to its broadband and high-speed properties [15, 16]. The typical configuration of the power detector usually includes the Schottky diode followed by a low-pass filter to extract the DC component, as illustrated in Figure 9.

The nonlinear characteristic between the current i(t) that passes through the diode and the input RF voltage VRF(t) is generally described by the Schottky law. By neglecting the parasitic resistance of the diode, this characteristic will be expressed by

$$i(t) = I\_s \left[ \exp\left(\frac{q\_o v\_{RF}(t)}{nKT}\right) - 1\right] \tag{22}$$

where Is is the saturation current, qo is the charge of the electron, n is the coefficient of ideality, K is the Boltzmann constant, and T is the temperature.

Knowing that the voltage VRF (t) can be expressed by

The plot of the q<sup>i</sup> points using the S-parameter measurement results of the proposed six-port circuits is shown in Figure 8. As can be seen, the q<sup>i</sup> points are positioned equidistantly from the origin and angularly spaced by 360 divided their number (i = 4 in this case). These results underline the performance of the fabricated six-port circuit and prove the high accuracy location of the q<sup>i</sup> points over the considered 60 GHz frequency band. Consequently, the magnitudes of the q<sup>i</sup> points are equal and closer to 1, while the argument difference is closer

Figure 7. Typical transmission phase measurements for the fabricated six-port circuits.

94 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

Figure 8. The q<sup>i</sup> points of the fabricated six-port circuit for the considered frequency band 60–65 GHz.

to 90 between two corresponding q<sup>i</sup> points.

$$
\sigma\_{\rm RF}(t) = A \cdot \cos \left(\omega\_{\rm RF} t\right), \ \omega\_{\rm RF} = 2\pi f\_{\rm RF} \tag{23}
$$

On the other hand, considering that the input signal VRF (t) has a low power and that it satisfies the condition, A < VT, then we can reexpress Eq. (22) by using the limited development of the exponential function to obtain

$$\dot{\mathbf{u}}(t) = I\_s \left[ \left( \frac{v\_{\rm RF}(t)}{nV\_T} \right) + \frac{1}{2} \left( \frac{v\_{\rm RF}(t)}{nV\_T} \right)^2 + \dots \right] \tag{24}$$

Moreover, the low-frequency equivalent circuit at the output of the power detector may be presented as follows (Figure 10) [16]:

The dynamic resistance of the diode RV represents the video resistance [16]. The latter with the resistor R and the capacitor C forms a first-order low-pass filter having a cutoff frequency fc:

Figure 9. Typical configuration of a zero-bias Schottky diode-based power detector.

Figure 10. Equivalent circuit for the output of the Schottky diode-based power detector.

$$f\_c = \frac{R\_v + R}{2\pi R\_v R \mathcal{C}}\tag{25}$$

and have the role of providing resistive input impedance that enables broadband operation

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At the diode output, a broadband low-pass filtering made from two pairs of quarter-wave reflectors has been performed. This operation allows extraction of the DC voltage signals while suppressing all undesired higher-frequency components. In order to maximize the output detection voltage, a high-impedance integrated grounded resistor of 4 KΩ has also been added at the diode detector output. It uses the same 100 Ω per square titanium oxide thin layer.

The simulated and measured return loss at the RF input of the fabricated power detector circuit is compared, with good agreement, in Figure 12. As can be observed, the measured impedance bandwidth at 10 dB covers the frequency range of 3.8 GHz, from 60 to 63.8 GHz,

The simulated and the measured results of output power versus input power, at 61.9 GHz, are shown in Figure 13. As can be seen, a good concordance is achieved between measurement and simulation based on the diode model using the Keysight's Advanced Design System

which represents a bandwidth of 6.13%, at the center frequency of 61.9 GHz.

Figure 12. Simulated and measured return loss of the fabricated millimeter-wave power detector.

[15]. The quarter-wave length radial stub provides RF ground.

Figure 11. Photograph of the fabricated power detector prototype.

By choosing a low-frequency cutoff fc compared to the RF input frequency of the power detector, the output voltage Vo(t) will therefore be proportional to the low-frequency or baseband (BB) components of the current i(t), particularly to the quadratic term of Eq. (24).

Then, by replacing the expression of the input RF voltage given by Eq. (23) in Eq. (24) and taking into account only the quadratic term of the equation, we obtain

$$\dot{a}(t) = \frac{I\_S}{2} \left( \frac{A \cdot \cos\left(2\pi f\_{RF}t\right)}{V\_T} \right)^2 \tag{26}$$

After a low-pass filtering operation, the output voltage will be expressed as follows:

$$w\_o(t) = \frac{R \cdot R\_V}{R + R\_V} \left(\frac{I\_S}{4V\_T^2}\right)^2 \cdot A^2 = \alpha \cdot P\_{RF} \tag{27}$$

The coefficient α represents the sensitivity of the power detector, usually expressed in volts/ watt. According to the formula, it can be seen that for the low power levels, the detector can perform power detection because the output voltage of the detector is proportional to the square of the input signal amplitude or, in other words, to the power of the RF signal.

The photograph of the fabricated millimeter-wave power detector circuit used in the proposed front-end receiver architecture is shown in Figure 11. It comprises three main parts: the input impedance matching circuit, an HSCH-9161 millimeter-wave GaAs Schottky diode (zero bias), and the detected DC voltage output circuit. Obviously, all parts are designed and implemented on a thin ceramic substrate (ε<sup>r</sup> = 9.9, h = 127 μm), using an MHMIC fabrication process. The input of the diode is well matched to 50 Ω using an accurate impedance matching stage to achieve the maximum transmission of the RF signal to diode input. The proposed impedance matching networks include an open-circuit stub of 0.16λ in parallel with the main 50 Ω microstrip line for matching purposes, as well as quarter-wave length-paralleled radial stubs attached to a metalized via-hole through a high-impedance quarter-wave length microstrip line, to prevent the RF signal leakage, providing DC ground. The integrated high-precision resistors of 100 Ω (parallel) are implemented on a 100 Ω per square titanium oxide thin layer Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation http://dx.doi.org/10.5772/intechopen.72715 97

Figure 11. Photograph of the fabricated power detector prototype.

<sup>f</sup> <sup>c</sup> <sup>¼</sup> Rv <sup>þ</sup> <sup>R</sup>

By choosing a low-frequency cutoff fc compared to the RF input frequency of the power detector, the output voltage Vo(t) will therefore be proportional to the low-frequency or baseband (BB) components of the current i(t), particularly to the quadratic term of Eq. (24).

Then, by replacing the expression of the input RF voltage given by Eq. (23) in Eq. (24) and

<sup>A</sup><sup>∙</sup> cos 2π<sup>f</sup> RF<sup>t</sup> VT <sup>2</sup>

> IS 4VT 2 <sup>2</sup>

The coefficient α represents the sensitivity of the power detector, usually expressed in volts/ watt. According to the formula, it can be seen that for the low power levels, the detector can perform power detection because the output voltage of the detector is proportional to the

The photograph of the fabricated millimeter-wave power detector circuit used in the proposed front-end receiver architecture is shown in Figure 11. It comprises three main parts: the input impedance matching circuit, an HSCH-9161 millimeter-wave GaAs Schottky diode (zero bias), and the detected DC voltage output circuit. Obviously, all parts are designed and implemented on a thin ceramic substrate (ε<sup>r</sup> = 9.9, h = 127 μm), using an MHMIC fabrication process. The input of the diode is well matched to 50 Ω using an accurate impedance matching stage to achieve the maximum transmission of the RF signal to diode input. The proposed impedance matching networks include an open-circuit stub of 0.16λ in parallel with the main 50 Ω microstrip line for matching purposes, as well as quarter-wave length-paralleled radial stubs attached to a metalized via-hole through a high-impedance quarter-wave length microstrip line, to prevent the RF signal leakage, providing DC ground. The integrated high-precision resistors of 100 Ω (parallel) are implemented on a 100 Ω per square titanium oxide thin layer

taking into account only the quadratic term of the equation, we obtain

Figure 10. Equivalent circuit for the output of the Schottky diode-based power detector.

96 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

voðÞ¼ t

i tðÞ¼ IS 2

After a low-pass filtering operation, the output voltage will be expressed as follows:

square of the input signal amplitude or, in other words, to the power of the RF signal.

R∙RV R þ RV <sup>2</sup>πRvRC (25)

<sup>∙</sup>A<sup>2</sup> <sup>¼</sup> <sup>α</sup>∙PRF (27)

(26)

and have the role of providing resistive input impedance that enables broadband operation [15]. The quarter-wave length radial stub provides RF ground.

At the diode output, a broadband low-pass filtering made from two pairs of quarter-wave reflectors has been performed. This operation allows extraction of the DC voltage signals while suppressing all undesired higher-frequency components. In order to maximize the output detection voltage, a high-impedance integrated grounded resistor of 4 KΩ has also been added at the diode detector output. It uses the same 100 Ω per square titanium oxide thin layer.

The simulated and measured return loss at the RF input of the fabricated power detector circuit is compared, with good agreement, in Figure 12. As can be observed, the measured impedance bandwidth at 10 dB covers the frequency range of 3.8 GHz, from 60 to 63.8 GHz, which represents a bandwidth of 6.13%, at the center frequency of 61.9 GHz.

The simulated and the measured results of output power versus input power, at 61.9 GHz, are shown in Figure 13. As can be seen, a good concordance is achieved between measurement and simulation based on the diode model using the Keysight's Advanced Design System

Figure 12. Simulated and measured return loss of the fabricated millimeter-wave power detector.

Figure 13. Simulated and measured detected power versus input power at 61 GHz.

(ADS) software. The proposed power detector shows a measured dynamic range (detection range) in the linear region of more than 42 dB. A high sensitivity is also achieved; the minimum detectable input power level using digital voltmeter is approximately �48 dBm.

#### 3.3. V-band low-noise amplifier (LNA) implementation

A low-noise amplifier (LNA) represents the head amplifier of the receiving chain. It is often mounted as close as practical to the antenna, in order to amplify signals having a very low power level. However, the amplification of the signal received by the amplifier must meet two important criteria: maintain a stable and appropriate gain and control the noise figure (NF) of the receiver. In other words, a trade-off between the noise factor and the gain is therefore necessary in the LNA design. Generally, the noise factor F describes the signal-to-noise ratio degradation caused by the RF chain components. It is defined as the ratio of the input SNR (signal-to-noise ratio) to the output SNR of the receiver system:

$$\text{SNR} = \frac{\text{SNR}\_{\text{IN}}}{\text{SNR}\_{\text{OUT}}} \tag{28}$$

fabrication and integration with any planar fabrication technology, such as the miniature hybrid microwave integrated circuit (MHMIC) or the monolithic microwave integrated circuit

Figure 14. Photograph of typical low-noise amplifier TGA4600 implementation on a thin ceramic substrate.

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Technology 0.15 μm 3MI pHEMT

NF 4 dB Gain 13 dB Typical frequency range 57–65 GHz Input/output impedance 50 Ω Input/output return loss 26 dB/6 dB Reverse isolation 20 dB Stability factor >1

Table 1. Typical characteristics of TGA4600 low-noise amplifier at 60 GHz.

In this section, a high-gain 8 2 element microstrip patch antenna array has been designed to be integrated in the proposed 60 GHz millimeter-wave RF front-end receiver prototype, for high-data-rate indoor wireless applications. The proposed array configuration was simulated using the Advanced Design System (ADS) software from Keysight Technologies and tested using the vector network analyzer (E8362B) with millimeter-wave extension modules of the

(MMIC) [18].

same company.

In the designed front-end receiver prototype, the low-noise amplifier TGA4600 from TriQuint Semiconductor company has been selected. The latter has a reasonable noise factor, NF = 4 dB, allowing to significantly limit the noise contribution of the reception chain. The typical implementation of the employed low-noise amplifier with gold bonding wires and ribbons on a ceramic substrate is illustrated in a photograph of Figure 14. Typical characteristics at 60 GHz are also presented in Table 1.

#### 3.4. Millimeter-wave microstrip antenna array design

Millimeter-wave microstrip patch antennas are a promising alternative to the future wireless communication technologies in various fields including military, industrial, and commercial [17]. This is primarily due to their small size, low cost, and light weight, as well as the ease of Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation http://dx.doi.org/10.5772/intechopen.72715 99

Figure 14. Photograph of typical low-noise amplifier TGA4600 implementation on a thin ceramic substrate.


Table 1. Typical characteristics of TGA4600 low-noise amplifier at 60 GHz.

(ADS) software. The proposed power detector shows a measured dynamic range (detection range) in the linear region of more than 42 dB. A high sensitivity is also achieved; the mini-

A low-noise amplifier (LNA) represents the head amplifier of the receiving chain. It is often mounted as close as practical to the antenna, in order to amplify signals having a very low power level. However, the amplification of the signal received by the amplifier must meet two important criteria: maintain a stable and appropriate gain and control the noise figure (NF) of the receiver. In other words, a trade-off between the noise factor and the gain is therefore necessary in the LNA design. Generally, the noise factor F describes the signal-to-noise ratio degradation caused by the RF chain components. It is defined as the ratio of the input SNR

> SNR <sup>¼</sup> SNRIN SNROUT

In the designed front-end receiver prototype, the low-noise amplifier TGA4600 from TriQuint Semiconductor company has been selected. The latter has a reasonable noise factor, NF = 4 dB, allowing to significantly limit the noise contribution of the reception chain. The typical implementation of the employed low-noise amplifier with gold bonding wires and ribbons on a ceramic substrate is illustrated in a photograph of Figure 14. Typical characteristics at 60 GHz

Millimeter-wave microstrip patch antennas are a promising alternative to the future wireless communication technologies in various fields including military, industrial, and commercial [17]. This is primarily due to their small size, low cost, and light weight, as well as the ease of

(28)

mum detectable input power level using digital voltmeter is approximately �48 dBm.

3.3. V-band low-noise amplifier (LNA) implementation

Figure 13. Simulated and measured detected power versus input power at 61 GHz.

98 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

(signal-to-noise ratio) to the output SNR of the receiver system:

3.4. Millimeter-wave microstrip antenna array design

are also presented in Table 1.

fabrication and integration with any planar fabrication technology, such as the miniature hybrid microwave integrated circuit (MHMIC) or the monolithic microwave integrated circuit (MMIC) [18].

In this section, a high-gain 8 2 element microstrip patch antenna array has been designed to be integrated in the proposed 60 GHz millimeter-wave RF front-end receiver prototype, for high-data-rate indoor wireless applications. The proposed array configuration was simulated using the Advanced Design System (ADS) software from Keysight Technologies and tested using the vector network analyzer (E8362B) with millimeter-wave extension modules of the same company.

The geometry of the fabricated microstrip 8 2 antenna array, including its single patch element, is shown in the photograph at Figure 15. The proposed array architecture adopts a corporate feed network connected to a 50 Ω coplanar feed line to carry out on-wafer measurements through ground-signal-ground (GSG)-150 μm coplanar probes. The employed corporate microstrip feed network includes a Wilkinson power divider/combiner and multiple tee junctions, which are interconnected by microstrip lines of 50 Ω and 70.7 Ω characteristic impedances to allow impedance matching, as well as better control over the phase and amplitude of each single patch element. This approach provides high directivity, improves radiation efficiency, and reduces beam fluctuations, over the suggested frequency range, compared to other array configurations [17]. The geometrical parameters of the proposed antennas are Wa = 18.56 mm, La = 5.01 mm, W = 1.07 mm, L = 0.74 mm, d = 1.43 mm, and d<sup>1</sup> = 1.60 mm.

The simulated and measured return losses of the single patch antenna, as well as the 8 2 microstrip array, are compared in Figure 16. As can be seen, good concordances are achieved

> over the considered frequency range (60–65 GHz). However, the 10 dB antenna bandwidths cover from 60 GHz to 61.2 GHz for the single patch and from 60 GHz to 61.7 GHz for the 8 2 microstrip array, which represent, respectively, bandwidths of 1.2 GHz (2%) and 1.7 GHz

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Figure 17. 2D simulated radiation pattern of the proposed 8 2 microstrip antenna array at 60.5 GHz.

The simulated 2D radiation pattern in E-plane at 60.5 GHz is illustrated at Figure 17. As can be observed, the simulated radiation pattern maintains a high symmetry and a good broadside radiation pattern. The maximum gain and directivity are 16.8 dB and 17.9 dB, respectively. The simulated antenna efficiency is around 77.43%. The half-power beamwidth (HPBW) is about

In this section, we assemble all the components detailed above to form the final prototype of the fully integrated 60 GHz front-end receiver. The latter is shown in photograph at Figure 18. It therefore includes an 8 2 antenna array, a low-noise amplifier (LNA), a six-port circuit, and power detectors, integrated on a same 2.54 cm 2.54 cm ceramic substrate. In summary, its function and principle of operation are as follows: the RF signal enters at port 6, after being received by a 16-element patch antenna array (16.8 dB Gain) and amplified by the TGA4600 LNA from TriQuint Semiconductor (57–65 GHz, 13 dB Gain, and 4 dB noise figure). The reference signal from the local oscillator (LO) enters at port 5 through a microstrip to WR12 rectangular waveguide (RW) transition. In order to recover the low IF or the baseband signals,

The fabricated 60 GHz six-port front-end receiver has been tested using the test bench illustrated in the block diagram of Figure 19 and the corresponding photograph at Figure 20. In the transmitting part, the HP 8360 Series Synthesized Sweeper output is connected to a commercial K-Band power amplifier with a gain of around 10 dB. Limited by the upper output frequency range of the frequency synthesizer (40 GHz), an additional millimeter-wave frequency multiplier module (x3), model SFP-123KF-S1, from SAGE Millimeter, Inc., is then used to achieve a frequency around 61.71 GHz (20.57 GHz 3). The obtained signal is combined

12, and the first-side lobe is about 12.8 dB below the main lobe.

the four six-port outputs are connected to the RF power detectors.

3.5. Millimeter-wave multi-port (six-port) front-end receiver implementation

(2.83%), respectively.

Figure 15. Photograph of the fabricated prototype with the geometrical parameters of (a) the 8 2 microstrip array antenna and (b) the single patch.

Figure 16. Measured and simulated return loss of (a) the 8 2 microstrip array antenna and (b) the single patch.

Figure 17. 2D simulated radiation pattern of the proposed 8 2 microstrip antenna array at 60.5 GHz.

The geometry of the fabricated microstrip 8 2 antenna array, including its single patch element, is shown in the photograph at Figure 15. The proposed array architecture adopts a corporate feed network connected to a 50 Ω coplanar feed line to carry out on-wafer measurements through ground-signal-ground (GSG)-150 μm coplanar probes. The employed corporate microstrip feed network includes a Wilkinson power divider/combiner and multiple tee junctions, which are interconnected by microstrip lines of 50 Ω and 70.7 Ω characteristic impedances to allow impedance matching, as well as better control over the phase and amplitude of each single patch element. This approach provides high directivity, improves radiation efficiency, and reduces beam fluctuations, over the suggested frequency range, compared to other array configurations [17]. The geometrical parameters of the proposed antennas are Wa = 18.56 mm, La = 5.01 mm, W = 1.07 mm, L = 0.74 mm, d = 1.43 mm, and d<sup>1</sup> = 1.60 mm.

100 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

The simulated and measured return losses of the single patch antenna, as well as the 8 2 microstrip array, are compared in Figure 16. As can be seen, good concordances are achieved

Figure 15. Photograph of the fabricated prototype with the geometrical parameters of (a) the 8 2 microstrip array

Figure 16. Measured and simulated return loss of (a) the 8 2 microstrip array antenna and (b) the single patch.

antenna and (b) the single patch.

over the considered frequency range (60–65 GHz). However, the 10 dB antenna bandwidths cover from 60 GHz to 61.2 GHz for the single patch and from 60 GHz to 61.7 GHz for the 8 2 microstrip array, which represent, respectively, bandwidths of 1.2 GHz (2%) and 1.7 GHz (2.83%), respectively.

The simulated 2D radiation pattern in E-plane at 60.5 GHz is illustrated at Figure 17. As can be observed, the simulated radiation pattern maintains a high symmetry and a good broadside radiation pattern. The maximum gain and directivity are 16.8 dB and 17.9 dB, respectively. The simulated antenna efficiency is around 77.43%. The half-power beamwidth (HPBW) is about 12, and the first-side lobe is about 12.8 dB below the main lobe.

#### 3.5. Millimeter-wave multi-port (six-port) front-end receiver implementation

In this section, we assemble all the components detailed above to form the final prototype of the fully integrated 60 GHz front-end receiver. The latter is shown in photograph at Figure 18. It therefore includes an 8 2 antenna array, a low-noise amplifier (LNA), a six-port circuit, and power detectors, integrated on a same 2.54 cm 2.54 cm ceramic substrate. In summary, its function and principle of operation are as follows: the RF signal enters at port 6, after being received by a 16-element patch antenna array (16.8 dB Gain) and amplified by the TGA4600 LNA from TriQuint Semiconductor (57–65 GHz, 13 dB Gain, and 4 dB noise figure). The reference signal from the local oscillator (LO) enters at port 5 through a microstrip to WR12 rectangular waveguide (RW) transition. In order to recover the low IF or the baseband signals, the four six-port outputs are connected to the RF power detectors.

The fabricated 60 GHz six-port front-end receiver has been tested using the test bench illustrated in the block diagram of Figure 19 and the corresponding photograph at Figure 20. In the transmitting part, the HP 8360 Series Synthesized Sweeper output is connected to a commercial K-Band power amplifier with a gain of around 10 dB. Limited by the upper output frequency range of the frequency synthesizer (40 GHz), an additional millimeter-wave frequency multiplier module (x3), model SFP-123KF-S1, from SAGE Millimeter, Inc., is then used to achieve a frequency around 61.71 GHz (20.57 GHz 3). The obtained signal is combined

Figure 18. The fabricated six-port front-end receiver prototype.

Figure 20. Photograph of the test bench for the fabricated 60 GHz six-port front-end receiver.

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Figure 21. Experimental constellation results of the demodulated MPSK/MQAM signals.

Figure 19. Block diagram of the test bench for the fabricated six-port front-end receiver prototype.

with the 600 MHz intermediate frequency (IF) signal from Agilent E4438 source, through a balanced RF mixer. In order to transmit the modulated signal, a horn antenna operating at the 60 GHz band, with a gain of about 22 dBi, is also employed.

In the receiving part, a frequency multiplier (x6) is used to generate the LO signal having the frequency of 62.19 GHz, at port 5 of the fabricated 60 GHz front-end receiver prototype (signal coming from Anritsu 68347C Synthesized Signal Generator at 10.38 GHz (6 10.38 GHz)). An attenuator and phase shifter are also used to control the power level and phase of the LO signal in millimeter-wave band (V-band), respectively. The four output signals, from the 60 GHz front-end receiver prototype, are displayed and recorded using a Tektronix digital phosphor oscilloscope (DPO7054), with 1 MΩ input impedance.

The demodulation results of various signals, from 4 to 32 symbols, are therefore obtained as shown in Figure 21. As can be observed in these captures, the symbols of BPSK, 8PSK, and 16PSK demodulations are distributed evenly around the circle. Likewise, for the QPSK constellation diagram, a quasi-perfectly square shape is achieved, whereas for the 16QAM and Millimeter-Wave Multi-Port Front-End Receivers: Design Considerations and Implementation http://dx.doi.org/10.5772/intechopen.72715 103

Figure 20. Photograph of the test bench for the fabricated 60 GHz six-port front-end receiver.

Figure 21. Experimental constellation results of the demodulated MPSK/MQAM signals.

with the 600 MHz intermediate frequency (IF) signal from Agilent E4438 source, through a balanced RF mixer. In order to transmit the modulated signal, a horn antenna operating at the

Figure 19. Block diagram of the test bench for the fabricated six-port front-end receiver prototype.

102 Advanced Electronic Circuits - Principles, Architectures and Applications on Emerging Technologies

In the receiving part, a frequency multiplier (x6) is used to generate the LO signal having the frequency of 62.19 GHz, at port 5 of the fabricated 60 GHz front-end receiver prototype (signal coming from Anritsu 68347C Synthesized Signal Generator at 10.38 GHz (6 10.38 GHz)). An attenuator and phase shifter are also used to control the power level and phase of the LO signal in millimeter-wave band (V-band), respectively. The four output signals, from the 60 GHz front-end receiver prototype, are displayed and recorded using a Tektronix digital phosphor

The demodulation results of various signals, from 4 to 32 symbols, are therefore obtained as shown in Figure 21. As can be observed in these captures, the symbols of BPSK, 8PSK, and 16PSK demodulations are distributed evenly around the circle. Likewise, for the QPSK constellation diagram, a quasi-perfectly square shape is achieved, whereas for the 16QAM and

60 GHz band, with a gain of about 22 dBi, is also employed.

Figure 18. The fabricated six-port front-end receiver prototype.

oscilloscope (DPO7054), with 1 MΩ input impedance.

32QAM, the points are almost equidistant, proving the front-end discrimination's qualities in both amplitude and phase, according to the six-port demodulation theory [12]. As can be seen, the phase and amplitude errors are minimal, not exceeding a few percent for each constellation point. It should be pointed out that those errors are mainly due to synchronization of transmitting and receiving equipment (phase noise), as well as the fabrication tolerances (symmetry of the constellation points).

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