**3. Manufacturing of waveguide devices**

After addressing the first application of low-cost 3D printing in the microwave field, another step is introduced. Now, the printed devices are meant to be measured, and therefore, they have to be metallized first. These two requirements set a frame of limitations, which confine the usability of this technology—at least the low-cost version—and are crucial to understand the electrical responses of the devices presented later in this chapter.

First, the three main restrictions are thoroughly explained. Then, a full set of waveguide devices and two horn antennas are presented. They cover from relatively simple straight sections to fully operational devices such as a waveguide diplexer with challenging specifications. They try to push low-cost 3D printing to its limits, also in terms of frequency and bandwidth.

#### **3.1. Limitations of low-cost 3D printing process**

The main drawbacks of low-cost 3D printing when manufacturing fully functional devices may be summarized as follows: the working frequency, the metallization process, and the structural issues.

#### *3.1.1. Dimensions*

The first one is intrinsically related to the dimensions of the design and the printing accuracy, that is, the layer height. In a normal low-cost machine, the minimum value is ±0.1 mm. It marks the minimum variation of dimensions that can be built, and, therefore, it establishes an upper limit for the working frequency. It is unrealistic to print devices with working band above 15 GHz. Besides this limit, this issue becomes even more evident when dealing with sensitive devices, where a tiny variation of dimensions leads to a very different electrical behavior.

Printing accuracy is also related to the achievable matching losses in a waveguide device. In an experiment performed with printed 50-mm-long WR75 waveguide sections such as the one in **Figure 5**, the matching level is similar for the waveguides regardless of the paint they are covered with: around 20–25 dB in Ku band, as it appears in **Figure 6**.

All things considered, this limitation is probably the easiest to deal with. There is a quite clear limit regarding frequency, and computational design tools are of great help to account for precision. That means that before obtaining the final model of the device, all geometrical dimensions should have the number of decimal digits in agreement with the printing accuracy.

#### *3.1.2. Metallization*

In all these cases, very complex designs with reduced price are rapidly available. Some geometries involve many difficult details, even with the available visualization tools nowadays. These cases are therefore good examples of how 3D printing can help to reach a clearer com-

**Figure 4.** Printed layers of an ortho-mode transducer (OMT) based on turnstile junction, working in W-band.

munication between the different teams involved in an engineering product.

**Figure 3.** Printed model of a Ka-band diplexer with two band-pass filters with elliptical response.

**Figure 2.** 3D–printed prototype of a Ku-band 16-port power combiner.

98 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

The second drawback includes all the aspects related to the metallization process. As it has been mentioned, it is imperative to cover the inner surface of the printed waveguide devices with metal paint, in order to confine the electromagnetic energy within the structures. Consequently, the devices cannot be printed as a single piece, as it would be desirable, but they have to be printed in separate parts instead. Their inner part must be accessible for painting and that may limit the feasible geometries in some cases. This may be seen as damaging one of the biggest advantages of 3D printing, yet it is a trade-off and from this perspective the total cost is very low.

Two other metal paints with silver content, supplied by Ferro [11] and RS [12], were considered. In both cases, the conductivity was provided, although the value given was for DC. In the same experiment with the waveguide sections, insertion losses were also analyzed. Their provided nominal conductivity was σ = 105 S/m (RS) and σ = 1.6–5·10<sup>6</sup> S/m (Ferro), which is one or two orders of magnitude below silver conductivity itself (σ = 6.3·10<sup>7</sup> S/m). However, the effective conductivity is better for RS paint in Ku band, as shown in **Figure 7**. Such results are critical and RS paint is the natural choice to metallize 3D–printed devices. Its price was approximately 60 \$/20 g at the time of the experiment. The effective conductivity used in subsequent simulations was σ = 5·10<sup>4</sup> S/m. Metallization was performed by hand and each section was painted twice to ensure a homogeneous layer with no air gaps or drops of paint remaining. In fact, conductivity is not the only issue affecting the response. That may explain why the effective conductivity of RS paint is higher than Ferro, although for nominal values it is the other way around. The density of the paint may play an important role as well. For instance, the Ferro paint is not very thick and therefore it may not be well absorbed by the plastic, resulting in a defective metallization process.

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The problems derived from the metallization process are not only related to the paint itself. Since the waveguide structures need to be split, the eventual alignment turns out to be critical. The different parts must be joined back together and glued, since any misalignment would cause a significant variation in the electrical response. Extra parts such as fastening pieces

Finally, the influence of the plastic spools may be worth mentioning, since the quality of the plastic affects how well the paint is slicked to the plastic. Plastic characteristics may vary from one supplier

**Figure 7.** Measured insertion losses of 3D printed 50-mm-long WR75 waveguides painted with RS and Ferro paint.

may also been used, as mentioned later in the chapter.

**Figure 5.** 3D printed WR75 waveguide section ready to be measured.

**Figure 6.** Measured matching level of 3D printed 50-mm-long WR75 waveguide sections covered with RS and Ferro paint.

Besides this trade-off, it is crucial to find a metal-loaded paint with good quality and a reasonable price. It is a hard task, since the obtained conductivity is not very high and depends on frequency. It causes relevant insertion losses which may clearly deteriorate and even *hide* narrow-band responses. For this work, several commercial paints were tested before manufacturing complex devices. The first one was a repair kit for rear windscreen defoggers [10]. The conductivity value was claimed to be high due to the silver content, yet no specific value was provided. Conducted tests showed high insertion losses, and thus the paint was left out.

Two other metal paints with silver content, supplied by Ferro [11] and RS [12], were considered. In both cases, the conductivity was provided, although the value given was for DC. In the same experiment with the waveguide sections, insertion losses were also analyzed. Their provided nominal conductivity was σ = 105 S/m (RS) and σ = 1.6–5·10<sup>6</sup> S/m (Ferro), which is one or two orders of magnitude below silver conductivity itself (σ = 6.3·10<sup>7</sup> S/m). However, the effective conductivity is better for RS paint in Ku band, as shown in **Figure 7**. Such results are critical and RS paint is the natural choice to metallize 3D–printed devices. Its price was approximately 60 \$/20 g at the time of the experiment. The effective conductivity used in subsequent simulations was σ = 5·10<sup>4</sup> S/m.

Metallization was performed by hand and each section was painted twice to ensure a homogeneous layer with no air gaps or drops of paint remaining. In fact, conductivity is not the only issue affecting the response. That may explain why the effective conductivity of RS paint is higher than Ferro, although for nominal values it is the other way around. The density of the paint may play an important role as well. For instance, the Ferro paint is not very thick and therefore it may not be well absorbed by the plastic, resulting in a defective metallization process.

The problems derived from the metallization process are not only related to the paint itself. Since the waveguide structures need to be split, the eventual alignment turns out to be critical. The different parts must be joined back together and glued, since any misalignment would cause a significant variation in the electrical response. Extra parts such as fastening pieces may also been used, as mentioned later in the chapter.

Finally, the influence of the plastic spools may be worth mentioning, since the quality of the plastic affects how well the paint is slicked to the plastic. Plastic characteristics may vary from one supplier

**Figure 7.** Measured insertion losses of 3D printed 50-mm-long WR75 waveguides painted with RS and Ferro paint.

Besides this trade-off, it is crucial to find a metal-loaded paint with good quality and a reasonable price. It is a hard task, since the obtained conductivity is not very high and depends on frequency. It causes relevant insertion losses which may clearly deteriorate and even *hide* narrow-band responses. For this work, several commercial paints were tested before manufacturing complex devices. The first one was a repair kit for rear windscreen defoggers [10]. The conductivity value was claimed to be high due to the silver content, yet no specific value was provided. Conducted tests showed high insertion losses, and thus the paint was left out.

**Figure 6.** Measured matching level of 3D printed 50-mm-long WR75 waveguide sections covered with RS and Ferro paint.

**Figure 5.** 3D printed WR75 waveguide section ready to be measured.

100 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

to another, and even between spools or colors. Consequently, the plastic may absorb the paint differently. The behavior with humidity or temperature variations may vary as well. As a reasonable practice, it is recommended to maintain the material supplier in order to try to reduce such variations.

to this kind of issues, since a perfect contact between the body and the cover must be reached. In traditional mechanizing, screws are used to tackle this problem, but a tiny air gap in 3D–printed

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designs is almost unavoidable. Besides this, there might be a bending of the top layer.

**Figure 9.** Insertion losses for WR75 waveguide sections (E-plane and H-plane).

**Figure 10.** PLA-printed pieces of the waveguide load before being covered with the graphite powder.

#### *3.1.3. Structure*

This last limitation is more specific for passive waveguide devices such as filters, diplexers, or couplers. Its effect in the horn antennas presented later is much less important.

Since the structures must be split into pieces to metallize them, two main possible divisions are considered: E-plane or H-plane (**Figure 8**) [13]. The former consists in splitting the structure into two identical halves, following the so-called E-plane approach (cutting in the plane where the E-field is maximum), whereas the latter separates the structure in body and cover. Trying to determine the best approach, some experiments were conducted.

**Figure 9** shows the insertion-loss level of 40-mm-long WR75 waveguide sections covered with the same paint. The H-plane structure presents higher losses. That may be explained by considering problems assembling the structure or misalignments. H-plane structures are more sensitive

**Figure 8.** (a) CAD model and (b) 3D printed E-plane and H-plane structures at left and right, respectively.

to this kind of issues, since a perfect contact between the body and the cover must be reached. In traditional mechanizing, screws are used to tackle this problem, but a tiny air gap in 3D–printed designs is almost unavoidable. Besides this, there might be a bending of the top layer.

**Figure 9.** Insertion losses for WR75 waveguide sections (E-plane and H-plane).

to another, and even between spools or colors. Consequently, the plastic may absorb the paint differently. The behavior with humidity or temperature variations may vary as well. As a reasonable practice, it is recommended to maintain the material supplier in order to try to reduce such variations.

This last limitation is more specific for passive waveguide devices such as filters, diplexers, or

Since the structures must be split into pieces to metallize them, two main possible divisions are considered: E-plane or H-plane (**Figure 8**) [13]. The former consists in splitting the structure into two identical halves, following the so-called E-plane approach (cutting in the plane where the E-field is maximum), whereas the latter separates the structure in body and cover.

**Figure 9** shows the insertion-loss level of 40-mm-long WR75 waveguide sections covered with the same paint. The H-plane structure presents higher losses. That may be explained by considering problems assembling the structure or misalignments. H-plane structures are more sensitive

couplers. Its effect in the horn antennas presented later is much less important.

102 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

Trying to determine the best approach, some experiments were conducted.

**Figure 8.** (a) CAD model and (b) 3D printed E-plane and H-plane structures at left and right, respectively.

*3.1.3. Structure*

**Figure 10.** PLA-printed pieces of the waveguide load before being covered with the graphite powder.

On the other hand, E-plane structures show lower losses since they do not depend that much on the assembling. This is due to the absence of currents along the cutting plane. Consequently, dividing the structure into two halves is the best way to proceed and that is how it has been done in this work.

#### **3.2. 3D printed devices: experimental results**

#### *3.2.1. Waveguide load*

The first waveguide structure to be characterized is a load for Ku-band. It is split in several pieces, printed in plastic, and subsequently covered in Graphite 33 [14], a graphite lacquer for conductive coatings which shows a good electrical conductivity due to the high level of pure and fine graphite powder. Both inner and outer surfaces are coated. The two stages of the manufacturing process appear in **Figure 10** and **Figure 11**. The former shows the different parts of the load before coating them with the graphite powder, whereas the measurement setup with the assembled load appears in the latter.

The matching response is shown in **Figure 12**, with a matching level better than 21 dB in the whole band. Such result is equivalent to some commercial short waveguide loads working in the same band [15].

#### *3.2.2. Waveguide section*

After the loads, a simple two-port device is tested: a straight waveguide section. Although some of them were tested to decide between the different paints or the most suitable way to cut the structure, the S-parameters of a 3D–printed 50-mm-long WR75 waveguide section are presented in **Figure 13**. Perhaps a remarkable fact is the lack of symmetry between S11 and S22, which is due to the inherent asymmetries associated to the 3D–printing process. However, the matching level agrees with previously obtained values, as well as the insertion losses in Ku band.

*3.2.3. Filter: low-pass, high-pass, band-pass*

**Figure 12.** Matching response of the 3D printed load.

*3.2.3.1. Low-pass filter*

low-cost 3D printing technology.

After having analyzed simple waveguide elements such as the load or the straight section, a very challenging passive device is addressed. Filters are key components in the microwave field and their main purpose is to allow a good transmission of desired signals while rejecting the unwanted ones. Depending on their response, they are classified in low-pass, high-pass, band-pass or bandstop filters. In this work, examples of the first three groups are implemented using 3D–printed rectangular waveguides. The results were first presented in [6, 7, 16, 17]. For these and all the other passive devices, the mode-matching technique [18] has been the design tool, whereas Microwave

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Studio CST [19] has allowed performing the simulations including manufacturing effects.

Low-pass filters implemented in waveguide technology are based on corrugated structures. The different impedances caused by the varying height of the sections lead to rejection at higher frequencies. In this case, an eight-order low-pass filter with four corrugations has been designed and presented in detail in [6, 17]. It is an E-plane structure. The working bandwidth is from 11.9 GHz to 12.2 GHz. Specifications include 28-dB matching level and rejection higher than 50 dB from 13.75 GHz to 14 GHz. These requirement levels and the frequency bands are typical of satellite telecommunication systems in Ku band and therefore very demanding for

The printed parts of the filter and the assembled structure are presented in **Figure 14**. The most important part of this device is the corrugation width. In this case, it is set to 2.5 mm considering

**Figure 11.** Measurement setup of the waveguide load.

**Figure 12.** Matching response of the 3D printed load.

#### *3.2.3. Filter: low-pass, high-pass, band-pass*

After having analyzed simple waveguide elements such as the load or the straight section, a very challenging passive device is addressed. Filters are key components in the microwave field and their main purpose is to allow a good transmission of desired signals while rejecting the unwanted ones. Depending on their response, they are classified in low-pass, high-pass, band-pass or bandstop filters. In this work, examples of the first three groups are implemented using 3D–printed rectangular waveguides. The results were first presented in [6, 7, 16, 17]. For these and all the other passive devices, the mode-matching technique [18] has been the design tool, whereas Microwave Studio CST [19] has allowed performing the simulations including manufacturing effects.

#### *3.2.3.1. Low-pass filter*

On the other hand, E-plane structures show lower losses since they do not depend that much on the assembling. This is due to the absence of currents along the cutting plane. Consequently, dividing the structure into two halves is the best way to proceed and that is how it has been

The first waveguide structure to be characterized is a load for Ku-band. It is split in several pieces, printed in plastic, and subsequently covered in Graphite 33 [14], a graphite lacquer for conductive coatings which shows a good electrical conductivity due to the high level of pure and fine graphite powder. Both inner and outer surfaces are coated. The two stages of the manufacturing process appear in **Figure 10** and **Figure 11**. The former shows the different parts of the load before coating them with the graphite powder, whereas the measurement

The matching response is shown in **Figure 12**, with a matching level better than 21 dB in the whole band. Such result is equivalent to some commercial short waveguide loads working in

After the loads, a simple two-port device is tested: a straight waveguide section. Although some of them were tested to decide between the different paints or the most suitable way to cut the structure, the S-parameters of a 3D–printed 50-mm-long WR75 waveguide section are presented in **Figure 13**. Perhaps a remarkable fact is the lack of symmetry between S11 and S22, which is due to the inherent asymmetries associated to the 3D–printing process. However, the matching

level agrees with previously obtained values, as well as the insertion losses in Ku band.

done in this work.

*3.2.1. Waveguide load*

the same band [15].

*3.2.2. Waveguide section*

**3.2. 3D printed devices: experimental results**

104 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

setup with the assembled load appears in the latter.

**Figure 11.** Measurement setup of the waveguide load.

Low-pass filters implemented in waveguide technology are based on corrugated structures. The different impedances caused by the varying height of the sections lead to rejection at higher frequencies. In this case, an eight-order low-pass filter with four corrugations has been designed and presented in detail in [6, 17]. It is an E-plane structure. The working bandwidth is from 11.9 GHz to 12.2 GHz. Specifications include 28-dB matching level and rejection higher than 50 dB from 13.75 GHz to 14 GHz. These requirement levels and the frequency bands are typical of satellite telecommunication systems in Ku band and therefore very demanding for low-cost 3D printing technology.

The printed parts of the filter and the assembled structure are presented in **Figure 14**. The most important part of this device is the corrugation width. In this case, it is set to 2.5 mm considering

**Figure 13.** (a) Reflection and (b) transmission S-parameters for a printed WR75 waveguide section.

the printer accuracy. Besides this, it has been possible to avoid input/output transformers, that is, the filter height corresponds to the standard WR75. Consequently, insertion losses have been reduced as much as possible. Finally, it is very important to reach a perfect alignment.

in terms of return-loss and rejection level are the same as in the low-pass filter but for opposite bands, that is, return loss of 28 dB between 13.75 GHz and 14 GHz and rejection greater than 50 dB between 11.9 GHz and 12.2 GHz. From a full-wave design perspective, the filter is an H-plane structure, yet an E-plane implementation has been followed, as shown in **Figure 17**. **Figure 18** shows the measured results. In this case, the filter itself is less sensitive than the previous one; therefore, a quite good agreement is expected and achieved. The matching level is

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The last filter was introduced in [7, 16]. It is a band-pass one, where selective frequency response is directly related to the resonance phenomenon [20]. The process involves high standing waves and electromagnetic field intensities, and, therefore, insertion losses are

The band-pass filter in **Figure 19** is a third-order Chebychev filter. The figure shows both the CAD layout and the fully assembled structure. Inductive irises are used to couple between cavities. It is centered in 12 GHz and the bandwidth is 600 MHz. Return losses were specified to be 23 dB.

even better than the usual one.

expected higher than in high-pass or low-pass filters.

**Figure 14.** (a) CAD pieces and (b) assembled low-pass filter.

*3.2.3.3. Band-pass filter*

The measurement setup is presented in **Figure 15** and the results are shown in **Figure 16**. The main characteristic of the measured response is that it is wider than the simulated one. In addition, the matching level does not reach the intended level, yet it agrees with the aforementioned limits of the technology and proves how unrealistic it is to try to get more than 22–25 dB return loss in Ku band with low-cost additive manufacturing.

The rejection response differs from the expected one as well. Since the corrugations are the parts causing the reflection, it may be inferred that some modifications should be introduced in future designs. For instance, wider corrugations would allow a better metallization process and therefore a potentially better response. However, it would cause a larger structure with more sections and more insertion losses, which are now around 1.8 dB.

#### *3.2.3.2. High-pass filter*

The next design was also included in [6, 17]. It is a high-pass filter based on a waveguide section under cutoff with stepped taper for matching at both input and output ports. The requirements

**Figure 14.** (a) CAD pieces and (b) assembled low-pass filter.

in terms of return-loss and rejection level are the same as in the low-pass filter but for opposite bands, that is, return loss of 28 dB between 13.75 GHz and 14 GHz and rejection greater than 50 dB between 11.9 GHz and 12.2 GHz. From a full-wave design perspective, the filter is an H-plane structure, yet an E-plane implementation has been followed, as shown in **Figure 17**.

**Figure 18** shows the measured results. In this case, the filter itself is less sensitive than the previous one; therefore, a quite good agreement is expected and achieved. The matching level is even better than the usual one.

#### *3.2.3.3. Band-pass filter*

the printer accuracy. Besides this, it has been possible to avoid input/output transformers, that is, the filter height corresponds to the standard WR75. Consequently, insertion losses have been reduced as much as possible. Finally, it is very important to reach a perfect alignment.

The measurement setup is presented in **Figure 15** and the results are shown in **Figure 16**. The main characteristic of the measured response is that it is wider than the simulated one. In addition, the matching level does not reach the intended level, yet it agrees with the aforementioned limits of the technology and proves how unrealistic it is to try to get more than

The rejection response differs from the expected one as well. Since the corrugations are the parts causing the reflection, it may be inferred that some modifications should be introduced in future designs. For instance, wider corrugations would allow a better metallization process and therefore a potentially better response. However, it would cause a larger structure with

The next design was also included in [6, 17]. It is a high-pass filter based on a waveguide section under cutoff with stepped taper for matching at both input and output ports. The requirements

22–25 dB return loss in Ku band with low-cost additive manufacturing.

**Figure 13.** (a) Reflection and (b) transmission S-parameters for a printed WR75 waveguide section.

106 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

more sections and more insertion losses, which are now around 1.8 dB.

*3.2.3.2. High-pass filter*

The last filter was introduced in [7, 16]. It is a band-pass one, where selective frequency response is directly related to the resonance phenomenon [20]. The process involves high standing waves and electromagnetic field intensities, and, therefore, insertion losses are expected higher than in high-pass or low-pass filters.

The band-pass filter in **Figure 19** is a third-order Chebychev filter. The figure shows both the CAD layout and the fully assembled structure. Inductive irises are used to couple between cavities. It is centered in 12 GHz and the bandwidth is 600 MHz. Return losses were specified to be 23 dB.

**Figure 15.** Measurement setup for the low-pass filter connected to the VNA.

*3.2.4. Diplexer*

munication systems.

**Figure 17.** (a) CAD pieces and (b) assembled high-pass filter.

A more complex device is now presented: a diplexer, which is also included in [7]. It is a passive microwave device formed by two filtering structures joined by a three-port junction. Its main application is the use of a single antenna for both transmission and reception in com-

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For this design, real satellite communications specifications are used. The passbands are 11.9– 12.2 GHz for reception and 13.75–14 GHz for transmission. Return losses and out-of-band rejection are included in the specifications. The former is around 22 dB, whereas the latter is required to be more than 50 dB in complementary bands. For such requirements, different implementations may be used: two band-pass filters or a low-pass combined with a high-pass filter. In this case, the second approach is used since it is the implementation with less insertion losses.

**Figure 16.** Measurement and simulation results for the low-pass filter.

**Figure 20** shows the measured response together with the simulation. The effective conductivity has been considered in the simulations. The matching value is reduced in the 3D– printed model, yet both responses are very close. It shows good agreement with the mean value for 3D–printed waveguides in Ku band. Besides, insertion losses are 1.3 dB at the central frequency.

**Figure 17.** (a) CAD pieces and (b) assembled high-pass filter.

#### *3.2.4. Diplexer*

**Figure 20** shows the measured response together with the simulation. The effective conductivity has been considered in the simulations. The matching value is reduced in the 3D– printed model, yet both responses are very close. It shows good agreement with the mean value for 3D–printed waveguides in Ku band. Besides, insertion losses are 1.3 dB at the

**Figure 16.** Measurement and simulation results for the low-pass filter.

**Figure 15.** Measurement setup for the low-pass filter connected to the VNA.

108 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

central frequency.

A more complex device is now presented: a diplexer, which is also included in [7]. It is a passive microwave device formed by two filtering structures joined by a three-port junction. Its main application is the use of a single antenna for both transmission and reception in communication systems.

For this design, real satellite communications specifications are used. The passbands are 11.9– 12.2 GHz for reception and 13.75–14 GHz for transmission. Return losses and out-of-band rejection are included in the specifications. The former is around 22 dB, whereas the latter is required to be more than 50 dB in complementary bands. For such requirements, different implementations may be used: two band-pass filters or a low-pass combined with a high-pass filter. In this case, the second approach is used since it is the implementation with less insertion losses.

**Figure 18.** Measurement and simulation results for the high-pass filter.

However, the low-pass filter presents some challenges, as mentioned earlier. Again, the corrugations are the most sensitive part. Consequently, they have been made wider than in traditionally manufactured filters, also to have them painted more easily. Corrugations are 4.5-mm wide, which implies more sections to achieve the same electrical response.

transmission lines connected by two secondary (branch) lines. They present a narrow band-

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A 3-dB branch-line coupler working in Ku band has been designed and the results presented in [7]. Aiming to achieve a broad bandwidth, five branches have been designed. Isolation and matching levels are intended to be 25 dB in the working band. **Figure 23** shows the designed device from the simulation model to the measurement setup. Two double bends have been

The experimental response is shown in **Figure 24**. In terms of matching and isolation, the results are very satisfactory. The former is greater than 20 dB between 11 GHz and 14 GHz, although if the bandwidth from 11.5 GHz to 14 GHz is considered the matching value results to be greater than 25 dB. The isolation reached is very high as well, considering the limitations

The response of the printed branch-line coupler in terms of power coupled to each output port is evaluated on the enhanced part of **Figure 24**. The measurements are compared to a full-wave simulation including losses. They are lower in the transmitted port (P2) than in

width, although it can be improved by cascading several sections.

**Figure 19.** (a) CAD model and (b) assembled band-pass filter.

added to the coupler in order to attach the standard WR75 flanges.

of 3D printing. From 11 GHz to 14 GHz, it is better than 24 dB.

**Figure 21** shows the development of the diplexer: (a) the layout, (b) the CAD model, (c) one printed half, and (d) the device prepared to be measured. To join the filters, a three-port junction is designed. Moreover, a double bend is included to allow the waveguide flanges to be attached. Finally, a fastening piece has been printed to keep the two pieces close and avoid any possible air gaps.

The diplexer response is presented in **Figure 22** compared to a simulation with losses. Insertion losses are up to 3.5 dB in the passband. The matching level in both channels agrees with the achievable levels with low-cost 3D printing: approximately 20 dB. It is important to highlight that both responses have moved to lower frequencies as a result of possible printing inaccuracies.

#### *3.2.5. Branch-line coupler*

Directional couplers are four-port devices in which the power available at the input port is distributed between two output ports [20]. Perhaps the most common ones are the 3-dB couplers, widely used as input/output in balanced amplifier circuits, in beam-forming networks, as power dividers, and so on. In particular, branch-line couplers consist in two shunt

**Figure 19.** (a) CAD model and (b) assembled band-pass filter.

However, the low-pass filter presents some challenges, as mentioned earlier. Again, the corrugations are the most sensitive part. Consequently, they have been made wider than in traditionally manufactured filters, also to have them painted more easily. Corrugations are 4.5-mm

**Figure 21** shows the development of the diplexer: (a) the layout, (b) the CAD model, (c) one printed half, and (d) the device prepared to be measured. To join the filters, a three-port junction is designed. Moreover, a double bend is included to allow the waveguide flanges to be attached. Finally, a fas-

The diplexer response is presented in **Figure 22** compared to a simulation with losses. Insertion losses are up to 3.5 dB in the passband. The matching level in both channels agrees with the achievable levels with low-cost 3D printing: approximately 20 dB. It is important to highlight that both responses have moved to lower frequencies as a result of possible printing inaccuracies.

Directional couplers are four-port devices in which the power available at the input port is distributed between two output ports [20]. Perhaps the most common ones are the 3-dB couplers, widely used as input/output in balanced amplifier circuits, in beam-forming networks, as power dividers, and so on. In particular, branch-line couplers consist in two shunt

tening piece has been printed to keep the two pieces close and avoid any possible air gaps.

wide, which implies more sections to achieve the same electrical response.

**Figure 18.** Measurement and simulation results for the high-pass filter.

110 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

*3.2.5. Branch-line coupler*

transmission lines connected by two secondary (branch) lines. They present a narrow bandwidth, although it can be improved by cascading several sections.

A 3-dB branch-line coupler working in Ku band has been designed and the results presented in [7]. Aiming to achieve a broad bandwidth, five branches have been designed. Isolation and matching levels are intended to be 25 dB in the working band. **Figure 23** shows the designed device from the simulation model to the measurement setup. Two double bends have been added to the coupler in order to attach the standard WR75 flanges.

The experimental response is shown in **Figure 24**. In terms of matching and isolation, the results are very satisfactory. The former is greater than 20 dB between 11 GHz and 14 GHz, although if the bandwidth from 11.5 GHz to 14 GHz is considered the matching value results to be greater than 25 dB. The isolation reached is very high as well, considering the limitations of 3D printing. From 11 GHz to 14 GHz, it is better than 24 dB.

The response of the printed branch-line coupler in terms of power coupled to each output port is evaluated on the enhanced part of **Figure 24**. The measurements are compared to a full-wave simulation including losses. They are lower in the transmitted port (P2) than in

**Figure 20.** Measurement and simulation results for the band-pass filter.

the coupled one (P3), although the former has a double bend attached. It may be explained by considering that the coupled signal passes through the branches on its way to the output port. Moreover, misalignments are likely to happen, particularly in the small posts among the branches. They are the smallest part of the design, as well as the most sensitive one, as it happened with the corrugations in the low-pass filter.

and mass-produced metal horns. At the same time, they are robust enough and their smooth surface makes it feasible to print them as single structures, instead of splitting them into two parts as it occurred with the previous waveguide devices. It must be highlighted that the

**Figure 21.** (a) Diplexer layout, (b) half CAD model, (c) one printed half, (d) the device metallized and assembled, and

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The time required to print each horn antenna is approximately 2 hours, notably less than a traditional manufacturing process. Finally, the conformed profile of the second horn would have been very difficult to achieve by traditional means, and it would have required the use

Both horn antennas have been characterized in terms of matching level, gain, and radiation patterns. **Figure 26** shows the experimental setup for the pyramidal horn in the network analyzer and the anechoic chamber. The theoretical responses have been obtained with CST. Moreover, the measured gain of the antennas has been compared not only to the simulated response but also to the commercial model supplied by Flann working in the same

printed structures also include the waveguide input (the standard WR75 flange).

of molds. At the end, the total cost of each antenna is approximately 15 \$.

band.

(e) measurement setup.

#### *3.2.6. Horn antennas*

The two horn antennas [21] presented in this work fulfill standard horn specifications, according to available commercial models working in the same band, like the Flann horn in [22]. Their gain reaches 15 dBi at the center of the Ku band. The first one is a classical pyramidal horn, whereas the second one has a rectangular aperture and a conformed profile. Their input corresponds to the WR75 rectangular waveguide. The return-loss level for both antennas has been optimized to remain below 20 dB between 10 GHz and 15 GHz. Measurements show a remarkably good agreement with theoretical values, which makes the horns perfectly useful for real communication applications, for instance, indoor systems.

The process is presented in **Figure 25**, together with the CAD design of one horn. The antennas are monolayer structures, that is, only one layer of plastic has been used. Due to this fact, the antennas are extraordinary light—less than 15 gr.—contrary to the weight of traditional Additive Manufacturing of 3D Printed Microwave Passive Components http://dx.doi.org/10.5772/intechopen.74275 113

**Figure 21.** (a) Diplexer layout, (b) half CAD model, (c) one printed half, (d) the device metallized and assembled, and (e) measurement setup.

the coupled one (P3), although the former has a double bend attached. It may be explained by considering that the coupled signal passes through the branches on its way to the output port. Moreover, misalignments are likely to happen, particularly in the small posts among the branches. They are the smallest part of the design, as well as the most sensitive one, as it hap-

The two horn antennas [21] presented in this work fulfill standard horn specifications, according to available commercial models working in the same band, like the Flann horn in [22]. Their gain reaches 15 dBi at the center of the Ku band. The first one is a classical pyramidal horn, whereas the second one has a rectangular aperture and a conformed profile. Their input corresponds to the WR75 rectangular waveguide. The return-loss level for both antennas has been optimized to remain below 20 dB between 10 GHz and 15 GHz. Measurements show a remarkably good agreement with theoretical values, which makes the horns perfectly useful

The process is presented in **Figure 25**, together with the CAD design of one horn. The antennas are monolayer structures, that is, only one layer of plastic has been used. Due to this fact, the antennas are extraordinary light—less than 15 gr.—contrary to the weight of traditional

pened with the corrugations in the low-pass filter.

**Figure 20.** Measurement and simulation results for the band-pass filter.

112 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

for real communication applications, for instance, indoor systems.

*3.2.6. Horn antennas*

and mass-produced metal horns. At the same time, they are robust enough and their smooth surface makes it feasible to print them as single structures, instead of splitting them into two parts as it occurred with the previous waveguide devices. It must be highlighted that the printed structures also include the waveguide input (the standard WR75 flange).

The time required to print each horn antenna is approximately 2 hours, notably less than a traditional manufacturing process. Finally, the conformed profile of the second horn would have been very difficult to achieve by traditional means, and it would have required the use of molds. At the end, the total cost of each antenna is approximately 15 \$.

Both horn antennas have been characterized in terms of matching level, gain, and radiation patterns. **Figure 26** shows the experimental setup for the pyramidal horn in the network analyzer and the anechoic chamber. The theoretical responses have been obtained with CST. Moreover, the measured gain of the antennas has been compared not only to the simulated response but also to the commercial model supplied by Flann working in the same band.

*3.2.6.3. Antenna gain*

The gain over broadband has also been measured for both antennas. The pyramidal horn gain is presented in **Figure 31** compared with the simulated one and the commercial pyramidal horn provided by Flann [22]. It can be seen that the gain of the 3D–printed horn is not as high

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**Figure 23.** (a) Layout, (b) printed half, and (c) assembled and metallized branch-line coupler.

**Figure 22.** Measurement and simulation results for the diplexer.

#### *3.2.6.1. Matching level*

The horn antennas were designed with a return-loss level better than 20 dB between 10 GHz and 15 GHz. The measured responses and the simulated ones (σ = 5·10<sup>4</sup> S/m) are shown in **Figures 27** and **28**. Despite a return-loss level slightly worse than 20 dB between 10 GHz and 10.5 GHz for the conformed horn, the response of the reflection coefficient follows the expected one. However, a small ripple can be observed in both antennas. It is explained by considering multiple reflections due to the inherent roughness and small discontinuities on the walls.

#### *3.2.6.2. Radiation pattern*

The normalized radiation patterns in the E and H-plane—elevation and azimuth, respectively—have been obtained at 12.5 GHz for both antennas. The pattern corresponding to the classical pyramidal horn is shown in **Figure 29** along with the simulated co-polar response. The measured response follows the theoretical one very accurately, with a difference in the 3-dB beamwidth of only 1° (**Table 1**). With respect to the cross-polarization level, its measurement is limited by the anechoic chamber used in the horn characterization. In this case, its level is always 30 dB under the main lobe in both planes.

The conformed-profile horn shows a very good response as well, as presented in **Figure 30**. The radiation pattern matches the simulated one accurately and the 3-dB beamwidth barely differs 5° in the worst-case scenario (**Table 1**). Finally, the cross-polarization level remains more than 30 dB under the main lobe in both planes.

#### *3.2.6.3. Antenna gain*

*3.2.6.1. Matching level*

*3.2.6.2. Radiation pattern*

level is always 30 dB under the main lobe in both planes.

**Figure 22.** Measurement and simulation results for the diplexer.

114 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

more than 30 dB under the main lobe in both planes.

the walls.

The horn antennas were designed with a return-loss level better than 20 dB between 10 GHz and 15 GHz. The measured responses and the simulated ones (σ = 5·10<sup>4</sup> S/m) are shown in **Figures 27** and **28**. Despite a return-loss level slightly worse than 20 dB between 10 GHz and 10.5 GHz for the conformed horn, the response of the reflection coefficient follows the expected one. However, a small ripple can be observed in both antennas. It is explained by considering multiple reflections due to the inherent roughness and small discontinuities on

The normalized radiation patterns in the E and H-plane—elevation and azimuth, respectively—have been obtained at 12.5 GHz for both antennas. The pattern corresponding to the classical pyramidal horn is shown in **Figure 29** along with the simulated co-polar response. The measured response follows the theoretical one very accurately, with a difference in the 3-dB beamwidth of only 1° (**Table 1**). With respect to the cross-polarization level, its measurement is limited by the anechoic chamber used in the horn characterization. In this case, its

The conformed-profile horn shows a very good response as well, as presented in **Figure 30**. The radiation pattern matches the simulated one accurately and the 3-dB beamwidth barely differs 5° in the worst-case scenario (**Table 1**). Finally, the cross-polarization level remains The gain over broadband has also been measured for both antennas. The pyramidal horn gain is presented in **Figure 31** compared with the simulated one and the commercial pyramidal horn provided by Flann [22]. It can be seen that the gain of the 3D–printed horn is not as high

**Figure 23.** (a) Layout, (b) printed half, and (c) assembled and metallized branch-line coupler.

**Figure 25.** (a) CAD model of the pyramidal horn. (b) Horn antenna being printed with the low-cost PLA 3D

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**Figure 26.** Measurement arrangement for the pyramidal horn in (a) the network analyzer and (b) the anechoic

printer.

chamber.

**Figure 24.** Measurement and simulation results for the branch-line coupler. To the right, enhanced view of the transmission parameters.

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**Figure 25.** (a) CAD model of the pyramidal horn. (b) Horn antenna being printed with the low-cost PLA 3D printer.

**Figure 26.** Measurement arrangement for the pyramidal horn in (a) the network analyzer and (b) the anechoic chamber.

**Figure 24.** Measurement and simulation results for the branch-line coupler. To the right, enhanced view of the trans-

116 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

mission parameters.

**Figure 27.** Return-loss level over Ku-band for the classical pyramidal horn. Top right inset: Printed and metallized structure.

as the one of the commercial model, with differences up to 1.5 dB, especially in the lower band. There are discrepancies with the simulation too, caused by the limitations of the manufacturing process. However, the results are satisfactory, considering the intrinsic variability of the metallization and the substantial difference in terms of cost between the Flann horn and

**Figure 29.** Normalized radiation pattern of the pyramidal horn at frequency 12.5 GHz in (a) E-plane and (b) H-plane.

For the conformed-profile horn, the measured gain matches the simulated response with a remarkable accuracy, as shown in **Figure 32**. It presents a good behavior when compared with the Flann model as well, although some variations caused by manufacturing issues exist.

**E-plane H-plane**

Pyramidal horn 34° 33° 31.4° 30.7° Conformed horn 24.8° 30.2° 28.6° 31.1°

**Measured Simulated Measured Simulated**

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the printed one.

However, they do not exceed 1.35 dB.

**Table 1.** Simulated and measured 3-dB beamwidth.

**Figure 28.** Return-loss level over Ku-band for the conformed-profile horn. Top left inset: Printed and metallized structure.

**Figure 29.** Normalized radiation pattern of the pyramidal horn at frequency 12.5 GHz in (a) E-plane and (b) H-plane.

as the one of the commercial model, with differences up to 1.5 dB, especially in the lower band. There are discrepancies with the simulation too, caused by the limitations of the manufacturing process. However, the results are satisfactory, considering the intrinsic variability of the metallization and the substantial difference in terms of cost between the Flann horn and the printed one.

For the conformed-profile horn, the measured gain matches the simulated response with a remarkable accuracy, as shown in **Figure 32**. It presents a good behavior when compared with the Flann model as well, although some variations caused by manufacturing issues exist. However, they do not exceed 1.35 dB.


**Table 1.** Simulated and measured 3-dB beamwidth.

**Figure 27.** Return-loss level over Ku-band for the classical pyramidal horn. Top right inset: Printed and metallized

118 Emerging Microwave Technologies in Industrial, Agricultural, Medical and Food Processing

**Figure 28.** Return-loss level over Ku-band for the conformed-profile horn. Top left inset: Printed and metallized structure.

structure.

**Figure 30.** Normalized radiation pattern of the conformed-profile horn at frequency 12.5 GHz in (a) E-plane and (b) H-plane.

**4. Conclusions**

Flann.

authors' knowledge.

cially for institutions without extraordinary resources.

response, yet the discrepancies have also been explained.

Although benefits of additive manufacturing are well known to the engineer community, its real application to a particular field needs a detailed analysis, even more when considering a low-cost perspective. With the work presented throughout this chapter, it can be concluded that low-cost 3D printing presents great advantages for the microwave engineer. The promptness to print models of great quality or to implement traditionally unfeasible geometries may be used for two main applications: prototyping and manufacturing of functional devices. Both applications may also increase the quality of microwave engineering education, espe-

**Figure 32.** Conformed-profile horn gain over broadband compared with simulation and commercial horn antenna from

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In order to successfully address a low-cost 3D printing process in the microwave field, it is necessary to first take into account several key aspects: to set the maximum achievable working frequency, the best way to perform a metallization process, and the most suitable structure segmentation. This work has described all of them, emphasizing the best solutions as per

In addition, several prototypes of state-of-the art designs have been presented, together with real passive waveguide devices—individual filters, a diplexer, and a branch-line coupler—and two horn antennas. Experimental results have been discussed to evaluate their use in real-world applications. In general terms, the results agree quite well with the expected

**Figure 31.** Pyramidal horn gain over broadband compared with simulation and commercial horn antenna from Flann.

**Figure 32.** Conformed-profile horn gain over broadband compared with simulation and commercial horn antenna from Flann.
