**4. Interferometer based on band-stop filter for IFM**

The IFMS presented now is based on band-stop filter and is shown in Fig. 8. The advantage of using the new architecture is that one has in each channel only multi band-stop filters in‐ stead of delay lines and power splitter, as one finds in classical IFMS.

Each word is assigned to only one frequency sub-band to generate a one-step binary code. The response of each multi band-stop filter should be like the one shown in Fig. 9 (a) with discriminators 0, 1, 2, 3 and 4. The discriminator 0 provides the least-significant bit (LSB) and the discriminator 4 provides the most-significant bit (MSB). The form of these responses is suitable to implement the 1 bit A/D converters. Here, let us attribute value 1 if the inser‐ tion loss response for the multi band-stop filter is greater than 5 dB, and value 0 for the op‐ posite case. Fig. 9(b) shows the wave form of each 1 bit A/D converter output. According to this example the waveforms at the 1 bit A/D converter outputs are shown in Fig. 9(c). As seen in Fig. 4, this subsystem has its operating band from 2 to 4 GHz, which was divided into 32 sub-bands. Therefore, the resolution obtained was fR = 62.5 MHz.

Fig. 12(b) shows the frequency response obtained at ideal coupling distance between them. These distances are chosen to obtain the insertion loss greater than 10 dB over rejection band and also to get this band as large as required. One notices that the coupling between nonadjacent resonators is almost zero. This happens because their resonance frequencies are not very close and the distance between them is large enough. Therefore, the insertion of a new

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**Figure 9.** Responses for the IFMS from Fig. 8: (a) desired |S21|, (b) A/D converters output, and (c) generated code.

ed in the full chapter.

A model of two coupled resonators has been developed by the authors and will be present‐

resonator does not change the position of the others already inserted.

**Figure 8.** Architecture of an instantaneous frequency measurement subsystem (IFMS) using band-stop filters.

#### **4.1. Multi band-stop filter design and measurement**

Rectangular microstrip open loop resonators were chosen to design every discriminator of a five bit IFMS. Frequency response of those resonators presents a narrow rejection band and wide pass band [5] with first spurious out of the working band. Fig. 10(a) shows the top view of a resonator with resonance frequency at 1.9375 GHz. One can see in Fig. 10(b) that the first spurious occurs at 6.140 GHz. Still in this section, it will be shown how this re‐ sponse makes possible the fabrication of a wideband discriminator.

That resonator is placed near to a 50 Ω microstrip transmission line, which was de‐ signed with aid of quasi-static analysis and quasi-TEM approximation [8]-[9]. Fig. 11 shows the resonance frequency adjusted by the length l1 + l2 + l3+ l4 of the resonator, which must be approximately half wavelength long [8]. Additionally, there is a coupling gap g given by l2 - l3 - l4. Moreover, the coupling distance between the resonator and the main transmission line affects this resonance frequency. This distance also affects the bandwidth of the resonator [8].

Despite the narrow band of the isolated resonators, wide rejection bands are created from coupled arrays. Fig. 12(a) presents 3 sketches of one, two and three resonators, whose reso‐ nant frequencies are 2.02, 2.07 and 2.12 GHz, respectively. The line width for the resonators is fixed to be 0.5 mm along this chapter. The ideal coupling distance between resonators is obtained varying di,j using EM full wave software.

Fig. 12(b) shows the frequency response obtained at ideal coupling distance between them. These distances are chosen to obtain the insertion loss greater than 10 dB over rejection band and also to get this band as large as required. One notices that the coupling between nonadjacent resonators is almost zero. This happens because their resonance frequencies are not very close and the distance between them is large enough. Therefore, the insertion of a new resonator does not change the position of the others already inserted.

tion loss response for the multi band-stop filter is greater than 5 dB, and value 0 for the op‐ posite case. Fig. 9(b) shows the wave form of each 1 bit A/D converter output. According to this example the waveforms at the 1 bit A/D converter outputs are shown in Fig. 9(c). As seen in Fig. 4, this subsystem has its operating band from 2 to 4 GHz, which was divided

**Figure 8.** Architecture of an instantaneous frequency measurement subsystem (IFMS) using band-stop filters.

Rectangular microstrip open loop resonators were chosen to design every discriminator of a five bit IFMS. Frequency response of those resonators presents a narrow rejection band and wide pass band [5] with first spurious out of the working band. Fig. 10(a) shows the top view of a resonator with resonance frequency at 1.9375 GHz. One can see in Fig. 10(b) that the first spurious occurs at 6.140 GHz. Still in this section, it will be shown how this re‐

That resonator is placed near to a 50 Ω microstrip transmission line, which was de‐ signed with aid of quasi-static analysis and quasi-TEM approximation [8]-[9]. Fig. 11 shows the resonance frequency adjusted by the length l1 + l2 + l3+ l4 of the resonator, which must be approximately half wavelength long [8]. Additionally, there is a coupling gap g given by l2 - l3 - l4. Moreover, the coupling distance between the resonator and the main transmission line affects this resonance frequency. This distance also affects the

Despite the narrow band of the isolated resonators, wide rejection bands are created from coupled arrays. Fig. 12(a) presents 3 sketches of one, two and three resonators, whose reso‐ nant frequencies are 2.02, 2.07 and 2.12 GHz, respectively. The line width for the resonators is fixed to be 0.5 mm along this chapter. The ideal coupling distance between resonators is

**4.1. Multi band-stop filter design and measurement**

bandwidth of the resonator [8].

obtained varying di,j using EM full wave software.

sponse makes possible the fabrication of a wideband discriminator.

into 32 sub-bands. Therefore, the resolution obtained was fR = 62.5 MHz.

290 Radio Frequency Identification from System to Applications

**Figure 9.** Responses for the IFMS from Fig. 8: (a) desired |S21|, (b) A/D converters output, and (c) generated code.

A model of two coupled resonators has been developed by the authors and will be present‐ ed in the full chapter.

with its numbered resonators. The device is designed on a RT6010.2 substrate of relative die‐ lectric constant εr = 10.2 and thickness h = 1.27 mm. The 50 Ω transmission line width is 1.2 mm. The gap of every resonator and the distance between the main transmission line and the resonators are kept 0.1 mm for whole structure. Table I shows the coupling distances be‐

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**Figure 12.** (a) The open loop resonator arrays. The scale has been enhanced for a better comprehension of the devi‐

Still in Fig. 13(a) one sees four groups of resonators, whose frequency responses and A/D converter outputs are shown in Fig. 15(b). Looking carefully their correlation, Group 1 gives

ces, and (b) frequency response of 1, 2, and 3 resonators.

tween the resonators for this device.

**Figure 10.** (a) Physical structure of a resonator with resonance frequency at 1.9375 GHz, and (b) frequency response of the resonator over a wideband.

**Figure 11.** Open loop resonator.

As the desired insertion loss of the discriminator 1 is shown in Fig. 9(a), there must be four rejection bands, where the first one is from 2.125 GHz to 2.375 GHz, regarding the chosen operating band. The resonators are arranged one by one. Fig.13 (a) shows this discriminator with its numbered resonators. The device is designed on a RT6010.2 substrate of relative die‐ lectric constant εr = 10.2 and thickness h = 1.27 mm. The 50 Ω transmission line width is 1.2 mm. The gap of every resonator and the distance between the main transmission line and the resonators are kept 0.1 mm for whole structure. Table I shows the coupling distances be‐ tween the resonators for this device.

**Figure 10.** (a) Physical structure of a resonator with resonance frequency at 1.9375 GHz, and (b) frequency response

As the desired insertion loss of the discriminator 1 is shown in Fig. 9(a), there must be four rejection bands, where the first one is from 2.125 GHz to 2.375 GHz, regarding the chosen operating band. The resonators are arranged one by one. Fig.13 (a) shows this discriminator

of the resonator over a wideband.

292 Radio Frequency Identification from System to Applications

**Figure 11.** Open loop resonator.

**Figure 12.** (a) The open loop resonator arrays. The scale has been enhanced for a better comprehension of the devi‐ ces, and (b) frequency response of 1, 2, and 3 resonators.

Still in Fig. 13(a) one sees four groups of resonators, whose frequency responses and A/D converter outputs are shown in Fig. 15(b). Looking carefully their correlation, Group 1 gives the rejection band over 2 GHz; Group 2 gives the rejection band over 2.5 GHz, and so on. Fig. 13(b) presents the simulated results of the discriminator 1, which agree with the results shown in Fig. 9. One can see the insertion loss level is greater than 10 dB over all rejection bands, and is less than 5 dB over the pass bands. The output A/D converter should generate level zero for |S21| < - 5 dB and level 1 for |S21| > - 5 dB. Concerning all the involved di,j, the dimensions of this discriminator are 3 cm wide and 15 cm long. Following the same proce‐ dure, the others discriminators are projected, where new resonators configurations will give new desired rejection bands.

**Coupling distance between "i" and "j" resonators (mm)** *d1,2*= 0.6 *d13,14* = 1.4 *d2,3* = 0.8 *d14,15* = 1.6 *d3,4* = 0.5 *d15,16* = 1.3 *d4,5* = 0.3 *d16,17* = 0.7 *d5,6* = 0.2 *d17,18* = 0.4 *d7,8* = 0.6 *d19,20* = 1.3 *d8,9* = 1.2 *d20,21* = 1.4 *d9,10* = 0.4 *d21,22* = 1.6 *d10,11* = 1.1 *d22,23* = 1.2 *d11,12* = 1.1 *d23,24* = 1.1

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The Fig. 14(a)-(e) presents all the projected IFMS discriminators from Fig. 8, having between 23 and 25 resonators. The number of resonators depends on the desired rejection bands. Following the same principle, each group gives only one rejection band, so that discriminators with eight groups have eight rejection bands, as shown in Fig. 14(e). The others, without any specified group, have only one as shown in Fig. 14 (a) and (b). Fig. 15 shows that the simulated and meas‐

**Figure 14.** Bandstop filters for implementation of the: (a) discriminator 4 – MSB, (b) discriminator 3, (c) discriminator

2, (d) discriminator 1, and (e) discriminator 0 – LSB.

ured results of the five discriminators are in reasonable agreement with each other.

**Table 1.** Coupling Distances

**Figure 13.** (a) Layout of the discriminator 1, and (b) frequency response of the discriminator 1, and the output of the 1-bit A/D converter; 250 MHz for each rejected band.


**Table 1.** Coupling Distances

the rejection band over 2 GHz; Group 2 gives the rejection band over 2.5 GHz, and so on. Fig. 13(b) presents the simulated results of the discriminator 1, which agree with the results shown in Fig. 9. One can see the insertion loss level is greater than 10 dB over all rejection bands, and is less than 5 dB over the pass bands. The output A/D converter should generate level zero for |S21| < - 5 dB and level 1 for |S21| > - 5 dB. Concerning all the involved di,j, the dimensions of this discriminator are 3 cm wide and 15 cm long. Following the same proce‐ dure, the others discriminators are projected, where new resonators configurations will give

**Figure 13.** (a) Layout of the discriminator 1, and (b) frequency response of the discriminator 1, and the output of the

1-bit A/D converter; 250 MHz for each rejected band.

new desired rejection bands.

294 Radio Frequency Identification from System to Applications

The Fig. 14(a)-(e) presents all the projected IFMS discriminators from Fig. 8, having between 23 and 25 resonators. The number of resonators depends on the desired rejection bands. Following the same principle, each group gives only one rejection band, so that discriminators with eight groups have eight rejection bands, as shown in Fig. 14(e). The others, without any specified group, have only one as shown in Fig. 14 (a) and (b). Fig. 15 shows that the simulated and meas‐ ured results of the five discriminators are in reasonable agreement with each other.

**Figure 14.** Bandstop filters for implementation of the: (a) discriminator 4 – MSB, (b) discriminator 3, (c) discriminator 2, (d) discriminator 1, and (e) discriminator 0 – LSB.

dielectric losses. When designing an RFM it is important to decide which type of technology is adequate for a given application in terms of detection speed, power consumption and de‐

> Reconfigurable phase shifter

**Figure 16.** Architecture of a reconfigurable frequency measurement subsystem (RFM) based on phase shifters.

Device size will be mainly determined by the type of technology used to implement the subsystem; the most compact designs can be achieved monolithically, by having the components integrated into a single chip. A monolithic design can include all solid state, MEMS and ferroelectric implementations. Hybrid integrations use microwave laminates or substrates and tuning elements, these include solid state, MEMS and ferroelectric sur‐ face mountable components that can be embedded into the design. Hybrid integrations normally involve much larger circuit size compared to the monolithic counterpart, how‐ ever these components normally involve low cost and simple manufacturing and proto‐

The most reliable technology is the solid state transistor and the ferroelectric films, fol‐ lowed by the PIN and varactor diode ending with the MEMS components. MEMS pack‐ aging can improve device reliability by avoiding contamination or humidity of the movable parts of a switch or varactor. The objective of an RFM is to reduce the size of fixed IFMs by designing branches that can produce more than one bit in the identifica‐ tion subsystem. Size reduction is the main advantage of an RFM over a fixed IFM. A disadvantage over fixed IFMs is that there will be a switching time for the device, so the

This chapter presented two kinds of interferometers for IFM applications, the first type was a Coplanar Intedigital Interferometer and the second one was based on Multi band-stop fil‐ ters, which can substitute the interferometers in the IFM Architecture. For the first case, co‐ planar strips interdigital delay lines were fabricated, simulated and measured at a frequency

Voltage control

Fixed phase reference

> Vin GN D Vr ef D 1 D 4 Señal HAB . Conver t idor A/ D

Amplifier

Detector

A/D converter

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Output (multiple bit)

vice size.

Input signal

typing techniques.

frequency measurement is not instantaneous.

**6. Final considerations**

Limiting Amplifier

**Figure 15.** Frequency response of the: (A) Discriminator 4 – MSB, (B) Discriminator 3, (C) Discriminator 2, (D) Discrimi‐ nator 1, and (E) Discriminator 0 – LSB.

#### **5. Reconfigurable Frequency Measurement (RFM) designs**

Fixed IFM designs like the ones discussed in section IV have the advantage of providing in‐ stantaneous frequency identification while reconfigurable designs should do a sweep but are very compact in size, making them suitable for portable and handheld systems. RFMs include tuning elements [15] embedded in the designs to produce multibit frequency identi‐ fication using reconfigurable measurement branches.

An example of RFM architecture is shown in Fig. 16, this design includes a reconfigurable phase shifter used to produce more than one bit. The number of bits will depend on the amount of phase shifts produced by the reconfigurable design; each phase shift will corre‐ spond to a specific control voltage in the case of varactors, otherwise switches will be in "on" or "off" state to produce the different phase shifts. The other components shown in Fig. 16 operate in a similar way to the ones exposed in section IV. The RFM can also include reconfigurable bandstop filters [16] instead of the phase shifter to produce a branch that can produce more than one bit as an alternative design.

The switching speed of the tuning elements used in the reconfigurable phase shifter design will mainly determine the detection speed of the subsystem. Solid state components like PIN, varactor diodes, transistors and the use of ferroelectric materials will provide high tun‐ ing speeds, (10-6 seconds for the PIN and varactor diodes, 10-9 seconds for transistors and 10-10 seconds for the ferroelectric varactors) while the Micro Electromechanical Systems (MEMS) counterpart will provide slower tuning speeds (10-5 seconds) but with the advant‐ age of low power consumption compared with the solid state components. The use of ferro‐ electric materials results in high tuning speeds with the drawback of having generally high dielectric losses. When designing an RFM it is important to decide which type of technology is adequate for a given application in terms of detection speed, power consumption and de‐ vice size.

**Figure 16.** Architecture of a reconfigurable frequency measurement subsystem (RFM) based on phase shifters.

Device size will be mainly determined by the type of technology used to implement the subsystem; the most compact designs can be achieved monolithically, by having the components integrated into a single chip. A monolithic design can include all solid state, MEMS and ferroelectric implementations. Hybrid integrations use microwave laminates or substrates and tuning elements, these include solid state, MEMS and ferroelectric sur‐ face mountable components that can be embedded into the design. Hybrid integrations normally involve much larger circuit size compared to the monolithic counterpart, how‐ ever these components normally involve low cost and simple manufacturing and proto‐ typing techniques.

The most reliable technology is the solid state transistor and the ferroelectric films, fol‐ lowed by the PIN and varactor diode ending with the MEMS components. MEMS pack‐ aging can improve device reliability by avoiding contamination or humidity of the movable parts of a switch or varactor. The objective of an RFM is to reduce the size of fixed IFMs by designing branches that can produce more than one bit in the identifica‐ tion subsystem. Size reduction is the main advantage of an RFM over a fixed IFM. A disadvantage over fixed IFMs is that there will be a switching time for the device, so the frequency measurement is not instantaneous.
