**UWB Printed Antennas**

[32] Kong, L.-B., Zhong, S.-S., & Sun, Z., "Broadband microstrip element design of a DBDP shared-aperture SAR array", Microwave and Optical Technology Letters, Jan.

(2012). , 54(1), 133-136.

122 Advancement in Microstrip Antennas with Recent Applications

**Chapter 6**

**UWB Antennas for Wireless Applications**

what is currently used in different radio communica-tion systems [2].

**1.1. Different UWB Antenna Designs**

Currently, there is an increased interest in ultra-wideband (UWB) technology for use in sev‐ eral present and future applications. UWB technology received a major boost especially in 2002 since the US Federal Communication Commission (FCC) permitted the authorization of using the unlicensed frequency band starting from 3.1 to 10.6 GHz for commercial com‐ munication applications [1]. Although existing third-generation (3G) communication tech‐ nology can provide us with many wide services such as fast internet access, video telephony, enhanced video/music download as well as digital voice services, UWB –as a new technology– is very promising for many reasons. The FCC allocated an absolute band‐ width up to 7.5 GHz which is about 110% fractional bandwidth of the center frequency. This large bandwidth spectrum is available for high data rate communi-cations as well as radar and safety applications to operate in. The UWB technology has another advantage from the power consumption point of view. Due to spreading the ener-gy of the UWB signals over a large frequency band, the maximum power available to the antenna –as part of UWB sys‐ tem– will be as small as in order of 0.5mW according to the FCC spectral mask. This power is considered to be a small value and it is actually very close to the noise floor compared to

UWB antennas, key components of the UWB system, have received attention and significant research in recent years [3]-[28]. With theincreasing popularity of UWB systems, there have been breakthroughs in the design of UWB antennas. Implementation of a UWB system is facing many challenges and one of these challenges is to develop an appropriate antenna. This is because the antenna is an important part of the UWB system and it affects the overall performance of the system. Currently, there are many antenna designs that can achieve

> © 2013 Haraz and Sebak; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

> © 2013 Haraz and Sebak; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Osama Haraz and Abdel-Razik Sebak

http://dx.doi.org/10.5772/51403

**1. Introduction**

Additional information is available at the end of the chapter

### **UWB Antennas for Wireless Applications**

Osama Haraz and Abdel-Razik Sebak

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/51403

#### **1. Introduction**

Currently, there is an increased interest in ultra-wideband (UWB) technology for use in sev‐ eral present and future applications. UWB technology received a major boost especially in 2002 since the US Federal Communication Commission (FCC) permitted the authorization of using the unlicensed frequency band starting from 3.1 to 10.6 GHz for commercial com‐ munication applications [1]. Although existing third-generation (3G) communication tech‐ nology can provide us with many wide services such as fast internet access, video telephony, enhanced video/music download as well as digital voice services, UWB –as a new technology– is very promising for many reasons. The FCC allocated an absolute band‐ width up to 7.5 GHz which is about 110% fractional bandwidth of the center frequency. This large bandwidth spectrum is available for high data rate communi-cations as well as radar and safety applications to operate in. The UWB technology has another advantage from the power consumption point of view. Due to spreading the ener-gy of the UWB signals over a large frequency band, the maximum power available to the antenna –as part of UWB sys‐ tem– will be as small as in order of 0.5mW according to the FCC spectral mask. This power is considered to be a small value and it is actually very close to the noise floor compared to what is currently used in different radio communica-tion systems [2].

#### **1.1. Different UWB Antenna Designs**

UWB antennas, key components of the UWB system, have received attention and significant research in recent years [3]-[28]. With theincreasing popularity of UWB systems, there have been breakthroughs in the design of UWB antennas. Implementation of a UWB system is facing many challenges and one of these challenges is to develop an appropriate antenna. This is because the antenna is an important part of the UWB system and it affects the overall performance of the system. Currently, there are many antenna designs that can achieve

© 2013 Haraz and Sebak; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Haraz and Sebak; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

broad bandwidth to be used in UWB systems such as the Vivaldi antenna, bi-conical anten‐ na, log periodic antenna and spiral antenna as shown in Fig. 1. A Vivaldi antenna [3]-[4] is one of the candidate antennas for UWB operation. It has a directional radiation pattern and hence it is not suitable for either indoor wireless communication or mobile/portable devices which need omni-directional radiation patternsto enable easyand efficient communication between transmitters and receivers in all directions. Mono-conical and bi-conical antennas [5] have bulky structures with large physical dimensions which limit their applications. Al‐ so, log periodic [6] and spiral antennas [7] are two different UWB antennas that can operate in the 3.1-10.6 GHz frequency band but are not recommended for indoor wireless communi‐ cationapplications or mobile/portable devices. This is because they have large physical di‐ mensions as well as dispersive characteristics with frequency and severe ringing effect [6]. This is why we are looking for another candidate for UWB indoor wireless communications and mobile/portable devices that can overcome all these shortcomings. This candidate is the planar or printed monopole antenna [8]-[28]. Planar monopole antennas [8]-[10] with differ‐ ent shapes of polygonal (rectangular, trapezoidal...etc), circular, elliptical…etc have been proposed for UWB applications as shown in Fig. 2.

monopole antenna prototypes for UWB short-range wireless communication applications. The printed disc monopole antennas are chosen because they have small a size and omnidirectional radiation patterns with large bandwidth. In order to understand their operation mechanism that leads to the UWB characteristics, those antenna designs are numerically studied. Also, the important physical parameters which affect the antenna performances are investigated numerically using extensive parametric studies in order to obtain some quanti‐

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 127

**Figure 1.** (a) Vivaldi antenna [4] (b) Mono-conical and bi-conical antenna [5] (c) Log-periodic antenna [6] and (d) Spi‐

**Figure 2.** Modified shape planar antennas for UWB applications (a) rectangular, (b) circular and elliptical, (c) other

tative guidelines for designing these types of antennas.

ral and conical spiral antenna [7].

**Figure 3.** Planar PCB or printed antenna designs [8]-[20].

**Figure 4.** Printed antenna designs with single bandstop functions [21]-[28].

shapes.

#### **1.2. UWB Antennas for Wireless Communications**

Due to their wide frequency impedance bandwidth, simple structure, easy fabrication on printed circuit boards (PCBs), and omni-directional radiation patterns, printed PCB versions of planar monopole antennas are considered to be promising candidates for applications in UWB communications. Recent UWB antenna designs focus on small printed antennas be‐ cause of their ease of fabrication and their ability to be integrated with other components on the same PCBs [11]-[19]. Fig. 3 illustrates several realizations of planar PCB or printed anten‐ na deigns.

#### **1.3. UWB Antennas with Bandstop Function**

However, there are several existing NB communication systems operating below 10.6 GHz in the same UWB frequency band and may cause interference with the UWB systems such as IEEE 802.11a WLAN system or HIPERLAN/2 wireless system. These systems operate at 5.15-5.825 GHz which may cause interference with a UWB system. To avoid the interference with the existing wireless systems, a filter with bandstop characteristics maybe integrated with UWB antennas to achieve a notch function at the interfering frequency band [21]-[28]. Fig. 4 shows several developed bandstop antenna designs.

This chapter focuses on the development of different novel UWB microstrip-line-fed printed disc monopole and hybrid antennas with an emphasis of their frequency domain perform‐ ance. Different antenna configurations are proposed and designed in order to find a good candidate for UWB operation. The reasonable antenna candidate should satisfy UWB per‐ formance requirements including small size, constant gain, radiation pattern stability and phase linearity through the frequency band of interest. Also, the designed UWB antenna should have ease of manufacturing and integration with other mi-crowave components. We have simulated, designed, fabricated and then tested experi-mentally different printed disc monopole antenna prototypes for UWB short-range wireless communication applications. The printed disc monopole antennas are chosen because they have small a size and omnidirectional radiation patterns with large bandwidth. In order to understand their operation mechanism that leads to the UWB characteristics, those antenna designs are numerically studied. Also, the important physical parameters which affect the antenna performances are investigated numerically using extensive parametric studies in order to obtain some quanti‐ tative guidelines for designing these types of antennas.

**Figure 1.** (a) Vivaldi antenna [4] (b) Mono-conical and bi-conical antenna [5] (c) Log-periodic antenna [6] and (d) Spi‐ ral and conical spiral antenna [7].

**Figure 2.** Modified shape planar antennas for UWB applications (a) rectangular, (b) circular and elliptical, (c) other shapes.

**Figure 3.** Planar PCB or printed antenna designs [8]-[20].

broad bandwidth to be used in UWB systems such as the Vivaldi antenna, bi-conical anten‐ na, log periodic antenna and spiral antenna as shown in Fig. 1. A Vivaldi antenna [3]-[4] is one of the candidate antennas for UWB operation. It has a directional radiation pattern and hence it is not suitable for either indoor wireless communication or mobile/portable devices which need omni-directional radiation patternsto enable easyand efficient communication between transmitters and receivers in all directions. Mono-conical and bi-conical antennas [5] have bulky structures with large physical dimensions which limit their applications. Al‐ so, log periodic [6] and spiral antennas [7] are two different UWB antennas that can operate in the 3.1-10.6 GHz frequency band but are not recommended for indoor wireless communi‐ cationapplications or mobile/portable devices. This is because they have large physical di‐ mensions as well as dispersive characteristics with frequency and severe ringing effect [6]. This is why we are looking for another candidate for UWB indoor wireless communications and mobile/portable devices that can overcome all these shortcomings. This candidate is the planar or printed monopole antenna [8]-[28]. Planar monopole antennas [8]-[10] with differ‐ ent shapes of polygonal (rectangular, trapezoidal...etc), circular, elliptical…etc have been

Due to their wide frequency impedance bandwidth, simple structure, easy fabrication on printed circuit boards (PCBs), and omni-directional radiation patterns, printed PCB versions of planar monopole antennas are considered to be promising candidates for applications in UWB communications. Recent UWB antenna designs focus on small printed antennas be‐ cause of their ease of fabrication and their ability to be integrated with other components on the same PCBs [11]-[19]. Fig. 3 illustrates several realizations of planar PCB or printed anten‐

However, there are several existing NB communication systems operating below 10.6 GHz in the same UWB frequency band and may cause interference with the UWB systems such as IEEE 802.11a WLAN system or HIPERLAN/2 wireless system. These systems operate at 5.15-5.825 GHz which may cause interference with a UWB system. To avoid the interference with the existing wireless systems, a filter with bandstop characteristics maybe integrated with UWB antennas to achieve a notch function at the interfering frequency band [21]-[28].

This chapter focuses on the development of different novel UWB microstrip-line-fed printed disc monopole and hybrid antennas with an emphasis of their frequency domain perform‐ ance. Different antenna configurations are proposed and designed in order to find a good candidate for UWB operation. The reasonable antenna candidate should satisfy UWB per‐ formance requirements including small size, constant gain, radiation pattern stability and phase linearity through the frequency band of interest. Also, the designed UWB antenna should have ease of manufacturing and integration with other mi-crowave components. We have simulated, designed, fabricated and then tested experi-mentally different printed disc

proposed for UWB applications as shown in Fig. 2.

126 Advancement in Microstrip Antennas with Recent Applications

**1.2. UWB Antennas for Wireless Communications**

**1.3. UWB Antennas with Bandstop Function**

Fig. 4 shows several developed bandstop antenna designs.

na deigns.

**Figure 4.** Printed antenna designs with single bandstop functions [21]-[28].

#### **2. Operation Mechanism of UWB Monopole Antennas**

Printed disc monopole antennas are considered to be good candidates for UWB applications because they have a simple structure, easy fabrication, wideband characteristics, and omnidirectional radiation patterns [11]-[28]. The geometry of the reference printed circular disc monopole antenna is shown in Fig. 5. To determine the initial parameters of the printed cir‐ cular disc monopole antenna, we should first understand their operation mechanism. It has been shown that disc monopoles with a finite ground plane are capable of supporting multi‐ ple resonant modes instead of only one resonant mode (as in a conventional circular patch antenna) over a complete ground plane [29]. Overlapping closely spaced multiple resonance modes (f1, f2, f3, …, fN) as shown in Fig. 6 can achieve a wide bandwidth and this is the idea behind the UWB bandwidth of circular disc monopole antennas. The frequency of the first resonant mode can be determined by the size of the circular disc. At the first resonance f1, the disc antenna tends to behave like a quarter-wavelength monopole antenna, i.e. λ/4. That means the diameter of the circular disc is 2r = λ/4 at the first resonant frequency.

Then the higher order modes f2, f3, …,fN will be the harmonics of the first or fundamental mode of the disc. Unlike the conventional patch antennas with a complete ground plane, the ground plane of disc monopole antennas should be of a finite length LG to support multiple resonances and hence achieve wideband operation. The width of the ground plane W is found to be approximately twice the diameter of the disc or W=λ/2 at the first resonant [17].

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 129

The printed disc monopole antenna can be fed using different feeding techniques such as microstrip line, coplanar waveguide (CPW), aperture coupling, or proximity coupling. In the case of a microstrip line feed, the width of the microstrip feed line Wfeed is chosen to achieve a 50Ω characteristic impedance. The other antenna parameters such as the feed gap between the finite ground plane and the radiating circular disc d and the length of the finite ground plane LG can be determined using a full-wave EM numerical modeling techniques. The small feed gap between the finite ground plane and the radiating circular disc d is a very critical parameter which greatly affects the antenna impedance matching between the

**Figure 7.** The idea of integrating a bandstop filtering element to the reference circular disc monopole antenna.

To avoid interference with some existing wireless systems in the 5.15-5.825 GHz frequency band, a filter with bandstop characteristics maybe integrated with UWB antennas to achieve a notch function at the interfering frequency band. The idea of integrating a bandstop filter‐ ing element to the monopole antenna is illustrated in Fig. 7. Recently, several techniques have been introduced to achieve a single band notch within this frequency band. The most popular technique is embedding a narrow slot into the radiating patch. The slot may have different shapes such as C- shaped, slit ring resonator (SRR), L- shaped,U- or V- shaped, πshaped slot.…etc. Some other techniques are based on using parasitic strips, i.e., inverted Cshaped parasitic strip. Other techniques are based on using a slot defected ground structure

As mentioned in the introductory section of this chapter, there are several types of printed disc monopoles which exhibit ultra-wide impedance bandwidth. Here, different categories

of disc monopoles will be investigated both numerically and experimentally.

microstrip feedline and the radiating disc.

in the ground plane, i.e., H-shaped slot DGS.

**3. UWB Disc Monopole Antennas**

**Figure 5.** The configuration of the reference printed circular disc monopole antenna showing the necessary anten‐ na parameters.

**Figure 6.** The concept of overlapping closely-spaced multiple resonance modes for the reference circular disc monop‐ ole antenna (reproduced from [30]).

Then the higher order modes f2, f3, …,fN will be the harmonics of the first or fundamental mode of the disc. Unlike the conventional patch antennas with a complete ground plane, the ground plane of disc monopole antennas should be of a finite length LG to support multiple resonances and hence achieve wideband operation. The width of the ground plane W is found to be approximately twice the diameter of the disc or W=λ/2 at the first resonant [17].

The printed disc monopole antenna can be fed using different feeding techniques such as microstrip line, coplanar waveguide (CPW), aperture coupling, or proximity coupling. In the case of a microstrip line feed, the width of the microstrip feed line Wfeed is chosen to achieve a 50Ω characteristic impedance. The other antenna parameters such as the feed gap between the finite ground plane and the radiating circular disc d and the length of the finite ground plane LG can be determined using a full-wave EM numerical modeling techniques. The small feed gap between the finite ground plane and the radiating circular disc d is a very critical parameter which greatly affects the antenna impedance matching between the microstrip feedline and the radiating disc.

**Figure 7.** The idea of integrating a bandstop filtering element to the reference circular disc monopole antenna.

To avoid interference with some existing wireless systems in the 5.15-5.825 GHz frequency band, a filter with bandstop characteristics maybe integrated with UWB antennas to achieve a notch function at the interfering frequency band. The idea of integrating a bandstop filter‐ ing element to the monopole antenna is illustrated in Fig. 7. Recently, several techniques have been introduced to achieve a single band notch within this frequency band. The most popular technique is embedding a narrow slot into the radiating patch. The slot may have different shapes such as C- shaped, slit ring resonator (SRR), L- shaped,U- or V- shaped, πshaped slot.…etc. Some other techniques are based on using parasitic strips, i.e., inverted Cshaped parasitic strip. Other techniques are based on using a slot defected ground structure in the ground plane, i.e., H-shaped slot DGS.

#### **3. UWB Disc Monopole Antennas**

**2. Operation Mechanism of UWB Monopole Antennas**

128 Advancement in Microstrip Antennas with Recent Applications

Printed disc monopole antennas are considered to be good candidates for UWB applications because they have a simple structure, easy fabrication, wideband characteristics, and omnidirectional radiation patterns [11]-[28]. The geometry of the reference printed circular disc monopole antenna is shown in Fig. 5. To determine the initial parameters of the printed cir‐ cular disc monopole antenna, we should first understand their operation mechanism. It has been shown that disc monopoles with a finite ground plane are capable of supporting multi‐ ple resonant modes instead of only one resonant mode (as in a conventional circular patch antenna) over a complete ground plane [29]. Overlapping closely spaced multiple resonance modes (f1, f2, f3, …, fN) as shown in Fig. 6 can achieve a wide bandwidth and this is the idea behind the UWB bandwidth of circular disc monopole antennas. The frequency of the first resonant mode can be determined by the size of the circular disc. At the first resonance f1, the disc antenna tends to behave like a quarter-wavelength monopole antenna, i.e. λ/4.

That means the diameter of the circular disc is 2r = λ/4 at the first resonant frequency.

**Figure 5.** The configuration of the reference printed circular disc monopole antenna showing the necessary anten‐

**Figure 6.** The concept of overlapping closely-spaced multiple resonance modes for the reference circular disc monop‐

na parameters.

ole antenna (reproduced from [30]).

As mentioned in the introductory section of this chapter, there are several types of printed disc monopoles which exhibit ultra-wide impedance bandwidth. Here, different categories of disc monopoles will be investigated both numerically and experimentally.

#### **3.1. Printed Circular Disc Monopole Antenna with Two Steps and a Circular Slot**

determined. The circular slot inside the radiating patch acts as an impedance matching ele‐ ment which controls the antenna impedance matching as well as the antenna bandwidth. Also, the circular slot inside the radiating patch can be used for miniaturizing the monopole antenna. Also, it can be noticed that the rectangular steps have no remarkable effect on the overall antenna impedance bandwidth. The opti-mum values for feed gap width, slot radius and steps dimensions are d = 1 mm, RS = 3 mm and W1 (= 2W2) = 8 mm and L1 (= L2) = 3

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 131

**Figure 9.** Parametric studies of effect of (a) substrate width W (b) feed gap width d (c) circular slot radius RS and (d)

Cutting out two rectangular steps and a circular slot from the radiating patch to reduce the overall metallic area and hence reduce the antenna copper losses without affecting the an‐ tenna operation or disturbing the current distribution of the antenna is a challenging task. This can be done by investigating the antenna surface current distributions. Fig. 9 presents the antenna surface current and electric field distributions for the proposed disc monopole antenna. From the electric field distributions, it is noticed that the monopole antenna sup‐ ports multiple resonant modes. It can be seen that the current distribution is mainly located close to the radiating patch edges rather than in the center. For increasing the maximum achieved impedance bandwidth, the lower resonant frequency should be decreased. This can be done by increasing the antenna perimeter which directly affects lower resonant fre‐ quency and then the antenna impedance bandwidth. To increase the antenna perimeter, cut‐ ting out steps from the radiating patch are used here. This is simply because the surface current will take longer path when the antenna perimeter p is larger and the new antenna

steps dimensions W1 and L1 on antenna reflection coefficient.

mm, respectively.

For better understanding the antenna characteristics, the antenna reflection coefficient (S11) curves are plotted in decibel or dB scale, i.e. (S11dB = 20 log|S11| = –Return loss RL).The geometry and photograph of the proposed printed circular disc monopole antenna with two steps and a circular slot is shown in Fig. 8. The radiating element is fed by a 50Ω microstrip feed line with width of Wf = 4.4 mm. The substrate used in our design is Rogers RT/duroid 5880 high frequency laminate with thickness of h = 1.575 mm, relative permittivity of εr = 2.2 and loss tangent of tanδ = 0.0009. A finite ground plane of length LG and width W lies on the other side of the substrate. The feed gap of width d between the finite ground plane and the radiating patch is a very critical parameter for antenna matching purposes and to obtain wide bandwidth performance. This proposed antenna has a reduction in the overall antenna surface area compared to those reported in [16] and [19]. A parametric study is carried out to investigate the effect of antenna physical parameters such as the width of the substrate W, the width of the feed gap d, the radius of circular slot RS and the steps dimensions W1, W2, L1 and L2 on the performance of the proposed UWB antenna.

**Figure 8.** (a) Geometry and (b) photograph of the proposed microstrip line fed monopole antenna.

#### *3.1.1. Design Analysis*

During the parametric study, one parameter varies while all other parameters are kept fixed. The optimized antenna parameters are: W = 41 mm, L = 50 mm, LG = 18 mm, R = 10 mm, Δy = 2 mm, RS = 3 mm, W1 = 8 mm, W2 = 4 mm, L1 = 3 mm and L2 = 3 mm. Fig. 9 shows the simulated antenna reflection coefficient (20 log|S11|) curves using CST Microwave Studio TM package for different values of substrate width W, feed gap width d, slot radius RS and the steps dimensions W1, W2, L1 and L2. It can be noticed from results that the smallest sub‐ strate width for obtaining the maximum available bandwidth is W = 41 mm. It can be also seen that the reflection coefficient impedance bandwidth is greatly dependent on both the feed gap width d and the circular slot radius RS and by controlling these two parameters, the impedance matching between the radiating patch and the feed line can be easily control‐ led. By tuning the width of the feed gap d, the maximum achieved impedance bandwidth is determined. The circular slot inside the radiating patch acts as an impedance matching ele‐ ment which controls the antenna impedance matching as well as the antenna bandwidth. Also, the circular slot inside the radiating patch can be used for miniaturizing the monopole antenna. Also, it can be noticed that the rectangular steps have no remarkable effect on the overall antenna impedance bandwidth. The opti-mum values for feed gap width, slot radius and steps dimensions are d = 1 mm, RS = 3 mm and W1 (= 2W2) = 8 mm and L1 (= L2) = 3 mm, respectively.

**3.1. Printed Circular Disc Monopole Antenna with Two Steps and a Circular Slot**

L1 and L2 on the performance of the proposed UWB antenna.

130 Advancement in Microstrip Antennas with Recent Applications

**Figure 8.** (a) Geometry and (b) photograph of the proposed microstrip line fed monopole antenna.

During the parametric study, one parameter varies while all other parameters are kept fixed. The optimized antenna parameters are: W = 41 mm, L = 50 mm, LG = 18 mm, R = 10 mm, Δy = 2 mm, RS = 3 mm, W1 = 8 mm, W2 = 4 mm, L1 = 3 mm and L2 = 3 mm. Fig. 9 shows the simulated antenna reflection coefficient (20 log|S11|) curves using CST Microwave Studio TM package for different values of substrate width W, feed gap width d, slot radius RS and the steps dimensions W1, W2, L1 and L2. It can be noticed from results that the smallest sub‐ strate width for obtaining the maximum available bandwidth is W = 41 mm. It can be also seen that the reflection coefficient impedance bandwidth is greatly dependent on both the feed gap width d and the circular slot radius RS and by controlling these two parameters, the impedance matching between the radiating patch and the feed line can be easily control‐ led. By tuning the width of the feed gap d, the maximum achieved impedance bandwidth is

*3.1.1. Design Analysis*

For better understanding the antenna characteristics, the antenna reflection coefficient (S11) curves are plotted in decibel or dB scale, i.e. (S11dB = 20 log|S11| = –Return loss RL).The geometry and photograph of the proposed printed circular disc monopole antenna with two steps and a circular slot is shown in Fig. 8. The radiating element is fed by a 50Ω microstrip feed line with width of Wf = 4.4 mm. The substrate used in our design is Rogers RT/duroid 5880 high frequency laminate with thickness of h = 1.575 mm, relative permittivity of εr = 2.2 and loss tangent of tanδ = 0.0009. A finite ground plane of length LG and width W lies on the other side of the substrate. The feed gap of width d between the finite ground plane and the radiating patch is a very critical parameter for antenna matching purposes and to obtain wide bandwidth performance. This proposed antenna has a reduction in the overall antenna surface area compared to those reported in [16] and [19]. A parametric study is carried out to investigate the effect of antenna physical parameters such as the width of the substrate W, the width of the feed gap d, the radius of circular slot RS and the steps dimensions W1, W2,

**Figure 9.** Parametric studies of effect of (a) substrate width W (b) feed gap width d (c) circular slot radius RS and (d) steps dimensions W1 and L1 on antenna reflection coefficient.

Cutting out two rectangular steps and a circular slot from the radiating patch to reduce the overall metallic area and hence reduce the antenna copper losses without affecting the an‐ tenna operation or disturbing the current distribution of the antenna is a challenging task. This can be done by investigating the antenna surface current distributions. Fig. 9 presents the antenna surface current and electric field distributions for the proposed disc monopole antenna. From the electric field distributions, it is noticed that the monopole antenna sup‐ ports multiple resonant modes. It can be seen that the current distribution is mainly located close to the radiating patch edges rather than in the center. For increasing the maximum achieved impedance bandwidth, the lower resonant frequency should be decreased. This can be done by increasing the antenna perimeter which directly affects lower resonant fre‐ quency and then the antenna impedance bandwidth. To increase the antenna perimeter, cut‐ ting out steps from the radiating patch are used here. This is simply because the surface current will take longer path when the antenna perimeter p is larger and the new antenna with larger perimeter appears to be like a longer length monopole and then the lowest reso‐ nance frequency fL will be decreased according to [14]:

**Figure 10.** Simulated (a) surface current and (b) electric field distributions at the three re-sonant frequencies 3.3, 6.9 and 10.2 GHz.

$$
\varepsilon\_{eff} \approx \left(\varepsilon\_r + 1\right)/2 \tag{1}
$$

Laboratory at Concordia University. All scattering parameters measurements were carried out using Agilent E8364B programmable network analyzer (PNA). The measured and simu‐ lated reflection coefficient (S11) curves are presented in Fig. 11. It can be noticed that both measured and simulated results are in good agreement with each other and the measured 10 dB return loss bandwidth ranges from 3.0 to 11.4 GHz which covers the entire UWB fre‐ quency spectrum. Compared to the simulated results, the second resonant frequency at 7 GHz is shifted up while the third resonant frequency at 10 GHz is shifted down. This may be due to the sub-miniature version A (SMA) connector losses and/or substrate losses espe‐ cially at high frequencies (7-10 GHz). Even the loss effect of the substrate is modeled correct‐ ly and taken into account in the simulations; the simulation results did not change too much and did not agree with the measured results. In general, the proposed antenna exhibits an

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 133

UWB impedance bandwidth (3.1-10.6 GHz) in both simulated and measured results.

**Figure 11.** Measured and simulated reflection coefficient curves of the proposed antenna.

the phase seems to be linear across the whole UWB frequency range.

For further understanding the antenna performance, the Ansoft HFSS simulated maximum realized total directive gain in the boresight direction and the phase of reflection coefficient ∠S11 for the proposed antenna are presented in Fig. 12. The boresight of directional antenna is defined as the direction of maximum gain of the antenna. For most of antennas, the bore‐ sight is the axis of symmetry of the antenna, i.e. z-axis. It can be seen that the antenna has good gain stability across the frequency band of interest (3.1-10.6 GHz).It ranges from 3.4 dB to 5.2 dB with gain variation of about 2dB. The behavior of the phase of reflection coefficient ∠S11 versus frequency is also studied and shown in the same figure. It can be noticed that

$$\left(f\_L \left(\text{GHz}\right) = 300 \left/ \left(p\_\bullet \overleftarrow{\varepsilon\_{\text{eff}}}\right)\right.\tag{2}$$

where εeff is the effective dielectric constant and the perimeter p units are in millimeters.

For example, in the proposed antenna design, p = 71.4 mm, εr = 2.2, then εeff = 1.6 and the calculated lower resonant frequency using Eq. (2) is found to be fL ≈ 3.3 GHz. From the si‐ mulated and measured reflection coefficient results shown in Fig. 10, the lower resonant fre‐ quency is fL ≈ 3.3 GHz which agrees well with the calculated value.

#### *3.1.2. Experimental and Simulation Results*

A prototype of the microstrip-line-fed monopole antenna with optimized dimensions was fabricated as shown in Figure8and tested experimentally in the Applied Electromagnetics Laboratory at Concordia University. All scattering parameters measurements were carried out using Agilent E8364B programmable network analyzer (PNA). The measured and simu‐ lated reflection coefficient (S11) curves are presented in Fig. 11. It can be noticed that both measured and simulated results are in good agreement with each other and the measured 10 dB return loss bandwidth ranges from 3.0 to 11.4 GHz which covers the entire UWB fre‐ quency spectrum. Compared to the simulated results, the second resonant frequency at 7 GHz is shifted up while the third resonant frequency at 10 GHz is shifted down. This may be due to the sub-miniature version A (SMA) connector losses and/or substrate losses espe‐ cially at high frequencies (7-10 GHz). Even the loss effect of the substrate is modeled correct‐ ly and taken into account in the simulations; the simulation results did not change too much and did not agree with the measured results. In general, the proposed antenna exhibits an UWB impedance bandwidth (3.1-10.6 GHz) in both simulated and measured results.

with larger perimeter appears to be like a longer length monopole and then the lowest reso‐

**Figure 10.** Simulated (a) surface current and (b) electric field distributions at the three re-sonant frequencies 3.3, 6.9

where εeff is the effective dielectric constant and the perimeter p units are in millimeters.

quency is fL ≈ 3.3 GHz which agrees well with the calculated value.

*3.1.2. Experimental and Simulation Results*

For example, in the proposed antenna design, p = 71.4 mm, εr = 2.2, then εeff = 1.6 and the calculated lower resonant frequency using Eq. (2) is found to be fL ≈ 3.3 GHz. From the si‐ mulated and measured reflection coefficient results shown in Fig. 10, the lower resonant fre‐

A prototype of the microstrip-line-fed monopole antenna with optimized dimensions was fabricated as shown in Figure8and tested experimentally in the Applied Electromagnetics

*εeff* ≈(*ε<sup>r</sup>* + 1) / 2 (1)

*f <sup>L</sup>* (GHz) =300 /(*p εeff* ) (2)

nance frequency fL will be decreased according to [14]:

132 Advancement in Microstrip Antennas with Recent Applications

and 10.2 GHz.

**Figure 11.** Measured and simulated reflection coefficient curves of the proposed antenna.

For further understanding the antenna performance, the Ansoft HFSS simulated maximum realized total directive gain in the boresight direction and the phase of reflection coefficient ∠S11 for the proposed antenna are presented in Fig. 12. The boresight of directional antenna is defined as the direction of maximum gain of the antenna. For most of antennas, the bore‐ sight is the axis of symmetry of the antenna, i.e. z-axis. It can be seen that the antenna has good gain stability across the frequency band of interest (3.1-10.6 GHz).It ranges from 3.4 dB to 5.2 dB with gain variation of about 2dB. The behavior of the phase of reflection coefficient ∠S11 versus frequency is also studied and shown in the same figure. It can be noticed that the phase seems to be linear across the whole UWB frequency range.

**Figure 12.** The simulated gain and phase of reflection coefficient ∠S11 versus frequency of the proposed microstripline-fed monopole antenna.

**Figure 13.** Measured co-pol (blue solid line), cross-pol (red dashed line), Ansoft HFSS simulated co-pol (green dashdotted line) and cross-pol (magenta dotted line), (a) E-plane and (b) H-plane radiation patterns of the proposed antenna.

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 135

A modification can be made to the above designed antenna for achieving the bandstop func‐ tion to avoid possible interference to other existing WLAN systems. A very narrow arcshaped slot is cut away from the radiating patch as shown in Fig. 14 (a) will act as a filter element to make the antenna will not respond at the bandstop frequency. For perfect bandrejection performance of UWB antenna, the return loss of the stop-band notch should be al‐ most 0dB or the reflection coefficient is almost 1.0. However, in our first band-stop antenna design, we could achieve voltage standing wave ratio (VSWR) of about 8 (reflection coeffi‐ cient is 0.78 or -2.1 dB). The arc-shaped slot filter element di-mensions will control both the bandstop frequency fnotch and the rejection bandwidth of the band-notched filter BWnotch. The arc-shaped slot filter dimensions are: the radius of the slot R1, the thickness of the slot T and the slot angle 2α. Fig. 14 (b) illustrates the simulated reflection coefficient curves using both HFSS and CST MWS for comparison. From the simulation results, it can be seen that the band-notched characteristic in the 5.0-6.0 GHz band is achieved with good agreement

Parametric studies were carried out to address the effect of arc-shaped slot dimen-sions on the band-notched performance. Figures 15 shows the effect of varying the slot radius R1, slot thickness T and the slot angle 2α parameters on the simulated antenna ref-lection coefficient, respectively. From results in Fig. 15 (a) & (c), it can be seen that the notch frequency fnotch

*3.1.3. UWB Bandstop Antenna Design*

between them.

Fig. 13 shows the radiation characteristics for the proposed antenna. Both yz-cut plane (Eplane) and xz-cut plane (H-plane) radiation patterns have been simulated using Ansoft HFSS and measured in an anechoic chamber at the three resonant frequen-cies 3.3, 6.8, and 10.2 GHz. From the measured results, the proposed antenna has omni-directional radiation pattern in H-plane at lower frequency (3.3 GHz) and near omni-directional at higher fre‐ quencies (6.9 and 10.2 GHz) with good agreement with simula-tions. The measured E-plane radiation patterns agree with the simulations especially at lower frequency (3.3 GHz) while the agreement is not as good as the H-plane patterns at higher frequencies (6.9 and 10.2 GHz). There are some ripples and discrepancies in the measured radiation patterns especial‐ ly at the higher frequencies which may be due to sen-sitivity and accuracy of the measuring devices at higher frequencies in addition to the ef-fects of the SMA feed connector and the coaxial cable. The E-plane is identified by most of UWB antenna patterns which is perpen‐ dicular to H-plane (almost symmetric). Re-searchers in UWB antenna typically define Eplane as the plane containing the feedline and the maximum radiation of the antenna. Hplane is the plane perpendicular to E-plane.

We have investigated both simulated and measured E-plane patterns. From simu-lations, nulls in E-plane at θ = 90° depend on the size of the finite ground plane and the contact point of SMA feed connector in particular at the upper edge frequency. By searching several published UWB antennas of similar disc monopole antennas, similar behavior of measured results are reported in many papers including [31]-[34].

**Figure 13.** Measured co-pol (blue solid line), cross-pol (red dashed line), Ansoft HFSS simulated co-pol (green dashdotted line) and cross-pol (magenta dotted line), (a) E-plane and (b) H-plane radiation patterns of the proposed antenna.

#### *3.1.3. UWB Bandstop Antenna Design*

**Figure 12.** The simulated gain and phase of reflection coefficient ∠S11 versus frequency of the proposed microstrip-

Fig. 13 shows the radiation characteristics for the proposed antenna. Both yz-cut plane (Eplane) and xz-cut plane (H-plane) radiation patterns have been simulated using Ansoft HFSS and measured in an anechoic chamber at the three resonant frequen-cies 3.3, 6.8, and 10.2 GHz. From the measured results, the proposed antenna has omni-directional radiation pattern in H-plane at lower frequency (3.3 GHz) and near omni-directional at higher fre‐ quencies (6.9 and 10.2 GHz) with good agreement with simula-tions. The measured E-plane radiation patterns agree with the simulations especially at lower frequency (3.3 GHz) while the agreement is not as good as the H-plane patterns at higher frequencies (6.9 and 10.2 GHz). There are some ripples and discrepancies in the measured radiation patterns especial‐ ly at the higher frequencies which may be due to sen-sitivity and accuracy of the measuring devices at higher frequencies in addition to the ef-fects of the SMA feed connector and the coaxial cable. The E-plane is identified by most of UWB antenna patterns which is perpen‐ dicular to H-plane (almost symmetric). Re-searchers in UWB antenna typically define Eplane as the plane containing the feedline and the maximum radiation of the antenna. H-

We have investigated both simulated and measured E-plane patterns. From simu-lations, nulls in E-plane at θ = 90° depend on the size of the finite ground plane and the contact point of SMA feed connector in particular at the upper edge frequency. By searching several published UWB antennas of similar disc monopole antennas, similar behavior of measured

line-fed monopole antenna.

plane is the plane perpendicular to E-plane.

134 Advancement in Microstrip Antennas with Recent Applications

results are reported in many papers including [31]-[34].

A modification can be made to the above designed antenna for achieving the bandstop func‐ tion to avoid possible interference to other existing WLAN systems. A very narrow arcshaped slot is cut away from the radiating patch as shown in Fig. 14 (a) will act as a filter element to make the antenna will not respond at the bandstop frequency. For perfect bandrejection performance of UWB antenna, the return loss of the stop-band notch should be al‐ most 0dB or the reflection coefficient is almost 1.0. However, in our first band-stop antenna design, we could achieve voltage standing wave ratio (VSWR) of about 8 (reflection coeffi‐ cient is 0.78 or -2.1 dB). The arc-shaped slot filter element di-mensions will control both the bandstop frequency fnotch and the rejection bandwidth of the band-notched filter BWnotch. The arc-shaped slot filter dimensions are: the radius of the slot R1, the thickness of the slot T and the slot angle 2α. Fig. 14 (b) illustrates the simulated reflection coefficient curves using both HFSS and CST MWS for comparison. From the simulation results, it can be seen that the band-notched characteristic in the 5.0-6.0 GHz band is achieved with good agreement between them.

Parametric studies were carried out to address the effect of arc-shaped slot dimen-sions on the band-notched performance. Figures 15 shows the effect of varying the slot radius R1, slot thickness T and the slot angle 2α parameters on the simulated antenna ref-lection coefficient, respectively. From results in Fig. 15 (a) & (c), it can be seen that the notch frequency fnotch decreases by increasing both the arc-shaped slot radius R1 and the angle 2α while the notch bandwidth BWnotch is almost the same. On the other side, both the notch frequency and bandwidth increase at the same time by increasing the slot thickness T. For achieving a band-notched performance in the 5-6 GHz frequency band, the arc-shaped slot parameter dimensions are: R1 = 7.5 mm, T = 0.7 mm and 2α = 160°.

**3.2. Maple-leaf Shaped Monopole Antennas**

In this section, we developed different maple-leaf shaped monopole antennas with two bandrejection techniques for the 5.0-6.0 GHz frequency band. Fig. 16 (a) & (b) show the geometri‐ cal configuration and the photograph of the proposed UWB maple-leaf-shaped monopole antenna prototype. The radiating element consists of a maple-leaf-shaped patch as a radiat‐ ing element which represents the Canada flag symbol. The radiat-ing patch is fed by a micro‐ strip line and both are etched on a Rogers RT Duroid 5880 substrate with dielectric constant εr = 2.2, dielectric loss tangent tanδ = 0.0009, and thickness h = 1.575 mm. The proposed antenna parameters L1 ~ L10 are determined using an extensive parametric study and optimization in both Ansoft HFSS and CST MWS to address the effect of those parameters on the overall performance of the antenna. Details of the optimized parameters are summarized in Table 1. Our target here is to design a compact antenna for UWB operation. So, we tried to reduce the overall antenna size by reducing the substrate dimensions from 50 × 41 mm2 as in the previ‐ ous antenna design to 35.48 × 30.56 mm2 as in the present antenna design. Here, there is a reduction in the an-tenna size by almost 47% compared to our first proposed antenna proto‐

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 137

type, i.e. circular disc monopole antenna with two steps and a circular slot.

**Table 1.** Maple-leaf Shaped Printed Monopole Antenna Dimensions (Units in mm).

(stepped monopole antenna).

**Parameter W L LG W1 Wf d L1 L2** Value (mm) 30.48 35.56 12.95 5.59 4.06 0.84 2.27 7.47 Parameter L3 L4 L5 L6 L7 L8 L9 L10 Value (mm) 2.65 4.10 4.34 3.05 5.39 7.73 4.02 5.24

The maple-leaf shaped monopole antenna is used to achieve wider impedance matching bandwidth by introducing many leaf arms into the main radiating patch. This will lead to increasing the overall perimeter of the antenna and hence the monopole an-tenna looks big‐ ger in size than its real physical size. This is simply because the current takes paths close to the edges rather than inside the radiating patch. The proposed maple-leaf shaped monopole antenna has a wider bandwidth with smaller size compared to the first UWB antenna design

Fig. 17 (a) illustrates the simulated and measured reflection coefficient curves against the frequency for the designed maple-leaf antenna. It can be noticed from the re-sults that the proposed antenna exhibits a simulated impedance bandwidth from 3 to 13 GHz with good agreement between Ansoft HFSS and CST simulation programs while the measured impe‐ dance bandwidth becomes dual-band, one in 4.1-7.0 GHz and the other one in 8.7-13.3 GHz. The explanation for the difference between the measured and simulated results can be easily understood if we mention that both simulated reflection coefficient curves are already very close or even touch the -10 dB level in the region 7.0-9.0 GHz frequency band. So, if there is any manufacturing error in the antenna parameters L1 ~ L10 during the fabrication proposes of the antenna prototype will be a big issue. This is in addition to calibration errors during S-

**Figure 14.** (a) Geometry of the band-notched antenna, R1 = 7.5 mm, T = 0.7 mm and 2α = 160° (b) Simulated reflec‐ tion coefficient curves versus frequency.

**Figure 15.** Simulated reflection coefficient curves versus frequency for different values of (a) arc-shaped slot radius R1, (b) thickness of the slot T and (c) the slot angle 2α.

#### **3.2. Maple-leaf Shaped Monopole Antennas**

decreases by increasing both the arc-shaped slot radius R1 and the angle 2α while the notch bandwidth BWnotch is almost the same. On the other side, both the notch frequency and bandwidth increase at the same time by increasing the slot thickness T. For achieving a band-notched performance in the 5-6 GHz frequency band, the arc-shaped slot parameter

**Figure 14.** (a) Geometry of the band-notched antenna, R1 = 7.5 mm, T = 0.7 mm and 2α = 160° (b) Simulated reflec‐

**Figure 15.** Simulated reflection coefficient curves versus frequency for different values of (a) arc-shaped slot radius R1,

dimensions are: R1 = 7.5 mm, T = 0.7 mm and 2α = 160°.

136 Advancement in Microstrip Antennas with Recent Applications

tion coefficient curves versus frequency.

(b) thickness of the slot T and (c) the slot angle 2α.

In this section, we developed different maple-leaf shaped monopole antennas with two bandrejection techniques for the 5.0-6.0 GHz frequency band. Fig. 16 (a) & (b) show the geometri‐ cal configuration and the photograph of the proposed UWB maple-leaf-shaped monopole antenna prototype. The radiating element consists of a maple-leaf-shaped patch as a radiat‐ ing element which represents the Canada flag symbol. The radiat-ing patch is fed by a micro‐ strip line and both are etched on a Rogers RT Duroid 5880 substrate with dielectric constant εr = 2.2, dielectric loss tangent tanδ = 0.0009, and thickness h = 1.575 mm. The proposed antenna parameters L1 ~ L10 are determined using an extensive parametric study and optimization in both Ansoft HFSS and CST MWS to address the effect of those parameters on the overall performance of the antenna. Details of the optimized parameters are summarized in Table 1. Our target here is to design a compact antenna for UWB operation. So, we tried to reduce the overall antenna size by reducing the substrate dimensions from 50 × 41 mm2 as in the previ‐ ous antenna design to 35.48 × 30.56 mm2 as in the present antenna design. Here, there is a reduction in the an-tenna size by almost 47% compared to our first proposed antenna proto‐ type, i.e. circular disc monopole antenna with two steps and a circular slot.


**Table 1.** Maple-leaf Shaped Printed Monopole Antenna Dimensions (Units in mm).

The maple-leaf shaped monopole antenna is used to achieve wider impedance matching bandwidth by introducing many leaf arms into the main radiating patch. This will lead to increasing the overall perimeter of the antenna and hence the monopole an-tenna looks big‐ ger in size than its real physical size. This is simply because the current takes paths close to the edges rather than inside the radiating patch. The proposed maple-leaf shaped monopole antenna has a wider bandwidth with smaller size compared to the first UWB antenna design (stepped monopole antenna).

Fig. 17 (a) illustrates the simulated and measured reflection coefficient curves against the frequency for the designed maple-leaf antenna. It can be noticed from the re-sults that the proposed antenna exhibits a simulated impedance bandwidth from 3 to 13 GHz with good agreement between Ansoft HFSS and CST simulation programs while the measured impe‐ dance bandwidth becomes dual-band, one in 4.1-7.0 GHz and the other one in 8.7-13.3 GHz. The explanation for the difference between the measured and simulated results can be easily understood if we mention that both simulated reflection coefficient curves are already very close or even touch the -10 dB level in the region 7.0-9.0 GHz frequency band. So, if there is any manufacturing error in the antenna parameters L1 ~ L10 during the fabrication proposes of the antenna prototype will be a big issue. This is in addition to calibration errors during S- parameters measurement and the effect of SMA connector which was not taken into account during simulations. Also, the manufacturing tolerance as well as the effect of SMA connec‐ tor has been simulated in CST MWS program and simulation results are shown in Fig. 17 (b) and it is found from the obtained result that it confirms the above explanation.

**Figure 16.** (a) Geometry and (b) photograph of the proposed maple-leaf shaped printed monopole antenna prototype.

**Figure 18.** Measured (red solid line) and simulated (blue dashed line) (a) E-plane and (b) H-plane radiation patterns of

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 139

We developed two different band-notched antennas using two different tech-niques for band rejection. Fig. 19 (a) introduces the first proposed band-notched anten-na which is de‐ signed by modifying the above maple-leaf antenna by cutting a narrow H-shaped slot away from the radiating patch. The H-slot acts as a filtering element where slot dimensions con‐ trol the rejection band of the band-notched filter. Fig. 19 (b) presents the second proposed band-notched antenna which is designed by cutting two narrow rectangular slits in the ground plane making a DGS. In the maple-leaf band-stop antennas, we achieved VSWR of 10 (reflection coefficient is 0.82 or -1.7 dB) with H-shaped slot and VSWR of 24 (reflection coefficient is 0.92 or -0.7 dB) with two slits in the ground. It can be concluded that using two slits in the ground plane achieves better rejection characteristics compared to using narrow

the maple-leaf antenna.

*3.2.1. Bandstop Antenna Design Prototypes*

slots (either arc-shaped or H-shaped) in the radiating patch.

**Figure 19.** Photograph and geometry of the proposed bandstop antennas using (a) H-slot (b) two slits.

The antenna radiation characteristics across the whole UWB frequency band were also in‐ vestigated. Fig. 18 shows both the measured and simulated E- and H-plane radiation pat‐ terns at frequencies 3, 5, 7, and 9 GHz, respectively. The measured H-plane radiation patterns are very close to those obtained in the simulation. It can be noticed that the H-plane patterns are omni-directional at all frequencies of interest. The measured E-plane patterns follow the shapes of the simulated ones, though the agreement is not as good as the H-plane patterns. There are some fluctuations, ripples and distortions on the measured curves, which may be caused by the SMA feed connector and the coaxial cable.

**Figure 17.** (a) Measured and simulated reflection coefficient curves of the maple-leaf an-tenna (b) effect of fabrica‐ tion tolerance on the performance of maple-leaf antenna.

**Figure 18.** Measured (red solid line) and simulated (blue dashed line) (a) E-plane and (b) H-plane radiation patterns of the maple-leaf antenna.

#### *3.2.1. Bandstop Antenna Design Prototypes*

parameters measurement and the effect of SMA connector which was not taken into account during simulations. Also, the manufacturing tolerance as well as the effect of SMA connec‐ tor has been simulated in CST MWS program and simulation results are shown in Fig. 17 (b)

**Figure 16.** (a) Geometry and (b) photograph of the proposed maple-leaf shaped printed monopole antenna prototype.

The antenna radiation characteristics across the whole UWB frequency band were also in‐ vestigated. Fig. 18 shows both the measured and simulated E- and H-plane radiation pat‐ terns at frequencies 3, 5, 7, and 9 GHz, respectively. The measured H-plane radiation patterns are very close to those obtained in the simulation. It can be noticed that the H-plane patterns are omni-directional at all frequencies of interest. The measured E-plane patterns follow the shapes of the simulated ones, though the agreement is not as good as the H-plane patterns. There are some fluctuations, ripples and distortions on the measured curves,

**Figure 17.** (a) Measured and simulated reflection coefficient curves of the maple-leaf an-tenna (b) effect of fabrica‐

which may be caused by the SMA feed connector and the coaxial cable.

tion tolerance on the performance of maple-leaf antenna.

and it is found from the obtained result that it confirms the above explanation.

138 Advancement in Microstrip Antennas with Recent Applications

We developed two different band-notched antennas using two different tech-niques for band rejection. Fig. 19 (a) introduces the first proposed band-notched anten-na which is de‐ signed by modifying the above maple-leaf antenna by cutting a narrow H-shaped slot away from the radiating patch. The H-slot acts as a filtering element where slot dimensions con‐ trol the rejection band of the band-notched filter. Fig. 19 (b) presents the second proposed band-notched antenna which is designed by cutting two narrow rectangular slits in the ground plane making a DGS. In the maple-leaf band-stop antennas, we achieved VSWR of 10 (reflection coefficient is 0.82 or -1.7 dB) with H-shaped slot and VSWR of 24 (reflection coefficient is 0.92 or -0.7 dB) with two slits in the ground. It can be concluded that using two slits in the ground plane achieves better rejection characteristics compared to using narrow slots (either arc-shaped or H-shaped) in the radiating patch.

**Figure 19.** Photograph and geometry of the proposed bandstop antennas using (a) H-slot (b) two slits.

In both techniques, we can control both the notch center frequency fnotch and the band‐ width BWnotch by adjusting the H-slot and the two slits dimensions, respectively. In the first band-notched antenna, we adjust the slot length LS, thickness WS, and location from the substrate edge DS to control the bandstop characteristic. In the second band-notched an‐ tenna, we control the bandstop characteristic by adjusting the two rectangular slits length LS, thickness WS, and distance between them S. The remarkable thing here is that the notch center frequency fnotch is controlled by adjusting the mean length of the slot or the two slits to be about one half-wavelength, i.e. λ/2 at the desired notched frequency. For example, the calculated mean length of the H-shaped slot is about 26 mm and the calculated λ/2 at the notch frequency fnotch = 5.5 GHz is 27.7 mm. It is found that the notch bandwidth BWnotch can be controlled by adjusting the thickness of the slot or the two slits.

The CST simulated antenna maximum realized gains in the bore-sight direction versus fre‐ quency for the maple-leaf antenna, band-notched antennas with H-slot and two slits are pre‐ sented in Fig. 23. It can be seen that the maple-leaf antenna gain is almost stable over the whole frequency band and it ranges from 2 dB to 4.3 dB with gain variation about 2.3 dB through the whole frequency band of interest. For band-notched antenna designs with Hslot and two slits, a sharp gain decrease is remarkably happened in the 5.0-6.0 GHz frequen‐ cy band. Gain results ensure that the band-notched antennas are not responding in the

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 141

**Figure 21.** Current distributions for the first bandstop antenna at the (a) radiating frequency f1 = 4 GHz, (b) bandstop

**Figure 22.** Current distributions for the second bandstop antenna at the (a) radiating frequency f1 = 4 GHz, (b) band‐

bandstop frequency range between 5.0 and 6.0 GHz.

frequency f2 = 5.5 GHz and (c) the radiating frequency f3 = 7 GHz.

stop frequency f2 = 5.5 GHz and (c) the radiating frequency f3 = 7 GHz.

Fig. 20 (a) & (b) present the simulated and measured reflection coefficient curves of both band-notched antennas with H-slot (WS = 0.65 mm, LS = 8.6 mm and DS = 18.6 mm) and two slits (WS = 0.5 mm, LS = 10.2 mm, and S = 3 mm), respectively. It is ob-vious from the results that the bandstop function in the 5.0-6.0 GHz is successfully achieved for both antenna de‐ signs. The discrepancies in the 7-9 GHz frequency band come from the maple-leaf antenna itself not from the filter elements for band rejection. It can also be noticed that these discrep‐ ancies in the 7-9 GHz frequency band are more re-markable in the first prototype than the second one. This is may be due to the effect of using DGS in the finite ground plane en‐ hanced the antenna performance in the 7-9 GHz frequency band.

**Figure 20.** Measured and simulated reflection coefficient curves for bandstop antennas (a) using an H-slot and (b) using two slits.

Fig. 21 and Fig. 22 show the CST simulated surface current distributions over different fre‐ quencies, i.e. 3, 5.5 and 7 GHz for both band-notched antenna designs with H-slot and two slits, respectively. It can be noticed that at the bandstop frequency 5.5 GHz, nearly all the currents are trapped at the H-shaped slot or two slits which are preventing the current from radiation while at the radiation frequencies 3 and 7 GHz, the current is uniformly distribut‐ ed through the whole radiating patch.

The CST simulated antenna maximum realized gains in the bore-sight direction versus fre‐ quency for the maple-leaf antenna, band-notched antennas with H-slot and two slits are pre‐ sented in Fig. 23. It can be seen that the maple-leaf antenna gain is almost stable over the whole frequency band and it ranges from 2 dB to 4.3 dB with gain variation about 2.3 dB through the whole frequency band of interest. For band-notched antenna designs with Hslot and two slits, a sharp gain decrease is remarkably happened in the 5.0-6.0 GHz frequen‐ cy band. Gain results ensure that the band-notched antennas are not responding in the bandstop frequency range between 5.0 and 6.0 GHz.

In both techniques, we can control both the notch center frequency fnotch and the band‐ width BWnotch by adjusting the H-slot and the two slits dimensions, respectively. In the first band-notched antenna, we adjust the slot length LS, thickness WS, and location from the substrate edge DS to control the bandstop characteristic. In the second band-notched an‐ tenna, we control the bandstop characteristic by adjusting the two rectangular slits length LS, thickness WS, and distance between them S. The remarkable thing here is that the notch center frequency fnotch is controlled by adjusting the mean length of the slot or the two slits to be about one half-wavelength, i.e. λ/2 at the desired notched frequency. For example, the calculated mean length of the H-shaped slot is about 26 mm and the calculated λ/2 at the notch frequency fnotch = 5.5 GHz is 27.7 mm. It is found that the notch bandwidth BWnotch

Fig. 20 (a) & (b) present the simulated and measured reflection coefficient curves of both band-notched antennas with H-slot (WS = 0.65 mm, LS = 8.6 mm and DS = 18.6 mm) and two slits (WS = 0.5 mm, LS = 10.2 mm, and S = 3 mm), respectively. It is ob-vious from the results that the bandstop function in the 5.0-6.0 GHz is successfully achieved for both antenna de‐ signs. The discrepancies in the 7-9 GHz frequency band come from the maple-leaf antenna itself not from the filter elements for band rejection. It can also be noticed that these discrep‐ ancies in the 7-9 GHz frequency band are more re-markable in the first prototype than the second one. This is may be due to the effect of using DGS in the finite ground plane en‐

**Figure 20.** Measured and simulated reflection coefficient curves for bandstop antennas (a) using an H-slot and (b)

Fig. 21 and Fig. 22 show the CST simulated surface current distributions over different fre‐ quencies, i.e. 3, 5.5 and 7 GHz for both band-notched antenna designs with H-slot and two slits, respectively. It can be noticed that at the bandstop frequency 5.5 GHz, nearly all the currents are trapped at the H-shaped slot or two slits which are preventing the current from radiation while at the radiation frequencies 3 and 7 GHz, the current is uniformly distribut‐

can be controlled by adjusting the thickness of the slot or the two slits.

140 Advancement in Microstrip Antennas with Recent Applications

hanced the antenna performance in the 7-9 GHz frequency band.

using two slits.

ed through the whole radiating patch.

**Figure 21.** Current distributions for the first bandstop antenna at the (a) radiating frequency f1 = 4 GHz, (b) bandstop frequency f2 = 5.5 GHz and (c) the radiating frequency f3 = 7 GHz.

**Figure 22.** Current distributions for the second bandstop antenna at the (a) radiating frequency f1 = 4 GHz, (b) band‐ stop frequency f2 = 5.5 GHz and (c) the radiating frequency f3 = 7 GHz.

**Figure 23.** Simulated gain curves versus frequency for all three maple-leaf antennas.

#### **3.3. Other Shaped Disc Monopole Antennas**

In this section we continue to enhance the UWB antenna performance to obtain a compact in size antenna with maximum possible impedance bandwidth for UWB opera-tion. We are considering the design of two compact omni-directional UWB antennas with different shape of radiating patches. The first design is the butterfly-shaped monopole antenna while the second one is trapezoidal-shaped monopole antenna with a bell-shaped cut as shown in Fig. 24 (a) and (b), respectively. The butterfly-shaped monopole an-tenna size is 35 × 35 mm2 which is bigger than the previous maple-leaf-shaped antenna (35.5 × 30.5 mm2) by about 13%. The other proposed design is the trapezoidal-shaped monopole antenna of size 34 × 30 mm2 which is smaller than the maple-leaf-shaped an-tenna by about 6%. The best candidate among all printed disc monopole antennas from the antenna size point of view is the trape‐ zoidal antenna with bell-shaped cut. Moreover, the candidate antenna still has UWB impe‐ dance bandwidth with reasonable stable radia-tion characteristics and constant gain through the desired frequency range.

**Figure 24.** Geometry and photograph of the (a) butterfly-shaped (b) trapezoidal-shaped monopole antenna.

has been fabricated for experimental investigation.

The trapezoidal-shaped antenna consists of a trapezoidal patch of dimensions L1 = 12 mm, L2 = 11 mm, W1 = 10 mm and bevel angle α = 55.7°. Two elliptical cuts have been cut out from the radiating patch forming a bell shaped cut. The first elliptical cut is of a major radius Rx1 = 10 mm and a minor radius Ry1 = 6 mm (elliptically ratio Rx1/Ry1 = 1.67). The second elliptical cut is of a minor radius Rx2 = 6 mm and a major radius Ry2 = 14 mm (elliptically ratio Ry2/Rx2 = 2.33). An antenna prototype of both structures with optimized parameters

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 143

**Figure 25.** Measured and simulated reflection coefficient curves of the (a) butterfly antenna and (b) trapezoidal antenna.

Both proposed antennas are etched on 1.575mm-thick Rogers RT 5880 substrate and fed by 50Ω characteristic impedance microstrip line. The finite ground plane length is LG = 10 mm and the feed gap width is d = 0.5 mm. The butterfly-shaped antenna consists of a radiating element of two overlapped elliptical discs of major radius a = 16.6 mm and a minor radius b = 10.4 mm (elliptically ratio a/b ≈ 1.6 forming the two wings of the butterfly). Two annular slot rings of an outer radius r1 = 2 mm and an inner radius r2 = 1 mm have been cut out from the radiating patch. They are located at distance c (= e) = 5.2 mm from the two ellipses' edges. These slot rings can increase the bandwidth of the proposed antenna and they are useful to reduce the overall metallic area.

**Figure 24.** Geometry and photograph of the (a) butterfly-shaped (b) trapezoidal-shaped monopole antenna.

**Figure 23.** Simulated gain curves versus frequency for all three maple-leaf antennas.

In this section we continue to enhance the UWB antenna performance to obtain a compact in size antenna with maximum possible impedance bandwidth for UWB opera-tion. We are considering the design of two compact omni-directional UWB antennas with different shape of radiating patches. The first design is the butterfly-shaped monopole antenna while the second one is trapezoidal-shaped monopole antenna with a bell-shaped cut as shown in Fig. 24 (a) and (b), respectively. The butterfly-shaped monopole an-tenna size is 35 × 35 mm2 which is bigger than the previous maple-leaf-shaped antenna (35.5 × 30.5 mm2) by about 13%. The other proposed design is the trapezoidal-shaped monopole antenna of size 34 × 30 mm2 which is smaller than the maple-leaf-shaped an-tenna by about 6%. The best candidate among all printed disc monopole antennas from the antenna size point of view is the trape‐ zoidal antenna with bell-shaped cut. Moreover, the candidate antenna still has UWB impe‐ dance bandwidth with reasonable stable radia-tion characteristics and constant gain through

Both proposed antennas are etched on 1.575mm-thick Rogers RT 5880 substrate and fed by 50Ω characteristic impedance microstrip line. The finite ground plane length is LG = 10 mm and the feed gap width is d = 0.5 mm. The butterfly-shaped antenna consists of a radiating element of two overlapped elliptical discs of major radius a = 16.6 mm and a minor radius b = 10.4 mm (elliptically ratio a/b ≈ 1.6 forming the two wings of the butterfly). Two annular slot rings of an outer radius r1 = 2 mm and an inner radius r2 = 1 mm have been cut out from the radiating patch. They are located at distance c (= e) = 5.2 mm from the two ellipses' edges. These slot rings can increase the bandwidth of the proposed antenna and they are

**3.3. Other Shaped Disc Monopole Antennas**

142 Advancement in Microstrip Antennas with Recent Applications

the desired frequency range.

useful to reduce the overall metallic area.

The trapezoidal-shaped antenna consists of a trapezoidal patch of dimensions L1 = 12 mm, L2 = 11 mm, W1 = 10 mm and bevel angle α = 55.7°. Two elliptical cuts have been cut out from the radiating patch forming a bell shaped cut. The first elliptical cut is of a major radius Rx1 = 10 mm and a minor radius Ry1 = 6 mm (elliptically ratio Rx1/Ry1 = 1.67). The second elliptical cut is of a minor radius Rx2 = 6 mm and a major radius Ry2 = 14 mm (elliptically ratio Ry2/Rx2 = 2.33). An antenna prototype of both structures with optimized parameters has been fabricated for experimental investigation.

**Figure 25.** Measured and simulated reflection coefficient curves of the (a) butterfly antenna and (b) trapezoidal antenna.

The measured and simulated reflection coefficient curves against frequency for butterfly and trapezoidal antennas are plotted in Fig. 25, respectively. It is observed from the results that the simulated reflection coefficient with Ansoft HFSS and CST are almost in good agreement and both antennas exhibit wide impedance bandwidth from 3 GHz to beyond 12 GHz (FBW is > 110%) for both antennas. The measured results shows that the both antenna designs still have wide impedance bandwidth covering the UWB frequency range. It is shown that there are different resonances occur at different frequencies across the UWB frequency range and the overlap among these resonances achieve the wide bandwidth characteristic of those types of printed monopole antenna. The measured and simulated E- and H-plane radiation patterns at frequencies 3, 5, 7 and 9 GHz are illustrated in Fig. 26 and Fig. 27, respectively. As expected, both antennas exhibit a dipole-like radiation patterns in E-plane and good omni-directional radiation patterns in H-plane.

Both physical and electrical properties of different UWB disc monopole antennas for shortrange wireless communications are summarized in Table 2. The comparison includes the overall antennas dimensions, 10 dB return loss bandwidth, realized gain and groub delay features. It can be seen that the Trapezoidal monopole antenna with bell-shaped cut is the good candidate among all proposed antenna designs in terms of both physical and electri‐

Dimensions (mm) 41 × 50 × 1.575 30.5 × 35.5 × 1.575 35 × 35 × 1.575 30 × 34 × 1.575

117% 52%, 42% 113% 112%

In this section, we investigate the transmission/reception (Tx/Rx) characteristics of different UWB antennas discussed above in both time and frequency domains. We set up various sce‐ narios and study the communication link between two identical prototype an-tennas. The distance between the transmitting and receiving antennas is assumed to be 30 cm which is approximately 3 wavelengths at the lower frequency of the considered band of operation (antennas are in the far field of each other). Two different scenarios are established for our study. The first one is the face-to-face scenario where the two identic-al antennas are placed in vertical position facing each other at a separation distance be-tween the two antennas of d as shown in Fig. 28(a). The second case is the end-to-end scenario where the two antennas are placed in horizontal position facing each other at a separation distance d as shown in Fig. 28(b). This study is carried out calculated in the E-plane (ϕ = 90°) at different observa‐

(Dual-band)

2.0~4.3 ±2.3

Group delay (ns) 4.2 2.7 1.5 4.2

**Maple-leaf antenna Butterfly antenna Trapezoidal antenna**

2.0~4.7 ±2.7

3.0~10.8 3.2~11.4

**with bell-shaped cut**

2.7~5.3 ±2.6

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 145

cal propoerties.

10 dB RL bandwidth

10 dB RL bandwidth

tion angles θ.

Realized gain (dB) 3.4~5.2

(GHz)

(%)

**Parameter Circular disc**

**slot**

±1.8

**monopole with two steps and a circular**

3.0~11.5 4.1~7.0, 8.7~13.3

**Table 2.** Comparison among Different UWB Antenna Design Prototypes.

**3.4. Transmission Characteristics of UWB Antennas**

**Figure 26.** Measured (red solid) and simulated (blue dashed) (a) E-plane and (b) H-plane radiation patterns for butter‐ fly antenna.

**Figure 27.** E- and H-plane radiation patterns of the trapezoidal antenna. Blue dashed lines for simulated and red solid lines for measured.

Both physical and electrical properties of different UWB disc monopole antennas for shortrange wireless communications are summarized in Table 2. The comparison includes the overall antennas dimensions, 10 dB return loss bandwidth, realized gain and groub delay features. It can be seen that the Trapezoidal monopole antenna with bell-shaped cut is the good candidate among all proposed antenna designs in terms of both physical and electri‐ cal propoerties.


**Table 2.** Comparison among Different UWB Antenna Design Prototypes.

The measured and simulated reflection coefficient curves against frequency for butterfly and trapezoidal antennas are plotted in Fig. 25, respectively. It is observed from the results that the simulated reflection coefficient with Ansoft HFSS and CST are almost in good agreement and both antennas exhibit wide impedance bandwidth from 3 GHz to beyond 12 GHz (FBW is > 110%) for both antennas. The measured results shows that the both antenna designs still have wide impedance bandwidth covering the UWB frequency range. It is shown that there are different resonances occur at different frequencies across the UWB frequency range and the overlap among these resonances achieve the wide bandwidth characteristic of those types of printed monopole antenna. The measured and simulated E- and H-plane radiation patterns at frequencies 3, 5, 7 and 9 GHz are illustrated in Fig. 26 and Fig. 27, respectively. As expected, both antennas exhibit a dipole-like radiation patterns in E-plane and good omni-directional

**Figure 26.** Measured (red solid) and simulated (blue dashed) (a) E-plane and (b) H-plane radiation patterns for butter‐

**Figure 27.** E- and H-plane radiation patterns of the trapezoidal antenna. Blue dashed lines for simulated and red solid

radiation patterns in H-plane.

144 Advancement in Microstrip Antennas with Recent Applications

fly antenna.

lines for measured.

#### **3.4. Transmission Characteristics of UWB Antennas**

In this section, we investigate the transmission/reception (Tx/Rx) characteristics of different UWB antennas discussed above in both time and frequency domains. We set up various sce‐ narios and study the communication link between two identical prototype an-tennas. The distance between the transmitting and receiving antennas is assumed to be 30 cm which is approximately 3 wavelengths at the lower frequency of the considered band of operation (antennas are in the far field of each other). Two different scenarios are established for our study. The first one is the face-to-face scenario where the two identic-al antennas are placed in vertical position facing each other at a separation distance be-tween the two antennas of d as shown in Fig. 28(a). The second case is the end-to-end scenario where the two antennas are placed in horizontal position facing each other at a separation distance d as shown in Fig. 28(b). This study is carried out calculated in the E-plane (ϕ = 90°) at different observa‐ tion angles θ.

**Figure 28.** Configuration of UWB transmission system in case of (a) face-to-face scenario and (b) end-to-end scenario.

#### *3.4.1. Time-Domain Characteristics*

For a complete description of the antenna characteristics, the time domain behavior is calcu‐ lated in the E-plane (ϕ = 90°) at different observation angles: θ = 0°, 30°, 60°, 90°. Referring to Fig. 29(a), the incident wave arriving at the receiving antenna is assumed to be the fourth derivative of a Gaussian function

$$\mathbf{A}\_{S\_l}(t) = A \left( \mathbf{3} - 6 \frac{\left( 4 \pi \right)}{\tau^2} t \right)^2 + \left( \frac{4 \pi}{\tau^2} \right)^2 t^4 \Big/ \bullet e^{-2 \pi \left( \frac{t}{\tau} \right)^2} \Big/ \mathbf{V} \Big/ \mathbf{m} \Big\} \tag{3}$$

**Figure 30.** CST Simulated radiation waveforms in the E-plane at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for (a) circular disc with two steps and a circular slot antenna (b) maple-leaf monopole antenna (c) butterfly

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 147

Since virtual probes are situated in the E-plane (ϕ = 90°), we expect the Tx/Rx system fre‐ quency-domain transfer function in face-to-face scenario to become more flat than end-toend scenario. The separation distance between two transmit and receive antennas is set to d = 30 cm. The simulated impulse responses for both scenarios are given in Fig. 31(a) and (b), respectively. It is shown the ringing effect is slightly less in the face-to-face case compared to the end-to-end case. Fig. 32 shows the simulated transmission coefficients |S21| against fre‐ quency at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for different UWB an‐

monopole antenna (d) trapezoidal monopole antenna.

*3.4.2. Frequency-Domain Characteristics*

tenna prototypes.

where A = 0.1 and τ = 0.175 ns. The normalized spectrum of this pulse is illustrated in Fig. 29(b), and proves to comply with the required FCC indoor emission mask. Further refining the pulse spectrum can be achieved by utilizing some optimization algorithms. The pulse spectrum is then multiplied by the normalized antenna transfer functions and an inverse Fourier transform (IFT) is performed to achieve the required time domain response. The output waveform at the receiving antenna terminal can therefore be expressed by where represents an ideal bandpass filter from 1 to 18 GHz.

Fig. 30 presents the CST Simulated radiation waveforms in the E-plane at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for different UWB antenna prototypes.

**Figure 29.** (a) Received UWB pulse shape and (b) spectrum of a single received UWB pulse [35].

**Figure 30.** CST Simulated radiation waveforms in the E-plane at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for (a) circular disc with two steps and a circular slot antenna (b) maple-leaf monopole antenna (c) butterfly monopole antenna (d) trapezoidal monopole antenna.

#### *3.4.2. Frequency-Domain Characteristics*

**Figure 28.** Configuration of UWB transmission system in case of (a) face-to-face scenario and (b) end-to-end scenario.

For a complete description of the antenna characteristics, the time domain behavior is calcu‐ lated in the E-plane (ϕ = 90°) at different observation angles: θ = 0°, 30°, 60°, 90°. Referring to Fig. 29(a), the incident wave arriving at the receiving antenna is assumed to be the fourth

> *τ* 2 )2 *t* 4

where A = 0.1 and τ = 0.175 ns. The normalized spectrum of this pulse is illustrated in Fig. 29(b), and proves to comply with the required FCC indoor emission mask. Further refining the pulse spectrum can be achieved by utilizing some optimization algorithms. The pulse spectrum is then multiplied by the normalized antenna transfer functions and an inverse Fourier transform (IFT) is performed to achieve the required time domain response. The output waveform at the receiving antenna terminal can therefore be expressed by where

Fig. 30 presents the CST Simulated radiation waveforms in the E-plane at different angles θ

) <sup>∙</sup>*e*-2*π*( *<sup>t</sup> τ* )2

(V / m) (3)

*3.4.1. Time-Domain Characteristics*

derivative of a Gaussian function

*si*

146 Advancement in Microstrip Antennas with Recent Applications

(*t*)= *<sup>A</sup>*(<sup>3</sup> - <sup>6</sup>( <sup>4</sup>*<sup>π</sup>*

represents an ideal bandpass filter from 1 to 18 GHz.

*<sup>τ</sup>* <sup>2</sup> )*<sup>t</sup>* <sup>2</sup> <sup>+</sup> ( <sup>4</sup>*<sup>π</sup>*

= 0°, 30°, 60°, 90° in face-to-face scenario for different UWB antenna prototypes.

**Figure 29.** (a) Received UWB pulse shape and (b) spectrum of a single received UWB pulse [35].

Since virtual probes are situated in the E-plane (ϕ = 90°), we expect the Tx/Rx system fre‐ quency-domain transfer function in face-to-face scenario to become more flat than end-toend scenario. The separation distance between two transmit and receive antennas is set to d = 30 cm. The simulated impulse responses for both scenarios are given in Fig. 31(a) and (b), respectively. It is shown the ringing effect is slightly less in the face-to-face case compared to the end-to-end case. Fig. 32 shows the simulated transmission coefficients |S21| against fre‐ quency at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for different UWB an‐ tenna prototypes.

**Figure 32.** CST Simulated transmission coefficients |S21| as function of frequency at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for (a) circular disc with two steps and a circular slot antenna (b) maple-leaf monopole

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 149

In this chapter, different UWB disc monopole antennas have been developed in microstrip PCB technology to achieve low profile and ease of integration. Parametric studies to see the effect of some antenna parameters on its performance have been numer-ically investigated. For further understanding the behavior of the proposed antennas, sur-face current distribu‐ tions have been simulated and presented. Different techniques for obtaining bandstop func‐ tion in the 5.0-6.0 GHz frequency band to avoid interference with other existing WLAN systems have been numerically and experimentally presented. The effects of band-notched parameters on the band-notch frequency and bandwidth have been studied. The chapter has investigated the frequency domain performances of different printed disc monopole anten‐ nas and hybrid antenna. Experimental as well as the simulated results have confirmed UWB characteristics of the proposed antennas with nearly stable omni-directional radiation prop‐ erties over the entire frequency band of interest. These features and their small sizes make

antenna (c) butterfly monopole antenna (d) trapezoidal monopole antenna.

them attractive for future UWB applications.

**4. Summary**

**Figure 31.** CST Simulated transmission coefficients |S21| as a function of frequency for different UWB antennas in case of (a) face-to-face scenario (b) end-to-end scenario.

**Figure 32.** CST Simulated transmission coefficients |S21| as function of frequency at different angles θ = 0°, 30°, 60°, 90° in face-to-face scenario for (a) circular disc with two steps and a circular slot antenna (b) maple-leaf monopole antenna (c) butterfly monopole antenna (d) trapezoidal monopole antenna.

#### **4. Summary**

**Figure 31.** CST Simulated transmission coefficients |S21| as a function of frequency for different UWB antennas in case

of (a) face-to-face scenario (b) end-to-end scenario.

148 Advancement in Microstrip Antennas with Recent Applications

In this chapter, different UWB disc monopole antennas have been developed in microstrip PCB technology to achieve low profile and ease of integration. Parametric studies to see the effect of some antenna parameters on its performance have been numer-ically investigated. For further understanding the behavior of the proposed antennas, sur-face current distribu‐ tions have been simulated and presented. Different techniques for obtaining bandstop func‐ tion in the 5.0-6.0 GHz frequency band to avoid interference with other existing WLAN systems have been numerically and experimentally presented. The effects of band-notched parameters on the band-notch frequency and bandwidth have been studied. The chapter has investigated the frequency domain performances of different printed disc monopole anten‐ nas and hybrid antenna. Experimental as well as the simulated results have confirmed UWB characteristics of the proposed antennas with nearly stable omni-directional radiation prop‐ erties over the entire frequency band of interest. These features and their small sizes make them attractive for future UWB applications.

#### **Acknowledgements**

This research is partially supported by the King Saud University - National Plan for Sciences and Technology (NPST) through Research Grant 09ELE858-02 and by KACST Technology Innovation Center in RFTONICS hosted by King Saud University.

[8] Su, S. W., Wong, K. L., & Tang, C. L. (2004). Ultra-wideband square planar monopole antenna for IEEE 802.16a operation in the 2-11 GHz band. *Microwave Opt. Tech-nol.*

UWB Antennas for Wireless Applications http://dx.doi.org/10.5772/51403 151

[9] Chen, Z. N., Ammann, M. J., & Chia, M. Y. W. (2003). Broadband square annular pla‐

[10] Chen, Z. N., Ammann, M. J., Chia, M. Y. W., & See, T. S. P. (2002). Annular planar mono-pole antennas. *IEE Proc. Microw. Antennas Propag.*, 149(4), 200-203, Aug.

[11] Ahmed, O. M., Elboushi, A., & Sebak, A. R. (2012). Design of Half Elliptical Ring Mo‐ nopole Antennas with Elliptical Slot in Ground Plane for Future UWB Applications,

[12] Ahmed, O. M. H., & Sebak, A. R. (2011). A Novel Printed Monopole Antenna for Fu‐ ture Ultrawideband Communication Systems. Microwave and Optical Technology

[13] Osama, M. H. Ahmed, & Sebak, Abdel-Razik. (2010). Planar Ultrawideband Antenna Array for Short-Range Wireless Communications. *Microwave and Optical Technology*

[14] Azenui, N. C., & Yang, H. Y. D. (2007). A printed crescent patch antenna for ultrawi‐

[15] Ahmed, O., & Sebak, A. R. (2008). A Printed Monopole Antenna with Two Steps and a Circular Slot for UWB Applications. *IEEE Antennas Wireless Propag. Lett.*, 7, 411-413.

[16] Hsu, C. H. (2007). Planar multilateral disc monopole antenna for UWB application.

[17] Chen, Z. N., Ammann, M. J., Chia, M. Y. W., & See, T. S. P. (2002). Annular planar monopole antennas. *IEE Proc. Microw. Antennas Propag*, 149(4), 200-203, Aug.

[18] Huang, C. Y., & Hsia, W. C. (2005). Planar elliptical antenna for ultra-wideband com‐

[19] Liang, J., Chiau, C. C., Chen, X., & Parini, C. G. (2005). Study of a printed circular disc monopole antenna for UWB systems. *IEEE Trans. Antennas Propag.*, 53(11), 3500-3504,

[20] Xuan Hui, Wu, & Ahmed, A. Kishk. (2008). Study of an Ultrawideband Omnidirec‐ tional Rolled Monopole Antenna With Trapezoidal Cuts. *IEEE Transactions on Anten‐*

[21] Ahmed, O. M. H., & Sebak, A. R. (2011). Numerical and Experimental Investigation of a Novel Ultrawideband Butterfly Shaped Printed Monopole Antenna with Band‐

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Nov.

#### **Author details**

Osama Haraz1\* and Abdel-Razik Sebak1,2\*

\*Address all correspondence to: osama\_m\_h@yahoo.com

1 Electrical and Computer Engineering Department, Concordia University, Canada

2 KACST Technology Innovation Center in RFTONICS, PSATRI, King Saud University, Sau‐ di Arabia

#### **References**


[8] Su, S. W., Wong, K. L., & Tang, C. L. (2004). Ultra-wideband square planar monopole antenna for IEEE 802.16a operation in the 2-11 GHz band. *Microwave Opt. Tech-nol. Lett.*, 42(6), 463-466, Sept.

**Acknowledgements**

**Author details**

di Arabia

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Osama Haraz1\* and Abdel-Razik Sebak1,2\*

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\*Address all correspondence to: osama\_m\_h@yahoo.com

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1 Electrical and Computer Engineering Department, Concordia University, Canada

2 KACST Technology Innovation Center in RFTONICS, PSATRI, King Saud University, Sau‐

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**Chapter 7**

**Printed Wide Slot Ultra-Wideband Antenna**

With the beginning of the new information era, necessity of wideband wireless communica‐ tions technology is increasing rapidly due to the need to support more users and to provide information with higher data transmitting rates. Ultra-wideband (UWB) technology could be the most suitable technologies that promise to revolutionize high data rate transmission and enable the personal area networking industry leading to new innovations and greater quality of services to the end users. A UWB system is found to be extremely useful and con‐ sists of various satisfying features such as high data rate, high precision ranging, fading ro‐ bustness, and low cost transceiver implementation. UWB is regarded as a very promising and fast emerging low-cost technology with uniquely attractive features inviting major ad‐ vances in wireless communications, sensor networking, radar, imaging, and positioning sys‐

Antennas are indispensable elements of any wireless communication systems. For UWB communication systems, the antennas must be of low profile, compact size, light weight, low cost and conformable to the architecture of the mounting devices. Amongst various types of antennas such as log periodic, TEM horn, stacked patch, spiral and planar structure, the antenna with planar profile seems to be the most preferred choice [3-5]. It has the ad‐ vantage of low profile in size, compactness, and easily embeddable into wireless devices or

In recent years, printed slot antennas are under consideration for use in UWB applications and are getting more and more popular because of the merits of wide frequency bandwidth, low profile, lightweight, ease of fabrication and integration with other devices or RF circui‐ tries. Compared to the electrical antennas, slot antennas have relatively large magnetic fields that tend not to couple strongly with near-by objects, which make them suitable for applica‐ tions wherein near-filed coupling is required to be minimized [6]. A conventional narrow

> © 2013 Azim and Islam; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

© 2013 Azim and Islam; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Rezaul Azim and Mohammad Tariqul Islam

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/51961

**1. Introduction**

tems [1, 2].

integratable with other RF circuitry


### **Printed Wide Slot Ultra-Wideband Antenna**

Rezaul Azim and Mohammad Tariqul Islam

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/51961

#### **1. Introduction**

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[24] Osama, M. H. Ahmed, & Abdel-Razik, Sebak. (2009). A Novel Maple-Leaf Shaped UWB Antenna with a 5.0-6.0 GHz Band-Notch Characteristic. Progress In Electro‐

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for UWB applications. *Microwave Opt. Tech. Lett.*, 40(5), Mar.

With the beginning of the new information era, necessity of wideband wireless communica‐ tions technology is increasing rapidly due to the need to support more users and to provide information with higher data transmitting rates. Ultra-wideband (UWB) technology could be the most suitable technologies that promise to revolutionize high data rate transmission and enable the personal area networking industry leading to new innovations and greater quality of services to the end users. A UWB system is found to be extremely useful and con‐ sists of various satisfying features such as high data rate, high precision ranging, fading ro‐ bustness, and low cost transceiver implementation. UWB is regarded as a very promising and fast emerging low-cost technology with uniquely attractive features inviting major ad‐ vances in wireless communications, sensor networking, radar, imaging, and positioning sys‐ tems [1, 2].

Antennas are indispensable elements of any wireless communication systems. For UWB communication systems, the antennas must be of low profile, compact size, light weight, low cost and conformable to the architecture of the mounting devices. Amongst various types of antennas such as log periodic, TEM horn, stacked patch, spiral and planar structure, the antenna with planar profile seems to be the most preferred choice [3-5]. It has the ad‐ vantage of low profile in size, compactness, and easily embeddable into wireless devices or integratable with other RF circuitry

In recent years, printed slot antennas are under consideration for use in UWB applications and are getting more and more popular because of the merits of wide frequency bandwidth, low profile, lightweight, ease of fabrication and integration with other devices or RF circui‐ tries. Compared to the electrical antennas, slot antennas have relatively large magnetic fields that tend not to couple strongly with near-by objects, which make them suitable for applica‐ tions wherein near-filed coupling is required to be minimized [6]. A conventional narrow

© 2013 Azim and Islam; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Azim and Islam; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

slot antenna has limited bandwidth, whereas wide-slot antennas exhibit wider bandwidth. Recently, different printed wide-slot antennas fed by a microstrip line or coplanar wave‐ guide have been reported [7, 8]. Apart from these antennas, monopole like slot antennas have also been reported to have wide bandwidth characteristics [9-11]. By using different tuning techniques or employing different slot shapes such as rectangle, circle, arc-shape, an‐ nular-ring, U-shaped [12-16], different slot antennas achieved wideband or ultra-wideband performance. A square slot antenna excited by a CPW-fed widened tuning stub was pro‐ posed in [17]. By properly choosing the location and size of the tuning stub, the proposed antenna achieved a bandwidth of 60% with an overall dimension of 72 mm × 72 mm. In [18], a novel broadband design of a CPW-fed square slot antenna loaded with conducting strips has been introduced. The -10 dB impedance bandwidth of the proposed slot antenna is more than 60%. In [19], a printed wide-slot antenna fed by a microstrip line is introduced. By em‐ ploying an arc-shaped slot and a square-patch feed, the antenna achieved an impedance bandwidth ranging from 1.82 GHz to 7.23 GHz. Although the antenna achieved a good im‐ pedance bandwidth with an overall dimension of 110 mm × 110 mm, it does not operate within the entire UWB. The design of a printed wide-slot antenna with a rotated slot is pre‐ sented in [20]. The impedance bandwidth of the antenna varies with the rotation angle of the slot and can maintain 50.2% with suitable angle. More recently, the design of a printed wide-slot antenna for wideband applications is proposed in [21]. The antenna consists of an E-shaped patch and E-shaped slot and achieves an impedance bandwidth of 120% (2.8 - 11.4 GHz). However, the antenna does not possess a compact profile having a dimension of 85 mm × 85 mm. A new CPW-fed tapered ring slot antenna was presented in [22]. With an overall size of 66.1 mm × 44 mm, the proposed antenna achieved an impedance bandwidth range of 8.9 GHz (ranging from 3.1 - 12 GHz). The actual bandwidth was, however, limited by the distortion of radiation patterns.

sists of two sections: the rectangular section of dimension *W3* × *L1* and the triangular section, which is tapered with a slant angle α = 900 for a length *L2* and has strong coupling to the feeding structure. The distance between bottom edge of the tuning stub and lower edge of the tapered shape slot is *h*. Therefore by properly selecting the slot shape and tuning stub, a good impedance bandwidth and radiation characteristics can be achieved. The overall size of the proposed antenna is only 22 mm × 24 mm, which can be considered as one of the

*L* 

Based on this design, some sensitive parameters are studied numerically in order to investi‐ gate the influence of the parameters on antenna performance. In the simulation only one pa‐ rameter was varied each time, where as the others were kept constant. All simulation was

Usually a large slot is used in a wide-slot antenna to achieve a high level of electromagnetic coupling to the tuning stub. Therefore variation of the tuning stub shape and slot shape will change the coupling; and thus control the impedance matching. In order to optimize the coupling between the microstrip-line and the tapered slot, different stub shapes are studied. The rectangular shape tuning stub is compared with four other stubs as shown in Figure 2. Figure 3 shows the simulated return loss curves for the five different stubs. It is observed that, for elliptical and circular shape tuning stubs, the impedance matching become very poor due to poor electromagnetic coupling between the feed-line and tapered slot. The rec‐ tangular shape tuning stub shows a good coupling with tapered shape slot proving a wider

Back View

*W3*

*L1 W1*

**Slot** 

*h* 

*L2*

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 155

*W* 

*w <sup>2</sup>*

smallest UWB slot antenna found in open literature.

Front View

**Figure 1.** Geometry and detailed view of the proposed slot antenna.

**3. Effects of designed parameters**

impedance matching for UWB application.

**3.1. Effect of the tuning stub**

*wf*

*w* 

**Tuning Stub** 

*l* 

carried out by employing Zeland's IE3D based on method of moment [23].

x y

z

In this chapter, a printed wide slot antenna that achieves a physically compact planar profile having sufficient impedance bandwidth and omnidirectional radiation pattern is proposed for UWB communication systems. By etching a microstrip fed rectangular tuning stub as ra‐ diating element and a tapered shape slot in the ground plane, the proposed antenna ach‐ ieved a UWB characteristics. The antenna structure is flat, and its design is simple and easy to fabricate.

#### **2. Antenna configuration**

The geometry and configuration of the proposed antenna is illustrated in Figure 1. The an‐ tenna consists of a tapered shape slot etched out of the ground plane and a microstrip line fed rectangular tuning stub for excitation. The tuning stub fed by microstrip line of 50 Ω characteristics impedance is printed on one side of an inexpensive FR4 substrate of thickness 1.6 mm, with relative permittivity 4.6 and loss tangent 0.02 while the slot is etched out on the other side. The reason for choosing FR4 substrate material is its low cost. Despite of rela‐ tively high loss tangent, the antenna fabricated on FR4 achieved moderate gain and efficient, which are sufficient for UWB wireless communications. The slot in the ground plane con‐ sists of two sections: the rectangular section of dimension *W3* × *L1* and the triangular section, which is tapered with a slant angle α = 900 for a length *L2* and has strong coupling to the feeding structure. The distance between bottom edge of the tuning stub and lower edge of the tapered shape slot is *h*. Therefore by properly selecting the slot shape and tuning stub, a good impedance bandwidth and radiation characteristics can be achieved. The overall size of the proposed antenna is only 22 mm × 24 mm, which can be considered as one of the smallest UWB slot antenna found in open literature.

**Figure 1.** Geometry and detailed view of the proposed slot antenna.

#### **3. Effects of designed parameters**

Based on this design, some sensitive parameters are studied numerically in order to investi‐ gate the influence of the parameters on antenna performance. In the simulation only one pa‐ rameter was varied each time, where as the others were kept constant. All simulation was carried out by employing Zeland's IE3D based on method of moment [23].

#### **3.1. Effect of the tuning stub**

slot antenna has limited bandwidth, whereas wide-slot antennas exhibit wider bandwidth. Recently, different printed wide-slot antennas fed by a microstrip line or coplanar wave‐ guide have been reported [7, 8]. Apart from these antennas, monopole like slot antennas have also been reported to have wide bandwidth characteristics [9-11]. By using different tuning techniques or employing different slot shapes such as rectangle, circle, arc-shape, an‐ nular-ring, U-shaped [12-16], different slot antennas achieved wideband or ultra-wideband performance. A square slot antenna excited by a CPW-fed widened tuning stub was pro‐ posed in [17]. By properly choosing the location and size of the tuning stub, the proposed antenna achieved a bandwidth of 60% with an overall dimension of 72 mm × 72 mm. In [18], a novel broadband design of a CPW-fed square slot antenna loaded with conducting strips has been introduced. The -10 dB impedance bandwidth of the proposed slot antenna is more than 60%. In [19], a printed wide-slot antenna fed by a microstrip line is introduced. By em‐ ploying an arc-shaped slot and a square-patch feed, the antenna achieved an impedance bandwidth ranging from 1.82 GHz to 7.23 GHz. Although the antenna achieved a good im‐ pedance bandwidth with an overall dimension of 110 mm × 110 mm, it does not operate within the entire UWB. The design of a printed wide-slot antenna with a rotated slot is pre‐ sented in [20]. The impedance bandwidth of the antenna varies with the rotation angle of the slot and can maintain 50.2% with suitable angle. More recently, the design of a printed wide-slot antenna for wideband applications is proposed in [21]. The antenna consists of an E-shaped patch and E-shaped slot and achieves an impedance bandwidth of 120% (2.8 - 11.4 GHz). However, the antenna does not possess a compact profile having a dimension of 85 mm × 85 mm. A new CPW-fed tapered ring slot antenna was presented in [22]. With an overall size of 66.1 mm × 44 mm, the proposed antenna achieved an impedance bandwidth range of 8.9 GHz (ranging from 3.1 - 12 GHz). The actual bandwidth was, however, limited

In this chapter, a printed wide slot antenna that achieves a physically compact planar profile having sufficient impedance bandwidth and omnidirectional radiation pattern is proposed for UWB communication systems. By etching a microstrip fed rectangular tuning stub as ra‐ diating element and a tapered shape slot in the ground plane, the proposed antenna ach‐ ieved a UWB characteristics. The antenna structure is flat, and its design is simple and easy

The geometry and configuration of the proposed antenna is illustrated in Figure 1. The an‐ tenna consists of a tapered shape slot etched out of the ground plane and a microstrip line fed rectangular tuning stub for excitation. The tuning stub fed by microstrip line of 50 Ω characteristics impedance is printed on one side of an inexpensive FR4 substrate of thickness 1.6 mm, with relative permittivity 4.6 and loss tangent 0.02 while the slot is etched out on the other side. The reason for choosing FR4 substrate material is its low cost. Despite of rela‐ tively high loss tangent, the antenna fabricated on FR4 achieved moderate gain and efficient, which are sufficient for UWB wireless communications. The slot in the ground plane con‐

by the distortion of radiation patterns.

154 Advancement in Microstrip Antennas with Recent Applications

**2. Antenna configuration**

to fabricate.

Usually a large slot is used in a wide-slot antenna to achieve a high level of electromagnetic coupling to the tuning stub. Therefore variation of the tuning stub shape and slot shape will change the coupling; and thus control the impedance matching. In order to optimize the coupling between the microstrip-line and the tapered slot, different stub shapes are studied. The rectangular shape tuning stub is compared with four other stubs as shown in Figure 2. Figure 3 shows the simulated return loss curves for the five different stubs. It is observed that, for elliptical and circular shape tuning stubs, the impedance matching become very poor due to poor electromagnetic coupling between the feed-line and tapered slot. The rec‐ tangular shape tuning stub shows a good coupling with tapered shape slot proving a wider impedance matching for UWB application.

improved by employing tapered slot structure, and a tapered-shape slot matched with a rec‐ tangular tuning stub can produce wider bandwidth than with a circular, elliptical, and

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 157

(a) (b) (c) (d)

2 3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

The gap between the slot and the ground plane determines the matching between the feed line and slot antenna. The effect of feed gap on the impedance matching was investigated in [7] and [21]. It was found that by enhancing the coupling between the slot and microstrip feed line, good impedance matching can be obtained. An optimum value of the impedance bandwidth can be obtained for a certain optimum value of coupling. However, if the cou‐ pling increases further from this optimum value, the impedance matching becomes worse due to over-coupling. Figure 6 shows the simulated results of the proposed antenna for dif‐ ferent feed gaps of -0.25, 0, 0.75 and 1.25 mm. It can be observed from the Figure that lower edge frequency of the operating band is highly dependent on the feed gap, while the feed

**Figure 4.** Different slot shape (a) Circular, (b) Elliptical, (c) Square and (d) Tapered.

Circular Elliptical Square Tapered Without slot


**3.3. Effect of feed gap**

**Figure 5.** Simulated return loss curves for different tuning slot shape.

Return loss(dB)

square-shaped slot.

**Figure 2.** Different tuning stub shape (a) Rectangular (b) Circular, (c) Square, (d) Elliptical and (e) Tapered.

**Figure 3.** Simulated return loss curves for different tuning stub shape

#### **3.2. Effect of slot shape**

The wide-slot antenna is well-known to have wide impedance bandwidth though its operat‐ ing bandwidth is limited due to the degradation of the radiation patterns at higher frequen‐ cies [7]. Through the numerical study on different slot shapes as shown in Figure 4, it is seen that currents flowing on the edge of the slot will increase the cross-polarization component in the *yz*-plane and cause the main beam to tilt away from the broadside direction in the *xz*plane. Unlike the conventional wide-slot antenna proposed in [17], the slot in the ground plane of the proposed antenna with tapered shape is surrounded by ground strips of small width, which makes the antenna very compact. Moreover, introduction of the tapered slot instead of the rectangular slot changes the electric field distribution by reducing the longest current path and reducing the slot size. As a result, the impedance matching is much im‐ proved, especially at lower frequencies, resulting in overall enhancement of operating band‐ width as shown in Figure 5. It is also observed that high-frequency performance can also be improved by employing tapered slot structure, and a tapered-shape slot matched with a rec‐ tangular tuning stub can produce wider bandwidth than with a circular, elliptical, and square-shaped slot.

**Figure 4.** Different slot shape (a) Circular, (b) Elliptical, (c) Square and (d) Tapered.

**Figure 5.** Simulated return loss curves for different tuning slot shape.

#### **3.3. Effect of feed gap**

(a) (b) (c) (d) (e)

2 3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

The wide-slot antenna is well-known to have wide impedance bandwidth though its operat‐ ing bandwidth is limited due to the degradation of the radiation patterns at higher frequen‐ cies [7]. Through the numerical study on different slot shapes as shown in Figure 4, it is seen that currents flowing on the edge of the slot will increase the cross-polarization component in the *yz*-plane and cause the main beam to tilt away from the broadside direction in the *xz*plane. Unlike the conventional wide-slot antenna proposed in [17], the slot in the ground plane of the proposed antenna with tapered shape is surrounded by ground strips of small width, which makes the antenna very compact. Moreover, introduction of the tapered slot instead of the rectangular slot changes the electric field distribution by reducing the longest current path and reducing the slot size. As a result, the impedance matching is much im‐ proved, especially at lower frequencies, resulting in overall enhancement of operating band‐ width as shown in Figure 5. It is also observed that high-frequency performance can also be

**Figure 2.** Different tuning stub shape (a) Rectangular (b) Circular, (c) Square, (d) Elliptical and (e) Tapered.

Circular Elliptical Rectangular Square Tapered


**Figure 3.** Simulated return loss curves for different tuning stub shape

Return loss(dB)

156 Advancement in Microstrip Antennas with Recent Applications

**3.2. Effect of slot shape**

The gap between the slot and the ground plane determines the matching between the feed line and slot antenna. The effect of feed gap on the impedance matching was investigated in [7] and [21]. It was found that by enhancing the coupling between the slot and microstrip feed line, good impedance matching can be obtained. An optimum value of the impedance bandwidth can be obtained for a certain optimum value of coupling. However, if the cou‐ pling increases further from this optimum value, the impedance matching becomes worse due to over-coupling. Figure 6 shows the simulated results of the proposed antenna for dif‐ ferent feed gaps of -0.25, 0, 0.75 and 1.25 mm. It can be observed from the Figure that lower edge frequency of the operating band is highly dependent on the feed gap, while the feed gap has a little effect on the upper edge frequencies. It is also observed that a feed gap of 0.75 mm can give the widest operating band with good return loss values. Table 1 is there‐ fore represents a good summary of the optimized parameters of the proposed antenna for achieving the ultra-wide impedance bandwidth.

**4. Antenna performance and characteristics**

**4.1. Input impedance characteristics and current distribution**

Front view

2 3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

Simulated Measured

Back

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 159

E8362C vector network analyzer.

ultra-wide frequency band.

**Figure 7.** Photograph of the realized antenna.


**Figure 8.** Simulated and measured return loss curves of the proposed antenna.

Return loss(dB)

A prototype of tapered shape slot antenna with optimal parameters tabulated in Table 1 is fabricated for experimental verification as shown in Figure 7. The antenna performance is measured in an anechoic chamber using Satimo's antenna measurement system and Agilent

The measured and simulated return loss curves of the proposed antenna are depicted in Fig‐ ure 8. It is seen that the proposed antenna exhibits a wideband performance from 3 to 11.2 GHz (115.5%) for -10 dB return loss value. The measured result agrees reasonably with the simulated one across the whole operating band. The disagreement between simulation and measurement is mainly due to the fabrication tolerance. It may also be due to the effect of the feeding cable as the antenna is small. Despite being physically small than the antenna proposed in [7, 13, 17, 20, 21], the antenna still achieved wide bandwidth to cover the entire

**Figure 6.** Simulated return loss curves for different feed gap.


**Table 1.** Optimized antenna parameters

#### **4. Antenna performance and characteristics**

gap has a little effect on the upper edge frequencies. It is also observed that a feed gap of 0.75 mm can give the widest operating band with good return loss values. Table 1 is there‐ fore represents a good summary of the optimized parameters of the proposed antenna for

> 2 3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

**Parameter Value(mm)** *W* 22

*L* 24

*W1* 2

*L1* 10.75

*w2* 11

*L2* 7

*W3* 18

*w* 13

*l* 7

*wf* 3

α 900

*h* 0.75


achieving the ultra-wide impedance bandwidth.

158 Advancement in Microstrip Antennas with Recent Applications


**Figure 6.** Simulated return loss curves for different feed gap.

**Table 1.** Optimized antenna parameters

Return loss(dB)

A prototype of tapered shape slot antenna with optimal parameters tabulated in Table 1 is fabricated for experimental verification as shown in Figure 7. The antenna performance is measured in an anechoic chamber using Satimo's antenna measurement system and Agilent E8362C vector network analyzer.

#### **4.1. Input impedance characteristics and current distribution**

The measured and simulated return loss curves of the proposed antenna are depicted in Fig‐ ure 8. It is seen that the proposed antenna exhibits a wideband performance from 3 to 11.2 GHz (115.5%) for -10 dB return loss value. The measured result agrees reasonably with the simulated one across the whole operating band. The disagreement between simulation and measurement is mainly due to the fabrication tolerance. It may also be due to the effect of the feeding cable as the antenna is small. Despite being physically small than the antenna proposed in [7, 13, 17, 20, 21], the antenna still achieved wide bandwidth to cover the entire ultra-wide frequency band.

view

**Figure 7.** Photograph of the realized antenna.

**Figure 8.** Simulated and measured return loss curves of the proposed antenna.

It is observed from the return loss curve that the proposed antenna is capable of supporting multiple resonances. The first resonance emerges at around 3.4 GHz, the second resonance at 6 GHz, third resonance at 8 GHz and fourth one at 10 GHz. The overlapping of these reso‐ nances, which are closely spaced over the spectrum leads to an ultra-wide operating band, which support the principle presented in [24].

The input impedance of proposed antenna is shown in Figure 9. Though there is variation in the frequency range from 5 - 8 GHz, it is seen in the Figure that the resistance is nearly flat and tends to 50 Ω values while the reactance is relatively constant at 0 Ω. Moreover, at these frequencies the measured phase of the input impedance is almost linear, which ensure that all the frequency components of the signal have the same delay leading to less pulse distor‐ tions. It is also seen that at the higher frequency end the resistance and reactance are getting away from 50 Ω and 0 Ω lines, respectively, i.e. the impedance matching is getting worse.

(c)

**Figure 10.** Simulated current distributions at (a) 3.4, (b) 6, (c) 8 and (d) 10 GHz.

(a)

(d)

(b)

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 161

Am-1

Satimo Starlab 0.6–18 GHz anechoic chamber at University of Hong Kong is used for the measurements of gain, total antenna efficiency, and radiation pattern [27]. This system uses the near-field measurement techniques that allow measurement of electric fields within the near-field of the antenna to calculate the equivalent far-field data of the antenna under test. The near-field of an antenna is the area close to the antenna, where the electric charge and electromagnetic induction effects occur. These effects fade out far more rapidly with increas‐ ing distance from the antenna (proportional to the cube of the distance) than the radiated electromagnetic far-field that fades out proportional to the distance. Near-field effects be‐ come negligible more than a few wavelengths away from the antenna. Once the near-field data have been measured, a Fourier transformation is used to calculate the equivalent farfield data. The antenna, mounted on the test board, is positioned in the center of a circular "arch" that contains 16 separate measurement probes. These probes are spaced equally apart

Am-1

Am-1

**4.2. Radiation characteristics**

Am-1

**Figure 9.** Input impedance and phase of the proposed slot antenna.

The return loss curve or the input impedance can only illustrate the antenna performance as a lumped load at the end of microstrip line [25]. The electromagnetic characteristics of the antenna can only be understood by examining the current distributions behavior at reso‐ nance frequencies. Simulated surface current distributions on the antenna close to the reso‐ nance frequencies are depicted in Figure 10. Figure 10(a) shows the current distribution at first resonance frequency of 3.4 GHz. The current pattern near the second resonance at around 6 GHz is shown in Figure 10(b), representing approximately a second order harmon‐ ic. Figure 10(c) present third order harmonic at 8 GHz. Figure 10(d) plots a more complicat‐ ed current pattern at 10 GHz, corresponding to the fourth order harmonic. These current distributions is also support the principle that the overlapping of closely spaced resonances resulting in UWB characterization of the proposed antenna. At these four frequencies the resonances are clearly observed on the edges of both the tapered shape slot and rectangular tuning stub.

**Figure 10.** Simulated current distributions at (a) 3.4, (b) 6, (c) 8 and (d) 10 GHz.

#### **4.2. Radiation characteristics**

It is observed from the return loss curve that the proposed antenna is capable of supporting multiple resonances. The first resonance emerges at around 3.4 GHz, the second resonance at 6 GHz, third resonance at 8 GHz and fourth one at 10 GHz. The overlapping of these reso‐ nances, which are closely spaced over the spectrum leads to an ultra-wide operating band,

The input impedance of proposed antenna is shown in Figure 9. Though there is variation in the frequency range from 5 - 8 GHz, it is seen in the Figure that the resistance is nearly flat and tends to 50 Ω values while the reactance is relatively constant at 0 Ω. Moreover, at these frequencies the measured phase of the input impedance is almost linear, which ensure that all the frequency components of the signal have the same delay leading to less pulse distor‐ tions. It is also seen that at the higher frequency end the resistance and reactance are getting away from 50 Ω and 0 Ω lines, respectively, i.e. the impedance matching is getting worse.

> Reistance Reactance Phase

2 4 6 8 10 12 Frequency(GHz)

The return loss curve or the input impedance can only illustrate the antenna performance as a lumped load at the end of microstrip line [25]. The electromagnetic characteristics of the antenna can only be understood by examining the current distributions behavior at reso‐ nance frequencies. Simulated surface current distributions on the antenna close to the reso‐ nance frequencies are depicted in Figure 10. Figure 10(a) shows the current distribution at first resonance frequency of 3.4 GHz. The current pattern near the second resonance at around 6 GHz is shown in Figure 10(b), representing approximately a second order harmon‐ ic. Figure 10(c) present third order harmonic at 8 GHz. Figure 10(d) plots a more complicat‐ ed current pattern at 10 GHz, corresponding to the fourth order harmonic. These current distributions is also support the principle that the overlapping of closely spaced resonances resulting in UWB characterization of the proposed antenna. At these four frequencies the resonances are clearly observed on the edges of both the tapered shape slot and rectangular


Phase(Degree)

which support the principle presented in [24].

160 Advancement in Microstrip Antennas with Recent Applications

**Figure 9.** Input impedance and phase of the proposed slot antenna.

Impedance(Ohm)

tuning stub.

Satimo Starlab 0.6–18 GHz anechoic chamber at University of Hong Kong is used for the measurements of gain, total antenna efficiency, and radiation pattern [27]. This system uses the near-field measurement techniques that allow measurement of electric fields within the near-field of the antenna to calculate the equivalent far-field data of the antenna under test. The near-field of an antenna is the area close to the antenna, where the electric charge and electromagnetic induction effects occur. These effects fade out far more rapidly with increas‐ ing distance from the antenna (proportional to the cube of the distance) than the radiated electromagnetic far-field that fades out proportional to the distance. Near-field effects be‐ come negligible more than a few wavelengths away from the antenna. Once the near-field data have been measured, a Fourier transformation is used to calculate the equivalent farfield data. The antenna, mounted on the test board, is positioned in the center of a circular "arch" that contains 16 separate measurement probes. These probes are spaced equally apart along the circular surface. The antenna is rotated horizontally through 360°, and the combi‐ nation of this rotation and the array of probes allows a full 3D scan of the antenna to be car‐ ried out, allowing full 3D radiation patterns to be measured, plotted, and analyzed. Information about antenna gain and efficiency can then be calculated from the far-field radi‐ ation pattern data. A coaxial cable was incorporated in the measurement system and the sys‐ tem was calibrated.

order harmonic introduced to patterns and both *H*- and *E*-plane become more directional but still retain omni-directionality. Polarization purity can only be seen at low frequency re‐ gion where the cross-to-co polarization ratio is around -20 dB, in contrast to higher frequen‐ cies, where the cross-polarization is dominant especially in *H*-plane. The slight asymmetry observed at higher frequencies in both in *H* -and *E* -plane may be due to the fact that micro‐

45

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45

135

90

**Figure 12.** Measured radiation patterns at 8 GHz and 10 GHz [solid line: co-polarization, crossed line: cross-polariza‐

Figure 13 depict the measured 3D radiation patterns at 3.4 and 8 GHz. In the patterns the red color indicates the stronger radiated *E*-field and the blue is the weakest ones. At low fre‐ quency of 3.4 GHz, the radiation patterns are almost omni-directional similar to a typical monopole antenna. The radiation is slightly weak in *z*-direction. At the higher frequency of 8 GHz, the radiation becomes slightly directional with a null in the *z-*direction due to higher order harmonics. The 3D omni-directional radiation patterns of the proposed antenna make

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(d) *H*-plane at 10 GHz

(b) *H*-plane at 8 GHz

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strip feed line itself act as a radiator.

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tion].

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(a) *E*-plane at 8 GHz


180

(c) *E*-plane at 10 GHz

**Figure 11.** Measured radiation patterns at 3.4 GHz and 6 GHz [solid line: co-polarization, crossed line: cross-polariza‐ tion].

Figures 11 and 12 show the measured 2D radiation patterns in two principal planes-namely, the *E-*(*xz*) and *H*-(*xy*) planes for four resonant frequencies of 3.4, 6, 8 and 10 GHz. It can be observed that at lower frequencies the antenna exhibit an omni-directional radiation pat‐ terns for *H*-plane and donut shape for *E*-plane with low cross-polarization field and patterns are about the same as that of a typical monopole antenna. As the frequency increases, higher order harmonic introduced to patterns and both *H*- and *E*-plane become more directional but still retain omni-directionality. Polarization purity can only be seen at low frequency re‐ gion where the cross-to-co polarization ratio is around -20 dB, in contrast to higher frequen‐ cies, where the cross-polarization is dominant especially in *H*-plane. The slight asymmetry observed at higher frequencies in both in *H* -and *E* -plane may be due to the fact that micro‐ strip feed line itself act as a radiator.

along the circular surface. The antenna is rotated horizontally through 360°, and the combi‐ nation of this rotation and the array of probes allows a full 3D scan of the antenna to be car‐ ried out, allowing full 3D radiation patterns to be measured, plotted, and analyzed. Information about antenna gain and efficiency can then be calculated from the far-field radi‐ ation pattern data. A coaxial cable was incorporated in the measurement system and the sys‐

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270

**Figure 11.** Measured radiation patterns at 3.4 GHz and 6 GHz [solid line: co-polarization, crossed line: cross-polariza‐

Figures 11 and 12 show the measured 2D radiation patterns in two principal planes-namely, the *E-*(*xz*) and *H*-(*xy*) planes for four resonant frequencies of 3.4, 6, 8 and 10 GHz. It can be observed that at lower frequencies the antenna exhibit an omni-directional radiation pat‐ terns for *H*-plane and donut shape for *E*-plane with low cross-polarization field and patterns are about the same as that of a typical monopole antenna. As the frequency increases, higher

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(b) *H*-plane at 3.4 GHz

180

(d) *H*-plane at 6 GHz

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tem was calibrated.

270


162 Advancement in Microstrip Antennas with Recent Applications

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tion].

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(a) *E*-plane at 3.4 GHz


180

(c) *E*-plane at 6 GHz

**Figure 12.** Measured radiation patterns at 8 GHz and 10 GHz [solid line: co-polarization, crossed line: cross-polariza‐ tion].

Figure 13 depict the measured 3D radiation patterns at 3.4 and 8 GHz. In the patterns the red color indicates the stronger radiated *E*-field and the blue is the weakest ones. At low fre‐ quency of 3.4 GHz, the radiation patterns are almost omni-directional similar to a typical monopole antenna. The radiation is slightly weak in *z*-direction. At the higher frequency of 8 GHz, the radiation becomes slightly directional with a null in the *z-*direction due to higher order harmonics. The 3D omni-directional radiation patterns of the proposed antenna make it suitable for being used in different wireless communications especially in mobile commu‐ nication.

**4.3. Gain and radiation efficiency**


and is similar to those proposed in [17] and [26].

Efficiency(%)

Gain(dBi)

**Figure 14.** Measured peak antenna gain.

**Figure 15.** Measured radiation efficiency.

rectional.

The measured peak gain of the proposed slot antenna is shown in Figure 14. From the Fig‐ ure, it can be seen that the proposed antenna achieves an average peak gain of 3.81 dBi. The maximum realized gain is 5.4 dBi at 9.8 GHz, where the radiation patterns become slight di‐

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 165

3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

3 4 5 6 7 8 9 10 11 12 Frequency(GHz)

The realized radiation efficiency of the proposed antenna is shown in Figure 15. It is seen that the fabricated antenna achieves an average radiation efficiency of 77.9% and the maxi‐ mum efficiency is 88.2%. Despite of fluctuations observed in the curves due to wider band‐ width, the proposed slot antenna achieves good gain and radiation efficiency with a compact profile in comparison with the other reported microstrip line fed planar antennas

**Figure 13.** Measured 3D radiation pattern in *xy*, *yz* and *xz*-planes at (a) 3.4 and (b) 8 GHz.

#### **4.3. Gain and radiation efficiency**

it suitable for being used in different wireless communications especially in mobile commu‐

**Figure 13.** Measured 3D radiation pattern in *xy*, *yz* and *xz*-planes at (a) 3.4 and (b) 8 GHz.

nication.

164 Advancement in Microstrip Antennas with Recent Applications

The measured peak gain of the proposed slot antenna is shown in Figure 14. From the Fig‐ ure, it can be seen that the proposed antenna achieves an average peak gain of 3.81 dBi. The maximum realized gain is 5.4 dBi at 9.8 GHz, where the radiation patterns become slight di‐ rectional.

**Figure 14.** Measured peak antenna gain.

**Figure 15.** Measured radiation efficiency.

The realized radiation efficiency of the proposed antenna is shown in Figure 15. It is seen that the fabricated antenna achieves an average radiation efficiency of 77.9% and the maxi‐ mum efficiency is 88.2%. Despite of fluctuations observed in the curves due to wider band‐ width, the proposed slot antenna achieves good gain and radiation efficiency with a compact profile in comparison with the other reported microstrip line fed planar antennas and is similar to those proposed in [17] and [26].

#### **5. Time domain behavior**

Since UWB systems directly transmit narrow pulses rather than continuous wave, the time domain performances of the UWB antenna is very crucial. A good time domain performan‐ ces is a primary requirement of UWB antenna. The antenna features can be optimized to avoid undesired pulse distortions. For a transmitting/receiving antenna system as shown in Fig. 16(a), the transfer function (S21 parameter) is required to have flat magnitude and linear phase response over the operating band to minimize the distortions in the received signal waveform and is defined as [28-30]

$$T(\theta,\phi,\alpha) = \frac{\sqrt{\eta\_0 Z\_0}}{Z\_0 + Z\_A(\alpha)} \vec{h}\_{\text{eff}}(\theta,\phi,\alpha) \tag{1}$$

Vector Network Analyzer

*d* = 0.5 m

Antenna System

Transmitting Antenna Receiving Antenna

(a)

(b)

**Figure 16.** Setup for transfer function and group delay measurement (a) schematic diagram and (b) in anechoic

3 4 5 6 7 8 9 10 11 Frequency(GHz)


Phase(degree)

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 167

chamber.


**Figure 17.** Magnitude and phase of the measured transfer function.




Magnitude(dB)



where *h* → *eff* is the complex vector effective height and *ZA*(*ω*)is the antenna input impedance. In terms of this transfer function the port-to-port S21 between transmitting and receiving an‐ tennas is

$$\mathcal{S}\_{21}(oo) = j o \vec{I}\_{TX}(oo) \bullet \vec{T}\_{RX}(oo) \frac{e^{-j\mathbb{R}R}}{2\pi R C\_0} \tag{2}$$

where ω=2π*f*, *f* is the operating frequency, *C0* is the velocity of light, (*θ*, *φ*) is the orientation and *T* → *TX* and *T* → *RX* are the transfer functions of the transmitting and receiving antennas re‐ spectively. If S21 is measured using two identical antennas, for single polarization, the trans‐ fer can be calculated as

$$T(oo) = \sqrt{\frac{2\pi Rc\_0}{jao} S\_{21}(oo)e^{jaoR} \Big|\_{C\_0}}\tag{3}$$

The distance, *R* is to be derived from the S21 data itself.

The group delay is defined as the negative derivative of the phase response with respect to frequency and usually used to evaluate the phase response of the transfer function. The group delay gives an indication of the time delay of an impulse signal at different frequen‐ cies. Ideally, when the phase response is strictly linear, the group delay variation is zero. The transfer functions and group delay between a pair of proposed antennas had been measured inside an anechoic chamber with dimension of 4 m × 4 m × 8 m using Satimo's StarLab antenna measuring equipment. Since UWB technology employed in short range communication systems, in the measurements the transmitting and receiving antennas are placed face-to face at distance 0.5 m apart as illustrates in Figure 16. The measurements were taken at different azimuth angle in *xz*-plane.

**5. Time domain behavior**

166 Advancement in Microstrip Antennas with Recent Applications

waveform and is defined as [28-30]

where *h* →

tennas is

and *T* → *TX* and *T* →

fer can be calculated as

Since UWB systems directly transmit narrow pulses rather than continuous wave, the time domain performances of the UWB antenna is very crucial. A good time domain performan‐ ces is a primary requirement of UWB antenna. The antenna features can be optimized to avoid undesired pulse distortions. For a transmitting/receiving antenna system as shown in Fig. 16(a), the transfer function (S21 parameter) is required to have flat magnitude and linear phase response over the operating band to minimize the distortions in the received signal

> 0 0 0 (,, ) (,, ) ( ) *eff A Z*

w

In terms of this transfer function the port-to-port S21 between transmitting and receiving an‐

 w p

where ω=2π*f*, *f* is the operating frequency, *C0* is the velocity of light, (*θ*, *φ*) is the orientation

spectively. If S21 is measured using two identical antennas, for single polarization, the trans‐

<sup>21</sup> <sup>0</sup>

w

*j C*

The group delay is defined as the negative derivative of the phase response with respect to frequency and usually used to evaluate the phase response of the transfer function. The group delay gives an indication of the time delay of an impulse signal at different frequen‐ cies. Ideally, when the phase response is strictly linear, the group delay variation is zero. The transfer functions and group delay between a pair of proposed antennas had been measured inside an anechoic chamber with dimension of 4 m × 4 m × 8 m using Satimo's StarLab antenna measuring equipment. Since UWB technology employed in short range communication systems, in the measurements the transmitting and receiving antennas are placed face-to face at distance 0.5 m apart as illustrates in Figure 16. The measurements were

 w

æ ö <sup>=</sup> ç ÷

r

qjw

0

*RX* are the transfer functions of the transmitting and receiving antennas re‐

è ø (3)

*jkR*


*eff* is the complex vector effective height and *ZA*(*ω*)is the antenna input impedance.

(1)

(2)

*T h Z Z* h

<sup>=</sup> <sup>+</sup>

() () () <sup>2</sup>

= · r r

*TX RX <sup>e</sup> S jT T RC* www

0

<sup>2</sup> ( ) ( ) *Rc j R T Se*

w

p

qjw

21

w

The distance, *R* is to be derived from the S21 data itself.

taken at different azimuth angle in *xz*-plane.

(b)

**Figure 16.** Setup for transfer function and group delay measurement (a) schematic diagram and (b) in anechoic chamber.

**Figure 17.** Magnitude and phase of the measured transfer function.

The magnitude and phase of the measured transfer function of the proposed antenna are shown in Figure 17. It is observed that the magnitudes of the transfer function are relatively smooth over the whole UWB frequency range and the variation is less than 10 dB. Linear phase response is also observed within the frequency range from 3 - 10 GHz as depicted in Figure 17. The measured group delay as shown in Figure 18 demonstrates relatively con‐ stant responses over the entire UWB frequency band. The average variation in the group de‐ lay is less than 1.3 ns, which corresponds very well to the phase of the transfer functions. This small variation in transfer function indicates that the proposed antenna does not distort the phase of the transmitted/received signals, which is a primary requirement of UWB appli‐ cations.

MO's StarLab antenna measurement equipment of Department of Electrical and Electronic

Printed Wide Slot Ultra-Wideband Antenna http://dx.doi.org/10.5772/51961 169

1 Institute of Space Science (ANGKASA), Universiti Kebangsaan Malaysia, Malaysia

[1] Allen, B., Dohler, M., Okon, E., Malik, W., Brown, A., Edwards, D. *Ultra-wideband An‐ tennas and Propagation for Communications, Radar and Imaging*. New York: John Wiley

[2] Chen, C. C., Rao, K. R., Lee, R. A New Ultrawide-bandwidth Dielectric-rod Antenna for Ground-Penetrating Radar Application. *IEEE Transactions on Antennas & Propaga‐*

[3] Chang, L. T., Burnside, W. D. An Ultrawide-Bandwidth Tapered Resistive TEM Horn Antenna. *IEEE Transactions on Antennas & Propagation*, 48, 1848–1857(2000).

[4] Shakib, M. N., Islam, M. T., Misran, N. Stacked Patch Antenna with Folded Patch Feed for Ultra-wideband Application. *IET Microwave Antennas & Propagation*, 4, 1456–

[5] Azim, R., Islam, M. T., Misran, N., Cheung, S. W., Yamada, Y. Planar UWB Antenna with Multi-Slotted Ground Plane. *Microwave & Optical Technology Letters*, 53, 966-968

[6] Schantz, H. G. UWB Magnetic Antennas. *Proceedings of the IEEE Antennas & Propaga‐ tion Society International Symposium*, 22-27 June (2003), Columbus, Ohio, USA, 604 -

[7] Liu, Y. F., Lan, K. L., Xue, Q., Chan, C. H. Experimental Studies of Printed Wide-slot Antenna for Wide-band Applications. *IEEE Antennas & Wireless Propagation Letters*, 3,

[8] Qu, S. W., Ruan, C., Wang, B. Z. Bandwidth Enhancement of Wide-slot Antenna fed by CPW and Microstrip Line. *IEEE Antennas & Wireless Propagation Letters*, 5, 15 – 17

Engineering, University of Hong Kong.

and Mohammad Tariqul Islam1,2

2 Department of Physics, University of Chittagong, Bangladesh

**Author details**

Rezaul Azim1

**References**

& Sons, Ltd. (2006).

*tion*, 51, 371-377(2003).

1461(2010).

(2011).

607.

(2006).

273 – 275 (2004).

**Figure 18.** Measured group delay of the proposed antenna.

#### **6. Conclusion**

The design of a compact printed wide slot antenna has been proposed and implemented for ultra-wideband applications. The proposed antenna consist of a tapered shape slot and rec‐ tangular tuning stub, and fabricated onto a 22 mm × 24 mm× 1.6 mm size FR4 dielectric sub‐ strate. The measured results show that the proposed antenna achieves good impedance matching constant gain, stable radiation patterns over an operating bandwidth of 3 to 11.2 GHz (115.5%) to cover the entire UWB. The stable radiation pattern with a maximum gain of 5.4 dBi and good time domain behaviors makes the proposed antenna a suitable candidate for practical UWB applications.

#### **Acknowledgements**

This work is funded by Universiti Kebangsaan Malaysia under the grant DIP-2012-06. The authors would like to thank Associate Professor S. W. Cheung for allowing using the SATI‐ MO's StarLab antenna measurement equipment of Department of Electrical and Electronic Engineering, University of Hong Kong.

#### **Author details**

The magnitude and phase of the measured transfer function of the proposed antenna are shown in Figure 17. It is observed that the magnitudes of the transfer function are relatively smooth over the whole UWB frequency range and the variation is less than 10 dB. Linear phase response is also observed within the frequency range from 3 - 10 GHz as depicted in Figure 17. The measured group delay as shown in Figure 18 demonstrates relatively con‐ stant responses over the entire UWB frequency band. The average variation in the group de‐ lay is less than 1.3 ns, which corresponds very well to the phase of the transfer functions. This small variation in transfer function indicates that the proposed antenna does not distort the phase of the transmitted/received signals, which is a primary requirement of UWB appli‐

> 3 4 5 6 7 8 9 10 11 Frequency(GHz)

The design of a compact printed wide slot antenna has been proposed and implemented for ultra-wideband applications. The proposed antenna consist of a tapered shape slot and rec‐ tangular tuning stub, and fabricated onto a 22 mm × 24 mm× 1.6 mm size FR4 dielectric sub‐ strate. The measured results show that the proposed antenna achieves good impedance matching constant gain, stable radiation patterns over an operating bandwidth of 3 to 11.2 GHz (115.5%) to cover the entire UWB. The stable radiation pattern with a maximum gain of 5.4 dBi and good time domain behaviors makes the proposed antenna a suitable candidate

This work is funded by Universiti Kebangsaan Malaysia under the grant DIP-2012-06. The authors would like to thank Associate Professor S. W. Cheung for allowing using the SATI‐


Delay(ns)

168 Advancement in Microstrip Antennas with Recent Applications

**Figure 18.** Measured group delay of the proposed antenna.

cations.

**6. Conclusion**

for practical UWB applications.

**Acknowledgements**

Rezaul Azim1 and Mohammad Tariqul Islam1,2


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170 Advancement in Microstrip Antennas with Recent Applications


**Chapter 8**

**Recent Trends in Printed Ultra-Wideband (UWB)**

After the Federal Communication Commission (FCC)'s authorization of frequency band of 3.1 to 10.6 GHz for unlicensed radio applications, ultra-wideband (UWB) technology be‐ come the most promising candidate for a wide range of applications that will provide signif‐ icant benefits for public safety, business and consumers, and attracted a lot attention both in industry and academia. The antennas are the key components of UWB system. In wireless communication system, an antenna can take various forms to fulfill the particular require‐ ment. As a result, an antenna may be a piece of conducting wire, an aperture, a patch, a re‐ flector, a lens, an assembly of elements (arrays). A good design of the antenna can fulfill the

Over the past few years, significant research efforts have been put into the design of UWB antennas and systems for communications. The UWB antenna is essential for providing wi‐ deband wireless communications based on the use of very narrow pulses on the order of nanoseconds, covering an ultra-wide bandwidth in the frequency domain, and over short distance at very low spectral power densities. In addition, the antennas required to have a non-dispersive characteristic in time and frequency, providing a narrow, pulse duration to enhance a high data throughput [1]. Different kinds of antennas suitable for use in UWB ap‐ plications have proposed in past few decade, each with its advantages and disadvantages. In this paper, a review on printed UWB antennas has been done historically. Then, a techni‐ que to miniaturize the antenna's physical size by shrinking the ground plane is proposed. To develop the design technique by which the antennas can be able to achieve both UWB operating bandwidth and the stable radiation pattern across the entire frequency band by reducing the ground plane effect is also described. Finally, the enhancement of operating bandwidth as well as the pattern bandwidth by further modified the ground plane is ach‐

> © 2013 Islam and Azim; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

© 2013 Islam and Azim; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Mohammad Tariqul Islam and Rezaul Azim

Additional information is available at the end of the chapter

system requirements and improve overall system performance.

ieved in order to fulfill the requirements defined by the FCC.

**Antennas**

http://dx.doi.org/10.5772/52056

**1. Introduction**
