**Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band**

Kazuyuki Seo

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/54662

### **1. Introduction**

Many kind of millimeter-wave automotive radars have been developed [1], [2]. The microstrip antenna becomes a good candidate when radar sensors are widely used in vehicle due to its advantages of low cost and low profile. Generally microstrip antennas are placed on the surface of a radar sensor and are connected to millimeter-wave circuits inside of the sensor via waveguides. Therefore, transitions from waveguide to microstrip line are required, as shown in Figure 1.

Rectangular waveguides were one of the earliest types of transmission lines used to transport microwave signals and are still used today for many applications. Because of the recent trend toward miniaturization and integration, a lot of microwave circuitry is currently fabricated using planar transmission lines, such as microstrip or strip line, rather than waveguide. There is, however, still a need for waveguides in many applications such as millimeter wave systems, and in some precision test applications.

Various types of millimeter-wave transitions from waveguide to microstrip line have been proposed. The ridge waveguide type [3], quasi-Yagi type [4], and planar waveguide type [5] have been studied as longitudinal connection of waveguide with microstrip line. With regard to vertical transitions, a conventional type of probe feeding has a wideband characteristic [6], [7], but it needs a metal short block with a quarter-wavelength on the substrate. The replace‐ ment of the metal short block is a patch element in the waveguide to achieve sufficient coupling between waveguide and microstrip line. The slot coupling type [8] achieves coupling between the microstrip line and the patch element in the waveguide by means of a slot, it is composed of two dielectric substrates without a metal short block. The proximity coupling type [9] has been developed more recently. It can be composed of a single dielectric substrate attached to the waveguide. A rectangular patch element on the lower plane of the dielectric substrate

© 2013 Seo; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Seo; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

couples with a microstrip line on the upper plane of the dielectric substrate. It is suitable for mass production. The proximity coupling type has been further developed for wideband [10].

waveguide is essentially *λg*/4 (*λg*: guided wavelength of the waveguide) above the substrate. Consequently, the electric current on the probe couples to the magnetic field of TE10 dominant mode of the waveguide as shown in Figure 4. Via holes are surrounding the waveguide in the structure in order to reduce the leakage of parallel plate mode transmitting into the substrate. Impedance matching could be achieved by controlling the length *ρ* of the probe and the length *Ss* of the upper waveguide. Each parameters in Figure

Back-short waveguide

Substrate with metal pattern

Upper ground

Upper waveguide

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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Lower ground

Microstrip line port Waveguide port

In order to reduce the leakage from the waveguide window at the insertion of the microstrip line, the width of the window should be narrow than the width of the cut off condition and is 0.9 mm in this case. *S*-parameters of the reflection *S*<sup>11</sup> and the transmission *S*21 are calculated by using an electromagnetic simulator based on the finite element method (Ansys HFSS) as shown in Figure 5. From the simulated results, this transition has wide frequency bandwidth.

3 are shown in Table 1 for example.

Dielectric substrate

*y z*

*x*

**Figure 2.** Probe transition with back-short

**Figure 1.** Construction of millimeter-wave automotive radar sensor and photograph for example

#### **2. Probe transition with back-short**

The transitions with short-circuited waveguide of 1/4 guided wavelength on the substrate are very popular [6], [7] because their principle of mode transformation is almost the same with that of ordinary transitions of a waveguide and a coaxial cable [11]. The probe transition connects a microstrip line and a waveguide as shown in Figure 2. A probe at one end of the microstrip line is inserted into the perpendicular waveguide whose one end is short-circuited by the back-short waveguide.

The configuration is shown in Figure 3. A dielectric substrate with conductor patterns on its both sides is placed on an open-ended waveguide (WR-12 standard waveguide). An aperture of the substrate is covered with an upper waveguide. A short circuit of the upper waveguide is essentially *λg*/4 (*λg*: guided wavelength of the waveguide) above the substrate. Consequently, the electric current on the probe couples to the magnetic field of TE10 dominant mode of the waveguide as shown in Figure 4. Via holes are surrounding the waveguide in the structure in order to reduce the leakage of parallel plate mode transmitting into the substrate. Impedance matching could be achieved by controlling the length *ρ* of the probe and the length *Ss* of the upper waveguide. Each parameters in Figure 3 are shown in Table 1 for example.

couples with a microstrip line on the upper plane of the dielectric substrate. It is suitable for mass production. The proximity coupling type has been further developed for wideband [10].

> Millimeterwave circuit

Digital signal processing unit

Transmitting antenna Receiving antenna

Waveguide

Transmitting antenna Receiving antenna

**2. Probe transition with back-short**

by the back-short waveguide.

**Figure 1.** Construction of millimeter-wave automotive radar sensor and photograph for example

The transitions with short-circuited waveguide of 1/4 guided wavelength on the substrate are very popular [6], [7] because their principle of mode transformation is almost the same with that of ordinary transitions of a waveguide and a coaxial cable [11]. The probe transition connects a microstrip line and a waveguide as shown in Figure 2. A probe at one end of the microstrip line is inserted into the perpendicular waveguide whose one end is short-circuited

The configuration is shown in Figure 3. A dielectric substrate with conductor patterns on its both sides is placed on an open-ended waveguide (WR-12 standard waveguide). An aperture of the substrate is covered with an upper waveguide. A short circuit of the upper

Waveguide-to-microstrip transition

250 Advancement in Microstrip Antennas with Recent Applications

In order to reduce the leakage from the waveguide window at the insertion of the microstrip line, the width of the window should be narrow than the width of the cut off condition and is 0.9 mm in this case. *S*-parameters of the reflection *S*<sup>11</sup> and the transmission *S*21 are calculated by using an electromagnetic simulator based on the finite element method (Ansys HFSS) as shown in Figure 5. From the simulated results, this transition has wide frequency bandwidth.

**Figure 3.** Detailed configuration of the probe transition with back-short

(a) Magnetic field distribution in *xz*-plane.

**Description Name**

Space between via holes *S* 0.5

**Table 1.** Parameters of probe transition with back-short


**3. Planar proximity coupling transition**

surrounding ground and the waveguide short electrically.

*S*| 11| [dB]

Broad wall length of

Length of back short

waveguide

**Value**

waveguide *<sup>a</sup>* 3.1 Narrow wall length of waveguide *<sup>b</sup>* 1.55 Width of microstrip line *Wm* 0.3 Length of inserted probe ρ 0.675

Thickness of substrate *T* 0.127 Diameter of via hole ϕ 0.2

66.5 71.5 76.5 81.5 86.5

Planar proximity coupling transitions shown in Figure 6 and Figure 7 have been proposed [9]. This transition can be composed of only a single dielectric substrate attached to the waveguide end and suitable for mass production. The conductor pattern with a notch (it is named a waveguide short pattern because of its function) and the microstrip line are located on the upper plane of the dielectric substrate. A rectangular patch element and a surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes are surrounding the aperture of the waveguide on the lower plane of the dielectric substrate to connect the

**Figure 5.** Reflection characteristic |*S*11| and insertion loss |*S*21| of probe transition with back-short

Frequency [GHz]

11.5 GHz 6.8 GHz

**(mm) Description Name**

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

*Ss* 0.61 Relative permittivity ε*<sup>r</sup>* 2.2

0.11 dB @ 76.5 GHz


*S*| 21| [dB] **Value (mm)** 253

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**Figure 4.** Magnetic and electric field lines


**Table 1.** Parameters of probe transition with back-short

*x*

Ground

(b) Lower pattern

TE10

Lower waveguide

Electric field

> Upper waveguide

*b*

Waveguide

*B*'

Waveguide port

Via hole

*A*

*T*

Upper waveguide

*a*

*B*

*y*

*B*' *B*

Port #1

Quasi TEM

Microstrip line

> *x y z*

*Ss*

*b*

(b) Electric field distribution in *yz*-plane.

*Ss*

(c) Sectional view in *yz*-plane

*z*

Waveguide short

*x y z*

**Figure 3.** Detailed configuration of the probe transition with back-short

field Waveguide

(a) Magnetic field distribution in *xz*-plane.

**Figure 4.** Magnetic and electric field lines

Magnetic Short

*A*'

Port #2

(a) Upper pattern

*A*

252 Advancement in Microstrip Antennas with Recent Applications

*S*

Ground

*l*

l*g* 

Probe current

er Microstrip line

f

Dielectric *A*' substrate

Microstrip line

*Wm*

Via holes

Probe

**Figure 5.** Reflection characteristic |*S*11| and insertion loss |*S*21| of probe transition with back-short

#### **3. Planar proximity coupling transition**

Planar proximity coupling transitions shown in Figure 6 and Figure 7 have been proposed [9]. This transition can be composed of only a single dielectric substrate attached to the waveguide end and suitable for mass production. The conductor pattern with a notch (it is named a waveguide short pattern because of its function) and the microstrip line are located on the upper plane of the dielectric substrate. A rectangular patch element and a surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes are surrounding the aperture of the waveguide on the lower plane of the dielectric substrate to connect the surrounding ground and the waveguide short electrically.

mode, TM01 fundamental resonant mode and TE10 fundamental transmission mode, respec‐ tively. Low transmission loss is realized by exchanging quasi TEM transmission mode and TE10 fundamental transmission mode with high efficiency utilizing TM01 fundamental resonant

*S*-parameters of the reflection *S*<sup>11</sup> and the transmission *S*<sup>21</sup> are calculated by using an electro‐ magnetic simulator based on the finite element method (Ansys HFSS) as shown in Figure 9. From the simulated results, frequency bandwidth of the planar proximity coupling transition

TEM *<sup>x</sup> <sup>y</sup>*

**Value**

Width of patch element *W* 2 Length of patch element *L* 1.1

Width of gap *G* 0.1 Relative permittivity ε*<sup>r</sup>* 2.2 Thickness of substrate *T* 0.127 Diameter of via hole ϕ 0.2

waveguide *<sup>a</sup>* 3.1 Narrow wall length of

Width of microstrip line *Wm* 0.3 Overlap length of inserted

TM01

TE10

**(mm) Description Name**

probe <sup>ρ</sup> 0.34

waveguide

**Value (mm)**

*b* 1.55

Electric field

*z*

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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255

mode. Each parameters in Figure 8 are shown in Table 2 for example.

is narrow than the ordinary probe transition with back-short.

Quasi

**Figure 8.** Electric field lines of each mode in *yz*-plane

**Description Name**

Space between via holes *S* 0.5

**Table 2.** Parameters of planar proximity coupling transition

Broad wall length of

**Figure 6.** Planar proximity coupling transition

**Figure 7.** Detailed configuration of planar proximity coupling transition

The microstrip line is inserted into the waveguide and overlaps on the rectangular patch element with overlap length *ρ* = 0.34 mm. The parameters of the transition are presented in Table 2 for example.

Figure 8 shows the electric field distribution of each mode in *yz*-plane. The modes of the microstrip line, the rectangular patch element and the waveguide are quasi TEM transmission mode, TM01 fundamental resonant mode and TE10 fundamental transmission mode, respec‐ tively. Low transmission loss is realized by exchanging quasi TEM transmission mode and TE10 fundamental transmission mode with high efficiency utilizing TM01 fundamental resonant mode. Each parameters in Figure 8 are shown in Table 2 for example.

*S*-parameters of the reflection *S*<sup>11</sup> and the transmission *S*<sup>21</sup> are calculated by using an electro‐ magnetic simulator based on the finite element method (Ansys HFSS) as shown in Figure 9. From the simulated results, frequency bandwidth of the planar proximity coupling transition is narrow than the ordinary probe transition with back-short.

**Figure 8.** Electric field lines of each mode in *yz*-plane

Substrate with metal pattern

*z*

*y*

**Figure 6.** Planar proximity coupling transition

Waveguide short

Dielectric substrate

Table 2 for example.

Microstrip line

*x*

254 Advancement in Microstrip Antennas with Recent Applications

*x y z*

**Figure 7.** Detailed configuration of planar proximity coupling transition

*A*'

*<sup>A</sup>*' (a) Upper pattern <sup>e</sup><sup>r</sup> *Wm*

*A*

*S*

f

Microstrip line Waveguide short

Via holes

Substrate

Waveguide

*B*

*W*

*<sup>L</sup> <sup>b</sup>*

*a*

*<sup>B</sup>*' (b) Lower pattern

*A*

*T*

*x*

(c) Sectional view in yz-plane

*B*' *B*

*b*

Ground

Port #1 Waveguide

*y*

Via hole

*z*

Port #2

The microstrip line is inserted into the waveguide and overlaps on the rectangular patch element with overlap length *ρ* = 0.34 mm. The parameters of the transition are presented in

Figure 8 shows the electric field distribution of each mode in *yz*-plane. The modes of the microstrip line, the rectangular patch element and the waveguide are quasi TEM transmission

*G*

Rectangular patch element

Surrounding ground

Rectangular patch element

Waveguide


**Table 2.** Parameters of planar proximity coupling transition

The quality factor QE of the patch element is given by

conductor loss, and dielectric loss.

waveguide [9] as follows:

given by

width for wideband.

1 *QE* <sup>=</sup> <sup>1</sup> *QWG* +

*QWG* <sup>=</sup> <sup>15</sup>*ωπε*0*εe<sup>L</sup> <sup>e</sup>ab* 2*Wet*

*QWG*<sup>|</sup>*We*<sup>=</sup>

2*aC π*

Broad wall length*a* of waveguide Narrow wall length*b* of waveguide

Effective relative permittivity e*<sup>e</sup>*

Effective length *Le* of patch element

Thickness of substrate *t*

**Table 3.** Relations between parameters and bandwidth

<sup>=</sup> <sup>15</sup>*ωπ* <sup>2</sup>

1 *QC* + 1 *QD*

Where, *Qwg*, *QC*, and *QD* are quality factors of the power transmitted into the waveguide,

The quality factor *QWG* is given with the cavity model and the dyadic Green's function of the

where, *ω*, *ε0* and *λ<sup>g</sup>* are angular frequency, permittivity in free space, and guided wavelength of waveguide. Relationship between the quality factor *QE* and the effective width *We* is solved

where, *C* is a constant value of 1.666. Equation (3) gives the minimum *Q* factor. *QWG* is then

The bandwidth increases with increasing *a*, while the effective width *We* is set to the optimum

The relationships between the parameters and the bandwidth are summarized in Table 3.

Effective width *We* of patch element <sup>p</sup> *<sup>W</sup> aC <sup>e</sup>* <sup>=</sup> <sup>2</sup>

Parameters

1 <sup>1</sup> - ( *<sup>λ</sup><sup>g</sup>* 2*a* )2

l<sup>e</sup> 2

*C*

1 ( sin ( *<sup>W</sup> <sup>e</sup> π* 2*a* ) ( *<sup>W</sup> <sup>e</sup><sup>π</sup>* 2*a* ) ) 2

*<sup>π</sup>* (3)

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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Bandwidth

Max.

(sin *<sup>C</sup>*)2 (4)

1 <sup>1</sup> - ( *<sup>λ</sup><sup>g</sup>* 2*a* )2

and the maximum bandwidth is obtained when *We* is expressed in (3) as follows:

*We* <sup>=</sup> <sup>2</sup>*ac*

*ε*0*εeL <sup>e</sup>b* 4*t*

(1)

257

(2)

**Figure 9.** Reflection characteristic |*S*11| and insertion loss |*S*21| of planar proximity coupling transition

#### **3.1. Bandwidth of planar proximity coupling transition**

The relationships between the parameters and the bandwidth were investigated to specify the optimum parameters for wideband [10]. Figure 10 shows an analytical model that uses a cavity model, which is used for the design of microstrip patch antennas, and the dyadic Green's function of the waveguide. *L*e and *W*<sup>e</sup> are the effective length and width of the patch element, including the fringing effect. *t* and *ε*<sup>e</sup> are the thickness and the effective relative permittivity of the dielectric substrate. The waveguide dimensions are *a* by *b*.

**Figure 10.** Analytical model using cavity model and dyadic Green's function of waveguide

The quality factor QE of the patch element is given by


256 Advancement in Microstrip Antennas with Recent Applications

*S*| 11| [dB]

66.5 71.5 76.5 81.5 86.5

Frequency [GHz]

The relationships between the parameters and the bandwidth were investigated to specify the optimum parameters for wideband [10]. Figure 10 shows an analytical model that uses a cavity model, which is used for the design of microstrip patch antennas, and the dyadic Green's function of the waveguide. *L*e and *W*<sup>e</sup> are the effective length and width of the patch element, including the fringing effect. *t* and *ε*<sup>e</sup> are the thickness and the effective relative permittivity

*We*

*t*

Substrate

Magnetic current

Waveguide

e*e*

*Le*

*a*

**Figure 10.** Analytical model using cavity model and dyadic Green's function of waveguide

**Figure 9.** Reflection characteristic |*S*11| and insertion loss |*S*21| of planar proximity coupling transition

**3.1. Bandwidth of planar proximity coupling transition**

of the dielectric substrate. The waveguide dimensions are *a* by *b*.

Rectangular patch element

*b*

*x*

*y*

*z*

5.3 GHz 2.9 GHz

0.18 dB@76.5 GGz


*S*| 21| [dB]

$$\frac{1}{\overline{Q\_E}} = \frac{1}{\overline{Q\_{WC}}} + \frac{1}{\overline{Q\_C}} + \frac{1}{\overline{Q\_D}} \tag{1}$$

Where, *Qwg*, *QC*, and *QD* are quality factors of the power transmitted into the waveguide, conductor loss, and dielectric loss.

The quality factor *QWG* is given with the cavity model and the dyadic Green's function of the waveguide [9] as follows:

$$Q\_{\rm NG} = \frac{15\omega\pi\varepsilon\_0\varepsilon\_r L\_{\rm } ab}{2W\_{\rm }t} \frac{1}{\sqrt{1 \cdot \left(\frac{\lambda\_g}{2\pi}\right)^2}} \frac{1}{\left(\frac{\sin\left(\frac{W\_{\rm \pi}}{2a}\right)}{\left(\frac{W\_{\rm \pi}}{2a}\right)}\right)^2} \tag{2}$$

where, *ω*, *ε0* and *λ<sup>g</sup>* are angular frequency, permittivity in free space, and guided wavelength of waveguide. Relationship between the quality factor *QE* and the effective width *We* is solved and the maximum bandwidth is obtained when *We* is expressed in (3) as follows:

$$\mathcal{W}\_e = \frac{2ac}{\pi} \tag{3}$$

where, *C* is a constant value of 1.666. Equation (3) gives the minimum *Q* factor. *QWG* is then given by

$$\left.Q\_{\rm WG}\right|\_{W\_{\sigma}=\frac{2\omega C}{\pi}} = \frac{15\omega\pi^2\epsilon\_0\epsilon\_r L\_r b}{4t} \frac{1}{\sqrt{1-\left(\frac{\lambda\_g}{2\omega}\right)^2}}\frac{\mathcal{C}}{(\sin\,\mathcal{C})^2} \tag{4}$$

The bandwidth increases with increasing *a*, while the effective width *We* is set to the optimum width for wideband.

The relationships between the parameters and the bandwidth are summarized in Table 3.


**Table 3.** Relations between parameters and bandwidth

### **4. Broadband microstrip-to-waveguide transition**

This section presents broadband techniques of the proximity coupling type transition. Refer to the 79 GHz UWB applications, 4 GHz bandwidth is required [12]. The proximity coupling type transition has bandwidth of 6.9 % (5.29 GHz) for the reflection coefficient below -15 dB [10]. Considering the tolerance for the manufacturing accuracy, much wider bandwidth is required. The boradband transition was presented using waveguide with large broad-wall [13]. Maximum width of the waveguide where higher order mode dose not propagate is applied and the distance from the edge of broad-wall of the waveguide to via holes on the broad-wall side of the waveguide is examined to have optimum length for wideband.

Substrate with Metal pattern

> *y z*

**Figure 11.** Configuration of broadband transition

Microstrip line

Dielectric substrate

*z*

Probe

*y*

Waveguide short

*x*

*x y*

*Wp G Wl* <sup>e</sup><sup>r</sup>

Port #2

Via holes

*b* Port #1

(c) Sectional view in yz -plane

*B*' *B*

*L*

Surrounding ground

*S*

*A*

*z*

*A*' (a) Upper pattern

> *A* '

*x*

**Figure 12.** Detailed configuration of broadband transition

f

Microstrip line Probe Waveguide short

Via holes

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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Dielectric substrate

Rectangular patch element

Surrounding ground

Waveguide with large broad-wall

*a*

*B*' (b) Lower pattern

Waveguide with large broad -wall

*T*

*A*

*W*

*Vx B*

*b*

*Vy*

Waveguide with large broad -wall

Rectangular patch element

#### **4.1. Transition structure**

Configuration of the transition is shown in Figure 11 and Figure 12. A microstrip line, a probe and a waveguide short are located on the upper plane of the dielectric substrate. A rectangular patch element and a surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes surround the aperture of the waveguide on the lower plane of the dielectric substrate to connect the surrounding ground and the waveguide short electrically. The required operation bandwidth is from 77 GHz to 81 GHz.

In terms of the bandwidth, it becomes wider as broad-wall length *a* of the waveguide increases, and narrow-wall length *b* of the waveguide decreases [10]. First, standard waveguide WR-10 can be applied for dominant mode propagation at the design frequency (79 GHz). Therefore narrow-wall length *b* of the waveguide is determined to be 1.27 mm which is the same as the narrow-wall length of WR-10 standard waveguide.

Next, the broad-wall length *a* of the waveguide is increased as large as possible to 3.1 mm where higher order mode in the waveguide dose not propagate. A rectangular patch element with the width *W* = 2.26 mm and the length *L* = 0.98 mm is located on the lower plane of the dielectric substrate at the center of the waveguide. The width *Wm* of the microstrip line is 0.3 mm corresponding to approximately 56 ohm of characteristic impedance. The probe with the width *Wp* = 0.35 mm is inserted into the waveguide and overlaps on the rectangular patch element with length *ρ* = 0.32 mm. The distance from the edge of broad-wall of the waveguide to via holes on the broad-wall side *Vy* is 0.46 mm. The distance from the edge of narrow-wall of the waveguide to via holes on the narrow- wall side *Vx* is 0.4 mm. The thickness of dielectric substrate *T* is 0.127 mm with relative permittivity *εr* is 2.2. The parameters of the transition are presented in Table 4.

#### **4.2. Design and numerical investigation**

The transition is investigated numerically by using the electromagnetic simulator based on the finite-element method (Ansys HFSS). In this calculation, loss tangent tan*δ* = 0.001 and con‐ ductivity *σ* = 5.8 × 107 S/m of a copper clad are used as loss factors. The reflection characteristic |*S*11| and the insertion loss |*S*21| of the transition with parameters in Table 4 are presented in Figure 13.

**Figure 11.** Configuration of broadband transition

**4. Broadband microstrip-to-waveguide transition**

258 Advancement in Microstrip Antennas with Recent Applications

required operation bandwidth is from 77 GHz to 81 GHz.

narrow-wall length of WR-10 standard waveguide.

**4.1. Transition structure**

presented in Table 4.

ductivity *σ* = 5.8 × 107

Figure 13.

**4.2. Design and numerical investigation**

This section presents broadband techniques of the proximity coupling type transition. Refer to the 79 GHz UWB applications, 4 GHz bandwidth is required [12]. The proximity coupling type transition has bandwidth of 6.9 % (5.29 GHz) for the reflection coefficient below -15 dB [10]. Considering the tolerance for the manufacturing accuracy, much wider bandwidth is required. The boradband transition was presented using waveguide with large broad-wall [13]. Maximum width of the waveguide where higher order mode dose not propagate is applied and the distance from the edge of broad-wall of the waveguide to via holes on the broad-wall side of the waveguide is examined to have optimum length for wideband.

Configuration of the transition is shown in Figure 11 and Figure 12. A microstrip line, a probe and a waveguide short are located on the upper plane of the dielectric substrate. A rectangular patch element and a surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes surround the aperture of the waveguide on the lower plane of the dielectric substrate to connect the surrounding ground and the waveguide short electrically. The

In terms of the bandwidth, it becomes wider as broad-wall length *a* of the waveguide increases, and narrow-wall length *b* of the waveguide decreases [10]. First, standard waveguide WR-10 can be applied for dominant mode propagation at the design frequency (79 GHz). Therefore narrow-wall length *b* of the waveguide is determined to be 1.27 mm which is the same as the

Next, the broad-wall length *a* of the waveguide is increased as large as possible to 3.1 mm where higher order mode in the waveguide dose not propagate. A rectangular patch element with the width *W* = 2.26 mm and the length *L* = 0.98 mm is located on the lower plane of the dielectric substrate at the center of the waveguide. The width *Wm* of the microstrip line is 0.3 mm corresponding to approximately 56 ohm of characteristic impedance. The probe with the width *Wp* = 0.35 mm is inserted into the waveguide and overlaps on the rectangular patch element with length *ρ* = 0.32 mm. The distance from the edge of broad-wall of the waveguide to via holes on the broad-wall side *Vy* is 0.46 mm. The distance from the edge of narrow-wall of the waveguide to via holes on the narrow- wall side *Vx* is 0.4 mm. The thickness of dielectric substrate *T* is 0.127 mm with relative permittivity *εr* is 2.2. The parameters of the transition are

The transition is investigated numerically by using the electromagnetic simulator based on the finite-element method (Ansys HFSS). In this calculation, loss tangent tan*δ* = 0.001 and con‐


S/m of a copper clad are used as loss factors. The reflection characteristic

**Figure 12.** Detailed configuration of broadband transition


Figure 14 shows the calculated electric field distributions in the *xy*-plane including *BB'*-line at 76.4 GHz, 79 GHz and 85.5 GHz. It is observed that fundamental mode of TM01 is excited at 76.4 GHz and 79 GHz in Figure 14 (a) and (b). On the other hand, a higher order mode is

> (b) Electric field intensity at 79 GHz

The length *L* of the rectangular patch element affects to the lower resonant frequency as shown in Figure 15. The lower resonant frequency can be controlled by the length *L* of the rectangular

*L* = 0.99 mm

64 69 74 79 84 89 94 Frequency [GHz]

*L* = 0.98 mm

*L* = 0.98 mm *L* = 0.99 mm

*L* = 0.97 mm

[V/m] 1.0e+5

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1.0e+4

Intensity

1.5e-2

(c) Electric field intensity at 85.5 GHz

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

observed at 85.5 GHz as shown in Figure 14 (c).

(a) Electric field intensity at 76.4 GHz

**Figure 14.** Electric field intensity distributions in *xy*-plane.

*L* = 0.97 mm

**Figure 15.** |*S*11| vs. length *L* of the patch element (Lower resonant frequency control)

*x*





B]




0

*4.2.1. Lower operation frequency by L*

*y*

*z*

patch element.

#### **Table 4.** Parameters of the broadband transition

It can be seen from the simulation results that the bandwidth for the reflection coefficient |*S*11| below -15 dB is 14.4 GHz, and the insertion loss |*S*21| is -0.28 dB from 77 GHz to 81 GHz. In this case, two different resonances are observed. Lower resonant frequency is 76.4 GHz and higher resonant frequency is 85.5 GHz.

**Figure 13.** Reflection characteristic |*S*11| and insertion loss |*S*21| of broadband transition

Figure 14 shows the calculated electric field distributions in the *xy*-plane including *BB'*-line at 76.4 GHz, 79 GHz and 85.5 GHz. It is observed that fundamental mode of TM01 is excited at 76.4 GHz and 79 GHz in Figure 14 (a) and (b). On the other hand, a higher order mode is observed at 85.5 GHz as shown in Figure 14 (c).

**Figure 14.** Electric field intensity distributions in *xy*-plane.

#### *4.2.1. Lower operation frequency by L*

**Description Name**

260 Advancement in Microstrip Antennas with Recent Applications

Overlap length of inserted

Distance from broad wall to via








0

B]

**Table 4.** Parameters of the broadband transition

higher resonant frequency is 85.5 GHz.

hole

**Value**

Broad wall length of waveguide *<sup>a</sup>* 3.1 Narrow wall length of

**(mm) Description Name**

Distance from narrow wall to via

waveguide

Width of patch element *W* 2.26 Length of patch element *L* 0.98

Width of microstrip line *Wm* 0.3 Width of probe *Wp* 0.35

probe <sup>ρ</sup> 0.32 Width of gap *<sup>G</sup>* 0.1

Thickness of substrate *T* 0.127 Relative permittivity ε*<sup>r</sup>* 2.2

Space between via holes *S* 0.4 Diameter of via hole ϕ 0.2

hole

It can be seen from the simulation results that the bandwidth for the reflection coefficient |*S*11| below -15 dB is 14.4 GHz, and the insertion loss |*S*21| is -0.28 dB from 77 GHz to 81 GHz. In this case, two different resonances are observed. Lower resonant frequency is 76.4 GHz and

Frequency [GHz]

**Figure 13.** Reflection characteristic |*S*11| and insertion loss |*S*21| of broadband transition

64 69 74 79 84 89 94

14.4 GHz

*Vy* 0.46

**Value (mm)**

*b* 1.27

*Vx* 0.4



B]

The length *L* of the rectangular patch element affects to the lower resonant frequency as shown in Figure 15. The lower resonant frequency can be controlled by the length *L* of the rectangular patch element.

**Figure 15.** |*S*11| vs. length *L* of the patch element (Lower resonant frequency control)

#### *4.2.2. Higher operation frequency by Vy*

The distance *Vy* from the edge of the broad-wall of the waveguide to via holes affects to the higher resonant frequency as shown in Figure 16. The higher resonant frequency can be controlled by the distance *Vy* from the edge of the broad-wall of the waveguide to via holes.

**Figure 17.** Impedance vs. overlap length ρ of the inserted probe

**Figure 18.** Impedance vs. width *Wp* of the probe

probe and the width *Wp* of the prove.

So, the impedance matching can be controlled by optimizing of the overlap length *ρ* of inserted

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263

**Figure 16.** |*S*11| vs. distance *Vy* from the edge of the broad-wall of the waveguide to via holes (Higher resonant fre‐ quency control)|

#### *4.2.3. Impedance matching by ρ and Wp*

The overlap length *ρ* of the inserted probe is most effective for the impedance matching to the waveguide as shown in Figure 17. Increment of the overlap length *ρ* of the inserted probe with rectangular patch element causes increase of inductance.

The width *W<sup>p</sup>* of the prove is also effective for the impedance matching to the waveguide as shown in Figure 18. Increment of width *W<sup>p</sup>* of the prove causes decrease of resistance. These Smith charts are observed from the waveguide *port#1* in Figure 12 (c) at a distance of 2.0 mm under the surrounding ground on the lower plane of the substrate.

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band http://dx.doi.org/10.5772/54662 263

**Figure 17.** Impedance vs. overlap length ρ of the inserted probe

*4.2.2. Higher operation frequency by Vy*

262 Advancement in Microstrip Antennas with Recent Applications


*4.2.3. Impedance matching by ρ and Wp*

rectangular patch element causes increase of inductance.

under the surrounding ground on the lower plane of the substrate.



*S*| 11| [dB]

quency control)|




0

The distance *Vy* from the edge of the broad-wall of the waveguide to via holes affects to the higher resonant frequency as shown in Figure 16. The higher resonant frequency can be controlled by the distance *Vy* from the edge of the broad-wall of the waveguide to via holes.

> 64 69 74 79 84 89 94 Frequency [GHz]

**Figure 16.** |*S*11| vs. distance *Vy* from the edge of the broad-wall of the waveguide to via holes (Higher resonant fre‐

The overlap length *ρ* of the inserted probe is most effective for the impedance matching to the waveguide as shown in Figure 17. Increment of the overlap length *ρ* of the inserted probe with

The width *W<sup>p</sup>* of the prove is also effective for the impedance matching to the waveguide as shown in Figure 18. Increment of width *W<sup>p</sup>* of the prove causes decrease of resistance. These Smith charts are observed from the waveguide *port#1* in Figure 12 (c) at a distance of 2.0 mm

*V***<sup>y</sup>** = 0.46 mm

*V***<sup>y</sup>** = 0.47 mm

*V***<sup>y</sup>** = 0.48 mm

*V***<sup>y</sup>** = 0.45 mm *V***<sup>y</sup>** = 0.46 mm *V***<sup>y</sup>** = 0.47 mm *V***<sup>y</sup>** = 0.48 mm

*V***<sup>y</sup>** = 0.45 mm

**Figure 18.** Impedance vs. width *Wp* of the probe

So, the impedance matching can be controlled by optimizing of the overlap length *ρ* of inserted probe and the width *Wp* of the prove.

#### *4.2.4. Wideband impedance matching by ρ and W<sup>p</sup>*

For the wideband impedance matching, both of the length *ρ* of the inserted probe and the width *Wp* of the probe are optimized as shown in Figure 19. It can be seen from the simulation results that both of *ρ* and *Wp* affect the wideband impedance matching. In these design, other parameters except *ρ* and *Wp* are same as in Table 4.

> to 81 GHz. A time gate function was used to exclude undesired waves, and high accuracy was achieved in this measurement. The distance between the center of the waveguides was set at 50 mm, which was long enough to distinguish between desired and undesired waves in the

Waveguide short Via holes with

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

controlled position

Rectangular patch element Aperture of waveguide with large broad-wall

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265

(a) Upper Plane of Substrate (b) Lower Plane of Substrate

Probe

Dielectric substrate

Microstrip line

Figure 23 shows the comparison of three designed transitions. Refer to the bandwidth, measured results are approximately 1.8 GHz decreased compared with the simulation results. For the insertion loss, the measured results are approximately 0.38 dB increased compared

In these results, design of increased bandwidth causes increase of insertion loss. Therefore, the bandwidth and the insertion loss is in tradeoff relation. So, the transition required each

application can be designed by optimizing of each parameters.

time domain.

**Figure 20.** Fabricated transitions

**Figure 21.** DUT in measurement

with the simulation results.

*4.3.2. Comparison of measured performance*

50 mm

**Figure 19.** Comparison of three type transitions

#### **4.3. Experiment**

Three transitions for the results shown in Figure 19 are fabricated. The photograph of the fabricated transitions are in Figure 20. Figure 20 (a) shows the upper plane of the substrate and is common for each design except the width *Wp* of the probe and the overlap length *ρ* of the inserted probe as each design. Figure 20 (b) shows the lower plane of the substrate and is common at each design.

#### *4.3.1. Measured banwidth*

Measured the reflection coefficient are shown in Figure 21. Maximum bandwidth for reflection coefficients below -15 dB is 15.1 GHz when *Wp* is 0.45 mm and *ρ* is 0.29 mm. In this measure‐ ment, the device-under-test (DUT) was composed of a pair of transitions with one microstrip line between them as shown in Figure 21. The measured |*S*11| and |*S*21| in Figure 22 were given by taking the transmission coefficient of the DUT, subtracting the loss of the microstrip line, and dividing by two. The loss of the microstrip line was measured as 0.05 dB/mm from 77 GHz

**Figure 20.** Fabricated transitions

*4.2.4. Wideband impedance matching by ρ and W<sup>p</sup>*

264 Advancement in Microstrip Antennas with Recent Applications

parameters except *ρ* and *Wp* are same as in Table 4.


**Figure 19.** Comparison of three type transitions

**4.3. Experiment**

common at each design.

*4.3.1. Measured banwidth*

*Wp* = 0.35 mm,

*Wp* = 0.45 mm,




B]




0

For the wideband impedance matching, both of the length *ρ* of the inserted probe and the width *Wp* of the probe are optimized as shown in Figure 19. It can be seen from the simulation results that both of *ρ* and *Wp* affect the wideband impedance matching. In these design, other

64 69 74 79 84 89 94

= 0.33 mm *Wp* = 0.4 mm,

= 0.3 mm

Frequency [GHz]

Three transitions for the results shown in Figure 19 are fabricated. The photograph of the fabricated transitions are in Figure 20. Figure 20 (a) shows the upper plane of the substrate and is common for each design except the width *Wp* of the probe and the overlap length *ρ* of the inserted probe as each design. Figure 20 (b) shows the lower plane of the substrate and is

Measured the reflection coefficient are shown in Figure 21. Maximum bandwidth for reflection coefficients below -15 dB is 15.1 GHz when *Wp* is 0.45 mm and *ρ* is 0.29 mm. In this measure‐ ment, the device-under-test (DUT) was composed of a pair of transitions with one microstrip line between them as shown in Figure 21. The measured |*S*11| and |*S*21| in Figure 22 were given by taking the transmission coefficient of the DUT, subtracting the loss of the microstrip line, and dividing by two. The loss of the microstrip line was measured as 0.05 dB/mm from 77 GHz

= 0.29 mm

16.7 GHz

15.5 GHz

14.4 GHz

to 81 GHz. A time gate function was used to exclude undesired waves, and high accuracy was achieved in this measurement. The distance between the center of the waveguides was set at 50 mm, which was long enough to distinguish between desired and undesired waves in the time domain.

#### *4.3.2. Comparison of measured performance*

Figure 23 shows the comparison of three designed transitions. Refer to the bandwidth, measured results are approximately 1.8 GHz decreased compared with the simulation results. For the insertion loss, the measured results are approximately 0.38 dB increased compared with the simulation results.

In these results, design of increased bandwidth causes increase of insertion loss. Therefore, the bandwidth and the insertion loss is in tradeoff relation. So, the transition required each application can be designed by optimizing of each parameters.

Three types of design are presented. It is confirmed by experiments that the most wideband transition exhibits a bandwidth of 19.1 % (15.1 GHz) for the reflection coefficient below -15 dB

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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267

Narrow-wall-connected microstrip-to-waveguide transition using V-shaped patch element in millimeter-wave band was proposed [14]. Since the microstrip line on the narrow-wall is perpendicular to the *E*-plane of the waveguide, the waveguide field does not couple directly to the microstrip line. The current on the V-shaped patch element flows along the inclined edges, then current on the V-shaped patch element couples to the microstrip line efficiently. Three types of the transitions are investigated. *S*-parameters of the reflection *S*11 and the transmission *S*21 are calculated by using an electromagnetic simulator based on the finite element method (Ansys HFSS). The numerical investigations of these transitions show some relations between the bandwidth and the insertion loss. It is confirmed that the improved transition exhibits an insertion loss of 0.6 dB from 76 to 77 GHz, and a bandwidth of 4.1 % (3.15

In some applications, narrow-wall-connected micro-strip-to-waveguide transition is required. Refer to the former developed proximity coupling type transition [9],[10], the microstrip line is located on the waveguide broad-wall and the microstrip line probe is parallel to *E*-plane of the waveguide, therefore, current on the rectangular patch element couples to the microstrip line efficiently. However, on the occasion of the microstrip line on the narrow-wall of the waveguide, the microstrip line probe is orthogonal to *E*-plane of the waveguide. Therefore, they do not couple essentially. To couple currents on the microstrip line and the patch element,

Configuration of the transition is shown in Figure 24 and Figure 25. The microstrip line and the waveguide short are located on the upper plane of the dielectric substrate. The V-shaped patch element and the surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes are surrounding the aperture of the waveguide on the lower plane of the substrate to connect the surrounding ground and the waveguide short electrically. The design frequency range is from 76 GHz to 77 GHz. The V-shaped patch element is designed as follows. Refer to the conventional proximity coupling type [9],[10], the current on rectangular patch element has only *y*-component which is parallel to *E*-plane of the waveguide shown in Figure 26 (a). In this case, the current on the rectangular patch element is parallel to the microstrip line, therefore, the current on the rectangular patch element couples to the microstrip line. On

a V-shaped patch element is applied instead of the rectangular patch element.

**5. Narrow-wall-connected microstrip-to-waveguide transition**

and insertion loss of -0.71 dB from 77 GHz to 81 GHz.

GHz) for the reflection coefficient below -15 dB.

**5.2. Transition structure and design**

*5.2.1. Transition structure*

**5.1. Background**

**Figure 22.** Measured bandwidth of three type transitions

**Figure 23.** Comparison of measured performance

#### **4.4. Conclusion**

Broadband microstrip-to-waveguide transition using waveguide with large broad-wall were developed in millimeter-wave band. By applying large broad-wall, the bandwidth is extended. Moreover, the distance from the edge of the broad-wall of the waveguide to via holes are examined to create double resonances, consequently the bandwidth is extended.

Three types of design are presented. It is confirmed by experiments that the most wideband transition exhibits a bandwidth of 19.1 % (15.1 GHz) for the reflection coefficient below -15 dB and insertion loss of -0.71 dB from 77 GHz to 81 GHz.

#### **5. Narrow-wall-connected microstrip-to-waveguide transition**

Narrow-wall-connected microstrip-to-waveguide transition using V-shaped patch element in millimeter-wave band was proposed [14]. Since the microstrip line on the narrow-wall is perpendicular to the *E*-plane of the waveguide, the waveguide field does not couple directly to the microstrip line. The current on the V-shaped patch element flows along the inclined edges, then current on the V-shaped patch element couples to the microstrip line efficiently. Three types of the transitions are investigated. *S*-parameters of the reflection *S*11 and the transmission *S*21 are calculated by using an electromagnetic simulator based on the finite element method (Ansys HFSS). The numerical investigations of these transitions show some relations between the bandwidth and the insertion loss. It is confirmed that the improved transition exhibits an insertion loss of 0.6 dB from 76 to 77 GHz, and a bandwidth of 4.1 % (3.15 GHz) for the reflection coefficient below -15 dB.

#### **5.1. Background**

In some applications, narrow-wall-connected micro-strip-to-waveguide transition is required. Refer to the former developed proximity coupling type transition [9],[10], the microstrip line is located on the waveguide broad-wall and the microstrip line probe is parallel to *E*-plane of the waveguide, therefore, current on the rectangular patch element couples to the microstrip line efficiently. However, on the occasion of the microstrip line on the narrow-wall of the waveguide, the microstrip line probe is orthogonal to *E*-plane of the waveguide. Therefore, they do not couple essentially. To couple currents on the microstrip line and the patch element, a V-shaped patch element is applied instead of the rectangular patch element.

#### **5.2. Transition structure and design**

#### *5.2.1. Transition structure*

**4.4. Conclusion**

**Bandwidth [GHz]**

1





B]




0

266 Advancement in Microstrip Antennas with Recent Applications

**Figure 23.** Comparison of measured performance

**0.35mm**

Measured Simulation

**Figure 22.** Measured bandwidth of three type transitions

*Wp* = 0.35 mm,

*Wp* = 0.45 mm,

= 0.29 mm

**Width (** *Wp* **) of probe**

**0.4 mm 0.45 mm**

Measured Simulation

= 0.3 mm

**0.35 mm 0.4 mm 0.45 mm Width (** *Wp* **) of probe**

**Loss [dB]**

64 69 74 79 84 89 94

15.1 GHz

13.9 GHz

11.6 GHz

Frequency [GHz]

= 0.33 mm *Wp* = 0.4 mm,

Broadband microstrip-to-waveguide transition using waveguide with large broad-wall were developed in millimeter-wave band. By applying large broad-wall, the bandwidth is extended. Moreover, the distance from the edge of the broad-wall of the waveguide to via holes are

examined to create double resonances, consequently the bandwidth is extended.

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

1

Configuration of the transition is shown in Figure 24 and Figure 25. The microstrip line and the waveguide short are located on the upper plane of the dielectric substrate. The V-shaped patch element and the surrounding ground are patterned on the lower plane of the dielectric substrate. Via holes are surrounding the aperture of the waveguide on the lower plane of the substrate to connect the surrounding ground and the waveguide short electrically. The design frequency range is from 76 GHz to 77 GHz. The V-shaped patch element is designed as follows. Refer to the conventional proximity coupling type [9],[10], the current on rectangular patch element has only *y*-component which is parallel to *E*-plane of the waveguide shown in Figure 26 (a). In this case, the current on the rectangular patch element is parallel to the microstrip line, therefore, the current on the rectangular patch element couples to the microstrip line. On the other hand, current on the patch element which is excited by electromagnetic field in the waveguide must have both of *x*-component and *y*-component in order to couple to the microstrip line on the waveguide narrow-wall. The current on the V-shaped patch element is divided to two directions along the side edge as shown in Figure 26 (b). Consequently, the current on the V-shaped patch element creates parallel component with the microstrip line, and effective coupling is achieved with the microstrip line.

y

*5.2.2. Transition design*

z x

Ground

**Figure 26.** Current distributions on patch element

**Description Name**

Overlap length of inserted

Shift length of microstrip line from center of waveguide

**Table 5.** Parameters of transition

Current

(a) Current on rectangle patch

Microstrip line

First, the rectangular patch element with the width *W* = 2.6 mm and the length *L* = 1.02 mm is located on the lower plane of the dielectric substrate at the center of the waveguide. Then, both sides of the rectangular patch element are cut by the patch-cut-angle *θ* = 30 degrees and also middle of upper horizontal edge of the rectangular patch element is cut as shown in Figure 25 (b). The microstrip line is located on the waveguide narrow-wall as shown in Figure 25 (a) with the shift length *Y* = 0.34 mm from the center of the waveguide. The microstrip line is located just above the side edge of the V-shaped patch element as shown in Figure 25 (b). The microstrip line is inserted into the waveguide and overlaps over the V-shaped patch element with the length *ρ* = 0.23 mm. The parameters of the transition are presented in Table 5.

Mode conversion from the waveguide to the microstrip line is achieved by the resonance of the V-shaped patch element. The dominant TE10 mode of the waveguide is converted to the quasi-TEM mode of the microstrip line. Figure 27 shows the calculated electric field intensity distribu‐ tioninthe*xy*-planeincluding*BB'*-line.Theelectricfieldintensity*E*includes*x,y*and*z*components of the electric field. The V-shaped patch element is resonated in two directions, by the reso‐

**(mm) Description Name**

**Value (mm)**

nance of current distribution along the both side edges of the V-shaped patch element.

Width of patch element *W* 2.6 Width of gap *G* 0.1 Length of patch element *L* 1.02 Thickness of substrate *T* 0.127 Patch cut angle θ 30 deg. Relative permittivity ε*<sup>r</sup>* 2.2

microstrip line <sup>ρ</sup> 0.23 Broad wall length of waveguide *<sup>a</sup>* 3.1 Width of cut patch element *Wc* 0.46 Narrow wall length of waveguide *b* 1.55 Length of cut patch element *Lc* 0.1 Diameter of via hole ϕ 0.2 Width of microstrip line *Wm* 0.3 Space between via holes *S* 0.5

**Value**

*Y* 0.34

Ground

Current Microstrip line

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269

(b) Current on V-shaped patch

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

**Figure 24.** Configuration of transiton

**Figure 25.** Detailed configurations of transition

**Figure 26.** Current distributions on patch element

#### *5.2.2. Transition design*

the other hand, current on the patch element which is excited by electromagnetic field in the waveguide must have both of *x*-component and *y*-component in order to couple to the microstrip line on the waveguide narrow-wall. The current on the V-shaped patch element is divided to two directions along the side edge as shown in Figure 26 (b). Consequently, the current on the V-shaped patch element creates parallel component with the microstrip line,

Waveguide short Microstrip line

y z x

*Y*

*Wm*

e*r*

*S*

Via holes

(a) Upper pattern

*A B*

y x

z

*G*

Dielectric substrate

Via hole

Waveguide

*W*

*Wc Lc*

*a*

*A*' *B*'

q

*L*

*a* Waveguide

*T*

Ground

*b*

*B*

Port #1

(c) Cross section view

Port #2 *A*'

Microstrip line

V-shaped patch element

Microstrip

*B*'

V-shaped patch element

(b) Lower pattern line

q

and effective coupling is achieved with the microstrip line.

268 Advancement in Microstrip Antennas with Recent Applications

Substrate with metal pattern

> *y z*

**Figure 24.** Configuration of transiton

Waveguide short

*A*

Dielectric substrate

**Figure 25.** Detailed configurations of transition

*x*

f

First, the rectangular patch element with the width *W* = 2.6 mm and the length *L* = 1.02 mm is located on the lower plane of the dielectric substrate at the center of the waveguide. Then, both sides of the rectangular patch element are cut by the patch-cut-angle *θ* = 30 degrees and also middle of upper horizontal edge of the rectangular patch element is cut as shown in Figure 25 (b). The microstrip line is located on the waveguide narrow-wall as shown in Figure 25 (a) with the shift length *Y* = 0.34 mm from the center of the waveguide. The microstrip line is located just above the side edge of the V-shaped patch element as shown in Figure 25 (b). The microstrip line is inserted into the waveguide and overlaps over the V-shaped patch element with the length *ρ* = 0.23 mm. The parameters of the transition are presented in Table 5.

Mode conversion from the waveguide to the microstrip line is achieved by the resonance of the V-shaped patch element. The dominant TE10 mode of the waveguide is converted to the quasi-TEM mode of the microstrip line. Figure 27 shows the calculated electric field intensity distribu‐ tioninthe*xy*-planeincluding*BB'*-line.Theelectricfieldintensity*E*includes*x,y*and*z*components of the electric field. The V-shaped patch element is resonated in two directions, by the reso‐ nance of current distribution along the both side edges of the V-shaped patch element.


**Table 5.** Parameters of transition

**Figure 27.** Electric field intensity distribution in *xy*-plane

#### **5.3. Numerical investigation**

#### *5.3.1. Operating frequency by L*

The reflection characteristic of the V-shaped patch element with the length *L* = 1.02 mm is presented in Figure 28. It can be seen from the simulation results that the bandwidth for |*S*11| below -15 dB is 2 GHz, and the insertion loss |*S*21| is 0.32 dB over the frequency range from 76 GHz to 77 GHz. The length *L* of the V-shaped patch element affects to the resonant frequency as shown in Figure 28. Increment of the length *L* of the V-shaped patch element causes the lower resonant frequency. So, operating frequency of this transition can be controlled by the length *L* of the V-shaped patch element.

#### *5.3.2. Impedance matching by ρ*

The overlap length *ρ* of the inserted microstrip line affects to the impedance as shown in Figure 29. Increment of the overlap length *ρ* causes increases of capacitive reactance at the desired frequency, and decrement of the overlap length *ρ* causes increases of inductive reactance. This Smith chart is observed from the waveguide *port#1* in Figure 25 (c) at a distance of 1.5 mm under the surrounding ground on the lower plane of the substrate. So, impedance matching can be controlled by adjusting the overlap length *ρ* to cancel reactive component.

**Figure 29.** Relation between Impedance and Length of Inserted Microstrip Line ρ from 66.5 GHz to 76.5 GHz

**Figure 28.** |*S*11| vs. length of V-shaped patch element *L* and transition characteristic |*S*21|

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271

**Figure 28.** |*S*11| vs. length of V-shaped patch element *L* and transition characteristic |*S*21|

Inte

nsit

[V/m] 2.0e+6

1.8e+2

1.5e-2

y

y

z x

**5.3. Numerical investigation**

*5.3.1. Operating frequency by L*

*5.3.2. Impedance matching by ρ*

**Figure 27.** Electric field intensity distribution in *xy*-plane

270 Advancement in Microstrip Antennas with Recent Applications

length *L* of the V-shaped patch element.

The reflection characteristic of the V-shaped patch element with the length *L* = 1.02 mm is presented in Figure 28. It can be seen from the simulation results that the bandwidth for |*S*11| below -15 dB is 2 GHz, and the insertion loss |*S*21| is 0.32 dB over the frequency range from 76 GHz to 77 GHz. The length *L* of the V-shaped patch element affects to the resonant frequency as shown in Figure 28. Increment of the length *L* of the V-shaped patch element causes the lower resonant frequency. So, operating frequency of this transition can be controlled by the

The overlap length *ρ* of the inserted microstrip line affects to the impedance as shown in Figure 29. Increment of the overlap length *ρ* causes increases of capacitive reactance at the desired frequency, and decrement of the overlap length *ρ* causes increases of inductive reactance. This Smith chart is observed from the waveguide *port#1* in Figure 25 (c) at a distance of 1.5 mm under the surrounding ground on the lower plane of the substrate. So, impedance matching

can be controlled by adjusting the overlap length *ρ* to cancel reactive component.

**Figure 29.** Relation between Impedance and Length of Inserted Microstrip Line ρ from 66.5 GHz to 76.5 GHz

#### *5.3.3. Bandwidth by θ*

The patch cut angle *θ* affects to the bandwidth for the reflection coefficient below -15 dB as shown in Figure 30. The transition characteristic change is investigated by change of the patch cut angle *θ* from 5 degrees to 50 degrees. Some parameters of *W*, *L*, *ρ*, *Wc*, *Lc* and *Y* are optimized at each patch cut angle *θ*. Least insertion loss |*S*21| is obtained at 30 degrees of the patch cut angle *θ*. In this case, the bandwidth for |*S*11| below -15 dB is 2 GHz and the insertion loss |*S*21| is -0.32 dB.

Waveguide

*z*

*x y*

**5.4. Design variety of transition**

**Table 6.** Parameters of wideband design

*5.4.3. Wideband and low loss design*

*5.4.1. Low loss design*

*5.4.2. Wideband design*

V-shaped patch element

**Figure 31.** Magnetic field distribution in *yz*-plane at *x* = 2.48 mm


Microstrip line

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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273

As shown in Figure 30, least insertion loss is obtained at the patch cut angle *θ* = 30 degrees. Configuration is shown in Figure 25 and design parameters are just the same as shown in Table 5. The bandwidth for |*S*11| below -15 dB is 2 GHz and the insertion loss |*S*21| is -0.32 dB.

In this design, the patch cut angle *θ* of 10 degrees is applied. Configuration is shown in Figure 30 and some parameters must be changed as Table 6, but other parameters are the same as Table 5. Characteristic of this transition is shown in Figure 4.13. The bandwidth for |*S*11| below

**Description Name**

The transition with the wideband design described before is modified. At the *y*-directional position *Cy* = 0.32 mm from the top of the V-shaped patch element, the V-shape patch element is cut to *x*-direction with the length *Cx* of 0.1mm as shown in Figure 32. The basic configuration

Length of patch element *L* 1.11 Patch cut angle θ 10 deg. Overlap length of inserted microstrip line ρ 0.28 Width of cut patch element *Wc* 0.5 Shift length of microstrip line from center of waveguide *Y* 0.385

**Value (mm)**

On the occasion of the patch cut angle *θ* = 10 degrees, the bandwidth is extended to 3.5 GHz, but the insertion loss |*S*21| is increased to -0.41 dB compared with the patch cut angle *θ* = 30 deg.

**Figure 30.** Bandwidth and transition characteristic |*S*21| related by patch cut angle θ

In the design with the small patch cut angle *θ*, current on the V-shaped patch element has small *x*-component, therefore the loss increases. The cause is that the magnetic field is small. This magnetic field is excited by the current of *x*-component on the V-shaped patch element and the magnetic field surrounds the microstrip line. Small magnetic field causes weak coupling between the V-shaped patch element and the microstrip line. Figure 4.31 shows calculated magnetic field distribution in the *yz*-plane at the *x* position of 2.48 mm. The magnetic field intensity *H* include *x,y* and *z* components of the magnetic field. On the other hand, in the design with large patch cut angle *θ*, the area of the V-shaped patch element decreases and the quality factor *Q* of the patch element increases, then bandwidth decreases.

**Figure 31.** Magnetic field distribution in *yz*-plane at *x* = 2.48 mm

#### **5.4. Design variety of transition**

#### *5.4.1. Low loss design*

*5.3.3. Bandwidth by θ*

272 Advancement in Microstrip Antennas with Recent Applications

is -0.32 dB.

*θ* = 30 deg.

0

B

an

d

width [G

H

z]

1

2

3

4

The patch cut angle *θ* affects to the bandwidth for the reflection coefficient below -15 dB as shown in Figure 30. The transition characteristic change is investigated by change of the patch cut angle *θ* from 5 degrees to 50 degrees. Some parameters of *W*, *L*, *ρ*, *Wc*, *Lc* and *Y* are optimized at each patch cut angle *θ*. Least insertion loss |*S*21| is obtained at 30 degrees of the patch cut angle *θ*. In this case, the bandwidth for |*S*11| below -15 dB is 2 GHz and the insertion loss |*S*21|

On the occasion of the patch cut angle *θ* = 10 degrees, the bandwidth is extended to 3.5 GHz, but the insertion loss |*S*21| is increased to -0.41 dB compared with the patch cut angle

0 10 20 30 40 50 60

Bandwidth Loss |*S*21|

Frequency [GHz]

In the design with the small patch cut angle *θ*, current on the V-shaped patch element has small *x*-component, therefore the loss increases. The cause is that the magnetic field is small. This magnetic field is excited by the current of *x*-component on the V-shaped patch element and the magnetic field surrounds the microstrip line. Small magnetic field causes weak coupling between the V-shaped patch element and the microstrip line. Figure 4.31 shows calculated magnetic field distribution in the *yz*-plane at the *x* position of 2.48 mm. The magnetic field intensity *H* include *x,y* and *z* components of the magnetic field. On the other hand, in the design with large patch cut angle *θ*, the area of the V-shaped patch element decreases and the quality

**Figure 30.** Bandwidth and transition characteristic |*S*21| related by patch cut angle θ

factor *Q* of the patch element increases, then bandwidth decreases.



L

oss *S*| 21| [d

B]




As shown in Figure 30, least insertion loss is obtained at the patch cut angle *θ* = 30 degrees. Configuration is shown in Figure 25 and design parameters are just the same as shown in Table 5. The bandwidth for |*S*11| below -15 dB is 2 GHz and the insertion loss |*S*21| is -0.32 dB.

#### *5.4.2. Wideband design*

In this design, the patch cut angle *θ* of 10 degrees is applied. Configuration is shown in Figure 30 and some parameters must be changed as Table 6, but other parameters are the same as Table 5. Characteristic of this transition is shown in Figure 4.13. The bandwidth for |*S*11| below -15 dB is 3.5 GHz and the insertion loss |*S*21| is -0.41 dB.


**Table 6.** Parameters of wideband design

#### *5.4.3. Wideband and low loss design*

The transition with the wideband design described before is modified. At the *y*-directional position *Cy* = 0.32 mm from the top of the V-shaped patch element, the V-shape patch element is cut to *x*-direction with the length *Cx* of 0.1mm as shown in Figure 32. The basic configuration is as shown in Figure 25 but the V-shaped patch element is modified as shown in Figure 32. Some parameters are optimized and changed as Table 7, although other parameters are the same as Table 5.

**5.5. Measured performance of three transitons**

of 4.1 % (3.15 GHz) for the reflection coefficient below -15 dB.

simulation results.

**Figure 33.** Fabricated transitons

The photograph of the fabricated transitions are shown in Figure 33. Figure 33 (a) shows the upper plane of the substrate and is common for each design except *y*-position of the microstrip line(*Y*). As described before, the shift length *Y* of the microstrip line from the center of the waveguide is changed at each design. Figure 34 shows the comparison of three designed transitions. Refer to the bandwidth, measured results agree with the simulation results. For the insertion loss, the measured results are approximately 0.3 dB increased compared with the

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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275

Three types of design are presented and as a compatible design of low loss and wideband, a new modified V-shape patch element is proposed. It is confirmed by experiments that the improved transition exhibits an insertion loss of 0.6 dB from 76 to 77 GHz, and a bandwidth

To get the wideband of the transition, the patch cut angle *θ* is kept to 10 degrees. To achieve strong coupling, this modification of V-shaped patch element is effective. Due to this structural modification, horizontal component of electric current on the patch element increases. Consequently strong coupling to the microstrip line is achieved. The bandwidth for |*S*11| below -15 dB is 3.1 GHz and the insertion loss |*S*21| is 0.34 dB. The bandwidth is 0.4 GHz narrow than the wideband design and the loss is approximately equal as the low loss design.

**Figure 32.** Lower pattern of the transition with modified V-shaped patch element


**Table 7.** Parameters of wideband and low loss design

#### **5.5. Measured performance of three transitons**

is as shown in Figure 25 but the V-shaped patch element is modified as shown in Figure 32. Some parameters are optimized and changed as Table 7, although other parameters are the

To get the wideband of the transition, the patch cut angle *θ* is kept to 10 degrees. To achieve strong coupling, this modification of V-shaped patch element is effective. Due to this structural modification, horizontal component of electric current on the patch element increases. Consequently strong coupling to the microstrip line is achieved. The bandwidth for |*S*11| below -15 dB is 3.1 GHz and the insertion loss |*S*21| is 0.34 dB. The bandwidth is 0.4 GHz narrow than

the wideband design and the loss is approximately equal as the low loss design.

**Figure 32.** Lower pattern of the transition with modified V-shaped patch element

**Table 7.** Parameters of wideband and low loss design

**Description Name**

Cut length in *x*-direction *Cx* 0.1 Cut length in y-direction *Cy* 0.32 Length of patch element *L* 1.08 Patch cut angle θ 10 deg. Overlap length of inserted microstrip line ρ 0.27 Width of cut patch element *Wc* 0.45 Shift length of microstrip line from center of waveguide *Y* 0.37

**Value (mm)**

same as Table 5.

274 Advancement in Microstrip Antennas with Recent Applications

The photograph of the fabricated transitions are shown in Figure 33. Figure 33 (a) shows the upper plane of the substrate and is common for each design except *y*-position of the microstrip line(*Y*). As described before, the shift length *Y* of the microstrip line from the center of the waveguide is changed at each design. Figure 34 shows the comparison of three designed transitions. Refer to the bandwidth, measured results agree with the simulation results. For the insertion loss, the measured results are approximately 0.3 dB increased compared with the simulation results.

Three types of design are presented and as a compatible design of low loss and wideband, a new modified V-shape patch element is proposed. It is confirmed by experiments that the improved transition exhibits an insertion loss of 0.6 dB from 76 to 77 GHz, and a bandwidth of 4.1 % (3.15 GHz) for the reflection coefficient below -15 dB.

#### **Figure 33.** Fabricated transitons

Case 1: Low Loss design Case 2: Wideband design Case 3: Wideband and Low loss design

[5] Deslandes, D, & Wu, K. Integrated microstrip and rectangular waveguide in planar

Planar Microstrip-To-Waveguide Transition in Millimeter-Wave Band

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[6] Ho, T. Q, & Shih, Y. C. Spectral-domain analysis of E-Plane waveguide to microstrip transitions, "IEEE Trans. Microw. Theory Tech., Feb. (1989). , 37(2), 388-392.

[7] Leong, Y, & Weinreb, S. Full band waveguide to microstrip probe transitions," IEEE MTT-S Int. Microw. Symp. Dig., Anaheim, CA, May (1999). MWSYM.1999.780219, 4,

[8] Grabherr, W, Hudder, B, & Menzel, W. Microstrip to waveguide transition compati‐ ble with mm-wave integrated circuits, "IEEE Trans. Microw. Theory Tech., Sep.

[9] Iizuka, H, Watanabe, T, Sato, K, & Nisikawa, K. Millimeter-wave microstrip line to waveguide transition fabricated on a single layer dielectric substrate," IEICE Trans.

[10] Iizuka, H, Sakakibara, K, & Kikuma, N. Millimeter-Wave Transition From Wave‐ guide to Two Microstrip Lines Using Rectangular Patch Element," IEEE Trans. Mi‐

[11] Bahl, I. J, Trivedi, D. K, & , A. s Guide to Microstrip Line,"Microwaves, May (1977). ,

[12] Strohm, K. M, Bloecher, H. L, Schneider, R, & Wenger, J. Development of Future Short Range Radar Technology," Radar Conference, 2005, EURAD 2005, European,

[13] Seo, K, Sakakibara, K, & Kikuma, N. Microstrip-to-waveguide Transition using Waveguide with Large Broad-wall in Millimeter-wave Band," IEEE International Conference on Ultra-Wideband, ICUWB2010, Sep. (2010). ICUWB.2010.5614169, 1,

[14] Seo, K, Sakakibara, K, & Kikuma, N. Narrow-Wall-Connected Microstrip-to-Wave‐ guide Transition Using V-Shaped Patch Element in Millimeter-Wave Band," IEICE

crow. Theory Tech., May. (2007). TMTT.2007.895139, 55(5), 899-905.

form, "IEEE Microw. Wireless Compon. Lett., Feb. (2001). , 11(2), 68-70.

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174-182.

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Commun., Jun. (2002). , E85-B(6), 1169-1177.

Trans. Commun., Oct. (2010). , E93-B(10), 2523-2530.

**Figure 34.** Comparison of each design

#### **Author details**

Kazuyuki Seo

Address all correspondence to: kazuyuki.seo@pillar.co.jp; k\_seo1951@pure.ocn.ne.jp

Process Development Dept., Nippon Pillar Packing Co., Ltd. Sanda City, Japan

#### **References**


Case 1: Low Loss design Case 2: Wideband design

Address all correspondence to: kazuyuki.seo@pillar.co.jp; k\_seo1951@pure.ocn.ne.jp

[1] Russel, M. E, Grain, A, Curran, A, Campbell, R. A, Drubin, C. A, & Miccioli, W. F. Millimeter-wave radar sensor for automotive intelligent cruise control(ICC)," IEEE

[2] Russel, M. E, Drubin, C. A, Marinilli, A. S, & Woodington, W. G. Integrated automo‐ tive sensors, "IEEE Trans. Microw. Theory Tech., Mar. (2002). , 50(3), 674-677.

[3] Yano, H. Y, Abdelmonem, A, Liang, J. F, & Zaki, K. A. Analysis and design of micro‐ strip to waveguide transition, "IEEE Trans. Microw. Theory Tech., Dec. (1994). ,

[4] Kaneda, N, Qian, Y, & Itoh, T. A broadband microstrip-to-waveguide transition us‐ ing quasi-Yagi antenna, "IEEE Trans. Microw. Thory Tech., Dec. (1999). MWSYM.

Process Development Dept., Nippon Pillar Packing Co., Ltd. Sanda City, Japan

Trans. Microw. Thory Tech., Dec. (1997). , 45(12), 2444-2453.

Measured Simulation Measured Simulation

0 0.5 1 1.5 2 2.5 3 3.5 4

**Case 1 Case 2 Case 3 Case 1 Case 2 Case 3**

**Bandwidth [G**

**Hz]**

123

Case 3: Wideband and Low loss design

123

276 Advancement in Microstrip Antennas with Recent Applications

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

**Figure 34.** Comparison of each design

42(12), 2371-2379.

1999.780218, 47(12), 2562-2567.

**Author details**

Kazuyuki Seo

**References**

**Loss [d**

**B]**


**Chapter 12**

**Drooped Microstrip Antennas for GPS Marine and**

The Navigation Satellite Timing and Ranging (NAVSTAR) GPS is a space-based system designed primarily for global real-time, all-weather navigation. There are 30 GPS satellites in six nearly circular, approximately 20,000 kilometer orbital planes, with an inclination of 550 relative to the equator [1]. Each satellite transmits two unique, Right Hand Circularly Polarized (RHCP) L band signals. The L1 (1.57542 GHz) carrier is bi-phase modulated with two pseudorandom noise sequences; the P and C/A codes. The L2 (1.2276 GHz) carrier is modulated only with the P code and is used mainly to determine and correct phase advance caused by the ionosphere. Superimposed on the P and C/A codes is the navigation message which contains, among other things, satellite ephemerides, clock biases, and ionosphere correction data [1]. Due to their light weight, reduced size, low cost, conformability, robustness, and ease of integration with MMIC, tremendous research has been reported over the last three decades into the use of microstrip antennas in GPS navigation [2]-[20]. Antenna designers are often faced with interrelated, strict, and conflicting performance requirements in order to meet the accuracy, continuity, and integrity of differential GPS, relative geodetic and hydrographic

The design specifications of a GPS antenna depend on the performance requirements peculiar to the application under consideration. A GPS user antenna requires RHCP and adequate copolarized radiation pattern coverage over almost the entire upper hemisphere to track all visible satellites. Moreover, the antenna should ideally provide a uniform response in ampli‐ tude and, more critically, in phase to the full visible satellite constellation [21]. The angle cutoff

> © 2013 Clark et al.; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use,

© 2013 Clark et al.; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

distribution, and reproduction in any medium, provided the original work is properly cited.

**Aerospace Navigation**

Ken G. Clark, Hussain M. Al-Rizzo,

Ayman Abbosh

**1. Introduction**

http://dx.doi.org/10.5772/55002

James M. Tranquilla, Haider Khaleel and

Additional information is available at the end of the chapter

surveying, ship-borne and aerospace navigation [21]-[27].
