**Drooped Microstrip Antennas for GPS Marine and Aerospace Navigation**

Ken G. Clark, Hussain M. Al-Rizzo, James M. Tranquilla, Haider Khaleel and

Ayman Abbosh

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/55002

#### **1. Introduction**

The Navigation Satellite Timing and Ranging (NAVSTAR) GPS is a space-based system designed primarily for global real-time, all-weather navigation. There are 30 GPS satellites in six nearly circular, approximately 20,000 kilometer orbital planes, with an inclination of 550 relative to the equator [1]. Each satellite transmits two unique, Right Hand Circularly Polarized (RHCP) L band signals. The L1 (1.57542 GHz) carrier is bi-phase modulated with two pseudorandom noise sequences; the P and C/A codes. The L2 (1.2276 GHz) carrier is modulated only with the P code and is used mainly to determine and correct phase advance caused by the ionosphere. Superimposed on the P and C/A codes is the navigation message which contains, among other things, satellite ephemerides, clock biases, and ionosphere correction data [1].

Due to their light weight, reduced size, low cost, conformability, robustness, and ease of integration with MMIC, tremendous research has been reported over the last three decades into the use of microstrip antennas in GPS navigation [2]-[20]. Antenna designers are often faced with interrelated, strict, and conflicting performance requirements in order to meet the accuracy, continuity, and integrity of differential GPS, relative geodetic and hydrographic surveying, ship-borne and aerospace navigation [21]-[27].

The design specifications of a GPS antenna depend on the performance requirements peculiar to the application under consideration. A GPS user antenna requires RHCP and adequate copolarized radiation pattern coverage over almost the entire upper hemisphere to track all visible satellites. Moreover, the antenna should ideally provide a uniform response in ampli‐ tude and, more critically, in phase to the full visible satellite constellation [21]. The angle cutoff

© 2013 Clark et al.; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Clark et al.; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

and roll-off characteristics of the radiation pattern can be altered to suit the application of interest. For example, fixed ground reference stations and relative static geodetic surveying demand a rapid fall-off near the horizon, a high cross-polarization rejection, and a front-toback gain ratio in excess of 20 dB to mitigate deleterious effects of severe multipath [28], [29].

[36]. However, a fundamental distinction exists in the relationship between a patch and the ground plane when compared with helical or dipole elements in that the ground plane of a microstrip antenna forms an integral part of the radiating structure and may not best be defined

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281

Building on our previous design experiences [28], [31]-[35], our research group at the Univer‐ sity of New Brunswick, Fredericton, NB, Canada was the first to rigorously investigate the potential performance enhancements and limitations involved when these design modifica‐ tions are applied to the more appealing microstrip antenna element [37]. The advantages associated with the microstrip antenna are such that one patent has been issued to a GPS manufacturer [38] based only on a downward drooped antenna structure. Neither the dimensions nor the performance of the proposed antennas were quantified in [38]. Later, a corner truncated square patch, partially enclosed within a flatly folded conducting wall, mounted on a pyramidal ground plane, was reported in [39]. However, neither the cross

In contrast to the antenna reported in [39], the drooped microstrip antennas introduced in this chapter have the ground plane and actual element deformed such that the corners or edges of the resonant cavity region fall away from the plane occupied by the element. A fundamental understanding of the operation and limitations of the drooped microstrip antenna is still lacking. A diffraction technique was attempted in [40] to model the effects of a sloping ground plane. This, however, was limited by the difficulty of implementing a realistic source term as well as the inclusion of finite lossy dielectric materials. A rigorous full-wave 3-D model, which incorporates the coaxial feed and detailed geometrical features of the drooped microstrip

For these reasons, we have performed the research reported in this chapter, which is the first to our knowledge that combines rigorous 3-D full-wave simulations and experimental measurements to provide a comprehensive characterization of downward and upward drooped microstrip antennas. A FDTD model has been developed, validated experimentally, and used to compute the input impedance and far-field radiation patterns. The FDTD model was used to examine the effects of a wide range of structural variations to gain an insight into the benefits and limitations of the proposed antennas. The parameters of interest include the location and angle of the bend, length of the ground plane, dielectric constant, and thickness of the substrate. Prototype structures were constructed, and their characteristics measured and then compared against simulated results. The authors wish to point out that the antennas described in this chapter are not intended to target multipath mitigation; on the contrary, they demonstrate the range of pattern modifications that could be accomplished by manipulating the orientation and size of the ground plane to suit GPS applications in marine and aerospace

The rest of the chapter is organized as follows: Section 2 summarizes the FDTD algorithm developed to perform the design and parametric studies and presents results from experimen‐ tal tests performed to validate the implementation of the model and to demonstrate its ability to correctly predict the behavior of the drooped antennas. Section 3 addresses the design proce‐ dure,introducesparametric studies,anddescribes thedroopedantennasconstructedandtested for the control of the radiation patterns. Finally, Section 4 provides concluding remarks.

polarization performance nor the phase center stability were provided in [39].

as a "secondary" source.

antenna, has not yet been reported.

navigation.

In real-time kinematic positioning, few if any of the above constraints may be effective [30] and it may be necessary to operate the antenna under less than optimal conditions in regard to cross polarization performance if a wide beamwidth is of precedence. Precise GPS hydro‐ graphic surveying on a vessel cruising at speeds of 10 to 20 knots in open oceans is a challenging task due to the rotational disturbances from a relatively harsh sea environment. Pitch and roll amplitudes as high as 100 to 150 may be encountered in stormy weather, which presents a major obstacle to GPS derived attitude determination [21], [22]. Another envisaged application for the drooped microstrip antennas introduced in this chapter involves normal pitch or roll maneuvers of a general aviation aircraft, which may cause loss of some satellite signals for a range of flight orientation.

There is significant interest in the commercial and military sectors to develop antennas that could cover much of the upper hemisphere, including GPS satellites at elevation angles as low as 100 , and to extend the coverage to negative elevation angles [3], [4], [15], [21]. This will lead to fewer occurrences of cycle slips and loss of lock to satellites while rising or setting, will maintain the proper Dilution of Precision (DoP) by maximizing the number of satellites in view, and will reduce the RMS error in range and velocity [1]. Notably, on the negative side, undesired multipath reflections off water and conducting bodies are also strongest at lowelevation angles. Nevertheless, whatever type of antenna is chosen, multipath reception will still have to be dealt with as a common problem [30]. It is fair to say that no single antenna design in the open literature has satisfactorily fulfilled all the above-mentioned requirements on coverage, phase center stability, and multipath rejection for real-time highly dynamic GPS marine and aerospace applications.

A pedestal ground plane is reported in [31], based on a trial-and-error experimental design approach, consisting of a cylindrical structure with a flat elevated center surrounded by sloping sides, to address beam shaping of crossed dipoles. This structure was found to be successful in improving the pattern coverage of the crossed dipole at low-elevation angles. Additional elements were also examined such as folded, serrated, rolled edges, monofilar, and quadrafiliar helices [32]-[35], although none achieved the same degree of radiation pattern control as the crossed dipole. This is attributed in part to the extent of ground plane illumina‐ tion produced by the different sources and serves to highlight the importance of the ground plane as a secondary source with which to produce pattern changes. Further modifications to the ground plane using choke rings [28] were investigated primarily for multipath rejection.

It is well known that a stand-alone microstrip antenna mounted on a flat ground plane suffers from a lack of pattern control and reduced gain at low-elevation angles. This may result in a loss of contact with satellites when the antenna is mounted on a highly-dynamic platform, an attractive use for this low-profile structure. Fundamental to the design of a patch antenna is the interaction with the ground plane. In fact, the size, orientation, and shape of the ground plane are among the most important parameters that have an influence on the radiation pattern [36]. However, a fundamental distinction exists in the relationship between a patch and the ground plane when compared with helical or dipole elements in that the ground plane of a microstrip antenna forms an integral part of the radiating structure and may not best be defined as a "secondary" source.

and roll-off characteristics of the radiation pattern can be altered to suit the application of interest. For example, fixed ground reference stations and relative static geodetic surveying demand a rapid fall-off near the horizon, a high cross-polarization rejection, and a front-toback gain ratio in excess of 20 dB to mitigate deleterious effects of severe multipath [28], [29]. In real-time kinematic positioning, few if any of the above constraints may be effective [30] and it may be necessary to operate the antenna under less than optimal conditions in regard to cross polarization performance if a wide beamwidth is of precedence. Precise GPS hydro‐ graphic surveying on a vessel cruising at speeds of 10 to 20 knots in open oceans is a challenging task due to the rotational disturbances from a relatively harsh sea environment. Pitch and roll

obstacle to GPS derived attitude determination [21], [22]. Another envisaged application for the drooped microstrip antennas introduced in this chapter involves normal pitch or roll maneuvers of a general aviation aircraft, which may cause loss of some satellite signals for a

There is significant interest in the commercial and military sectors to develop antennas that could cover much of the upper hemisphere, including GPS satellites at elevation angles as low

A pedestal ground plane is reported in [31], based on a trial-and-error experimental design approach, consisting of a cylindrical structure with a flat elevated center surrounded by sloping sides, to address beam shaping of crossed dipoles. This structure was found to be successful in improving the pattern coverage of the crossed dipole at low-elevation angles. Additional elements were also examined such as folded, serrated, rolled edges, monofilar, and quadrafiliar helices [32]-[35], although none achieved the same degree of radiation pattern control as the crossed dipole. This is attributed in part to the extent of ground plane illumina‐ tion produced by the different sources and serves to highlight the importance of the ground plane as a secondary source with which to produce pattern changes. Further modifications to the ground plane using choke rings [28] were investigated primarily for multipath rejection. It is well known that a stand-alone microstrip antenna mounted on a flat ground plane suffers from a lack of pattern control and reduced gain at low-elevation angles. This may result in a loss of contact with satellites when the antenna is mounted on a highly-dynamic platform, an attractive use for this low-profile structure. Fundamental to the design of a patch antenna is the interaction with the ground plane. In fact, the size, orientation, and shape of the ground plane are among the most important parameters that have an influence on the radiation pattern

, and to extend the coverage to negative elevation angles [3], [4], [15], [21]. This will lead to fewer occurrences of cycle slips and loss of lock to satellites while rising or setting, will maintain the proper Dilution of Precision (DoP) by maximizing the number of satellites in view, and will reduce the RMS error in range and velocity [1]. Notably, on the negative side, undesired multipath reflections off water and conducting bodies are also strongest at lowelevation angles. Nevertheless, whatever type of antenna is chosen, multipath reception will still have to be dealt with as a common problem [30]. It is fair to say that no single antenna design in the open literature has satisfactorily fulfilled all the above-mentioned requirements on coverage, phase center stability, and multipath rejection for real-time highly dynamic GPS

may be encountered in stormy weather, which presents a major

amplitudes as high as 100

range of flight orientation.

marine and aerospace applications.

as 100

to 150

280 Advancement in Microstrip Antennas with Recent Applications

Building on our previous design experiences [28], [31]-[35], our research group at the Univer‐ sity of New Brunswick, Fredericton, NB, Canada was the first to rigorously investigate the potential performance enhancements and limitations involved when these design modifica‐ tions are applied to the more appealing microstrip antenna element [37]. The advantages associated with the microstrip antenna are such that one patent has been issued to a GPS manufacturer [38] based only on a downward drooped antenna structure. Neither the dimensions nor the performance of the proposed antennas were quantified in [38]. Later, a corner truncated square patch, partially enclosed within a flatly folded conducting wall, mounted on a pyramidal ground plane, was reported in [39]. However, neither the cross polarization performance nor the phase center stability were provided in [39].

In contrast to the antenna reported in [39], the drooped microstrip antennas introduced in this chapter have the ground plane and actual element deformed such that the corners or edges of the resonant cavity region fall away from the plane occupied by the element. A fundamental understanding of the operation and limitations of the drooped microstrip antenna is still lacking. A diffraction technique was attempted in [40] to model the effects of a sloping ground plane. This, however, was limited by the difficulty of implementing a realistic source term as well as the inclusion of finite lossy dielectric materials. A rigorous full-wave 3-D model, which incorporates the coaxial feed and detailed geometrical features of the drooped microstrip antenna, has not yet been reported.

For these reasons, we have performed the research reported in this chapter, which is the first to our knowledge that combines rigorous 3-D full-wave simulations and experimental measurements to provide a comprehensive characterization of downward and upward drooped microstrip antennas. A FDTD model has been developed, validated experimentally, and used to compute the input impedance and far-field radiation patterns. The FDTD model was used to examine the effects of a wide range of structural variations to gain an insight into the benefits and limitations of the proposed antennas. The parameters of interest include the location and angle of the bend, length of the ground plane, dielectric constant, and thickness of the substrate. Prototype structures were constructed, and their characteristics measured and then compared against simulated results. The authors wish to point out that the antennas described in this chapter are not intended to target multipath mitigation; on the contrary, they demonstrate the range of pattern modifications that could be accomplished by manipulating the orientation and size of the ground plane to suit GPS applications in marine and aerospace navigation.

The rest of the chapter is organized as follows: Section 2 summarizes the FDTD algorithm developed to perform the design and parametric studies and presents results from experimen‐ tal tests performed to validate the implementation of the model and to demonstrate its ability to correctly predict the behavior of the drooped antennas. Section 3 addresses the design proce‐ dure,introducesparametricstudies,anddescribes thedroopedantennasconstructedandtested for the control of the radiation patterns. Finally, Section 4 provides concluding remarks.

#### **2. The contour path FDTD model: Experimental validation**

The basic microstrip antenna, which consists of a conducting patch radiator rotated by 450 with respect to the center of the ground plane, and separated from the ground plane by a thin dielectric substrate, and the downward drooped antenna are shown in Figure1.

In order to compute the input impedance, the instantaneous voltage and current are calculated at a fixed location in the coaxial feed by integrating the radial electric field and the magnetic field components encircling the inner conductor of the coaxial cable. A Gaussian pulse is used, and a DFT is performed to obtain broadband results. The impedance is then translated into the ground plane aperture by standard transmission line methods. The far-field radiation patterns are next determined by driving the model to steady state, using a sinusoidal wave at

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With pulse excitation, the fields must settle toward zero as energy escapes through the absorbing boundary. The condition used to judge if steady state is reached requires that the energy monitored at several observation points within the computational domain remains below 1% of the peak observed value with lower values enforced for cases continuing to display periodic oscillations in the fields. For a sinusoidal excitation, the solution must converge to an oscillation. The magnitude and phase at several observation points are extracted from the DFT at each temporal cycle. A 1% variation in steady state is permitted in phase. The methodology we followed involves extracting the antenna characteristics for a given geometry selected from a parametric study using the FDTD code; constructing a prototype; measuring the frequency response of the input impedance, far-field radiation patterns at the measured resonant frequency; and finally comparing simulated results against measurements. To validate the operation of the FDTD model, several antenna structures were simulated and measured. The first antenna used to validate the code was a flat rectangular microstrip. The 50 mm × 47 mm patch was constructed on a square ground plane, 150 mm in side length. The substrate has a relative dielectric constant, *ε<sup>r</sup>* = 4.2 and a thickness of 1.5 mm. The frequency spectrum of the real and imaginary parts of the input impedance and the far-field radiation

**Figure 2.** Measured and calculated input impedance for a 50 mm × 47 mm microstrip on a flat, 150 mm square

The calculated impedance correctly predicts the resonant frequency measured using a network analyzer. Similarly, excellent agreement is observed between the amplitude and phase of the

the fundamental resonant frequency of the antenna.

pattern in the *E* plane are shown in Figs. 2 and 3, respectively.

measured and simulated far-field radiation patterns.

ground plane.

**Figure 1.** (a) The reference flat microstrip antenna, (b) the downward geometry, (c) cross sectional view of the reso‐ nant cavity.

The element is driven by a 50 Ω coaxial cable passing through the ground plane and the substrate. The antenna operates at the L1 GPS frequency of 1.57542 GHz since the majority of commercial GPS receivers use only the L1 frequency. Due to the relatively low cost, time savings, and repeatability, computer simulations can often complement and reduce the empiricism involved in an otherwise purely experimental approach, particularly in the initial design phase, allowing the antennas to be characterized carefully prior to their construction. Because of the complex geometries involved, the task of modeling the drooped microstrip structures is by no means a simple endeavor; it better lends itself to numerical simulation techniques. The FDTD method has been adopted in this research due to its conceptual simplicity and ease of implementation. Because it is a time-domain scheme, it is straightfor‐ ward to impose a pulse excitation to perform broadband analysis using the Discrete Fourier Transform (DFT).

The FDTD algorithm is implemented in a 3-D Cartesian coordinate system with the formula‐ tion allowing for different spatial increments along each coordinate direction. Provision is made for modeling symmetrical objects by applying the Neumann boundary condition along one surface of the computational space. The antenna is excited either by a sinusoidal signal at the resonant frequency of the dominant mode or by a Gaussian pulse with a specified width and delay. The excitation is applied to either the electric or magnetic field, depending on whether a voltage or current source is desired.

In order to compute the input impedance, the instantaneous voltage and current are calculated at a fixed location in the coaxial feed by integrating the radial electric field and the magnetic field components encircling the inner conductor of the coaxial cable. A Gaussian pulse is used, and a DFT is performed to obtain broadband results. The impedance is then translated into the ground plane aperture by standard transmission line methods. The far-field radiation patterns are next determined by driving the model to steady state, using a sinusoidal wave at the fundamental resonant frequency of the antenna.

**2. The contour path FDTD model: Experimental validation**

282 Advancement in Microstrip Antennas with Recent Applications

The basic microstrip antenna, which consists of a conducting patch radiator rotated by 450

(a) (b)

(c)

nant cavity.

Transform (DFT).

whether a voltage or current source is desired.

**Figure 1.** (a) The reference flat microstrip antenna, (b) the downward geometry, (c) cross sectional view of the reso‐

The element is driven by a 50 Ω coaxial cable passing through the ground plane and the substrate. The antenna operates at the L1 GPS frequency of 1.57542 GHz since the majority of commercial GPS receivers use only the L1 frequency. Due to the relatively low cost, time savings, and repeatability, computer simulations can often complement and reduce the empiricism involved in an otherwise purely experimental approach, particularly in the initial design phase, allowing the antennas to be characterized carefully prior to their construction. Because of the complex geometries involved, the task of modeling the drooped microstrip structures is by no means a simple endeavor; it better lends itself to numerical simulation techniques. The FDTD method has been adopted in this research due to its conceptual simplicity and ease of implementation. Because it is a time-domain scheme, it is straightfor‐ ward to impose a pulse excitation to perform broadband analysis using the Discrete Fourier

The FDTD algorithm is implemented in a 3-D Cartesian coordinate system with the formula‐ tion allowing for different spatial increments along each coordinate direction. Provision is made for modeling symmetrical objects by applying the Neumann boundary condition along one surface of the computational space. The antenna is excited either by a sinusoidal signal at the resonant frequency of the dominant mode or by a Gaussian pulse with a specified width and delay. The excitation is applied to either the electric or magnetic field, depending on

)

dielectric substrate, and the downward drooped antenna are shown in Figure1.

respect to the center of the ground plane, and separated from the ground plane by a thin

with

With pulse excitation, the fields must settle toward zero as energy escapes through the absorbing boundary. The condition used to judge if steady state is reached requires that the energy monitored at several observation points within the computational domain remains below 1% of the peak observed value with lower values enforced for cases continuing to display periodic oscillations in the fields. For a sinusoidal excitation, the solution must converge to an oscillation. The magnitude and phase at several observation points are extracted from the DFT at each temporal cycle. A 1% variation in steady state is permitted in phase.

The methodology we followed involves extracting the antenna characteristics for a given geometry selected from a parametric study using the FDTD code; constructing a prototype; measuring the frequency response of the input impedance, far-field radiation patterns at the measured resonant frequency; and finally comparing simulated results against measurements. To validate the operation of the FDTD model, several antenna structures were simulated and measured. The first antenna used to validate the code was a flat rectangular microstrip. The 50 mm × 47 mm patch was constructed on a square ground plane, 150 mm in side length. The substrate has a relative dielectric constant, *ε<sup>r</sup>* = 4.2 and a thickness of 1.5 mm. The frequency spectrum of the real and imaginary parts of the input impedance and the far-field radiation pattern in the *E* plane are shown in Figs. 2 and 3, respectively.

**Figure 2.** Measured and calculated input impedance for a 50 mm × 47 mm microstrip on a flat, 150 mm square ground plane.

The calculated impedance correctly predicts the resonant frequency measured using a network analyzer. Similarly, excellent agreement is observed between the amplitude and phase of the measured and simulated far-field radiation patterns.

**Figure 4.** Stepped approximation of shallow, intermediate, and steep bend angles. Fine adjustments are made by varying the Δx/Δz ratio. The contour integral correction is made to adjacent *H* fields to define the actual surface.

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A rectangular microstrip with two 450 drooped edges fed by a coaxial cable was constructed and tested to compare measured and simulated input impedance, *E* and *H* plane patterns. The geometry of the antenna is shown in Figure 5 along with the phase of the *E* plane elevation cut. The measured phase displayed in Figure 5 showed a slight asymmetry due to the offset in antenna mount necessary to accommodate the bends and the connector. For comparison, we referenced the calculated far-field patterns to the same offset origin. The *E* plane and *H* plane patterns shown in Figs. 5 and 6 along with the impedance of Figure 7 reveal good

**Figure 5.** *E* plane (x-z plane) elevation phase for a 55 mm × 47 mm microstrip with a 45° bend, centered on a 83 mm ×

Before progressing to the double-bend antenna, a test was conducted to verify that the FDTD model maintained continuity at the point where the stepped approximation changed from a 2:1 to a 1:1 ratio. To accomplish this, we modeled a 40 mm × 50 mm bent microstrip antenna

using both ratios. A comparison of the input impedance and

correspondences between measured and simulated results.

75 mm ground plane.

near the change of an angle of 550

**Figure 3.** Normalized amplitude and phase of the far-field radiation pattern in the *E* plane for a 50 mm × 47 mm flat microstrip on a 150 mm square ground plane.

As a progression toward the drooped structure, a microstrip antenna with two edges bent down at an angle of 450 was modeled. This structure requires special consideration in that the boundary conditions imposed by the element and ground plane do not fall along the coordi‐ nate planes. The most straightforward approach is to discretize the sloping sides with the traditional stair-stepped approximation. However, it has been observed that this can result in a slight change in the resonant frequency of high *Q* structures [41]. To avoid errors associated with the conventional FDTD method, we used the contour path method introduced in [42]. In this approach, a field component adjacent to a boundary is not updated in terms of the spatial derivatives of the surrounding fields but by an integration of adjacent fields along the perimeter of the cell. This allows for partial or deformed cells, thus better approximating the drooped surfaces.

Three different step approximations, shown in Figure 4, were used to model the sloping sides, depending upon the angle of the bend. *For fine adjustments*, the ratio of the vertical to horizon‐ tal spatial increments is adjusted to yield the desired slope angle. This has an additional benefit of simplifying the implementation of the contour method when applied to field components adjacent to the sloped surfaces. Because each cell is truncated in the same way, the contour corrections proceeded without the need to calculate intercepts of the actual slope line at each cell.

**Figure 4.** Stepped approximation of shallow, intermediate, and steep bend angles. Fine adjustments are made by varying the Δx/Δz ratio. The contour integral correction is made to adjacent *H* fields to define the actual surface.

A rectangular microstrip with two 450 drooped edges fed by a coaxial cable was constructed and tested to compare measured and simulated input impedance, *E* and *H* plane patterns. The geometry of the antenna is shown in Figure 5 along with the phase of the *E* plane elevation cut. The measured phase displayed in Figure 5 showed a slight asymmetry due to the offset in antenna mount necessary to accommodate the bends and the connector. For comparison, we referenced the calculated far-field patterns to the same offset origin. The *E* plane and *H* plane patterns shown in Figs. 5 and 6 along with the impedance of Figure 7 reveal good correspondences between measured and simulated results.

As a progression toward the drooped structure, a microstrip antenna with two edges bent down at an angle of 450 was modeled. This structure requires special consideration in that the boundary conditions imposed by the element and ground plane do not fall along the coordi‐ nate planes. The most straightforward approach is to discretize the sloping sides with the traditional stair-stepped approximation. However, it has been observed that this can result in a slight change in the resonant frequency of high *Q* structures [41]. To avoid errors associated with the conventional FDTD method, we used the contour path method introduced in [42]. In this approach, a field component adjacent to a boundary is not updated in terms of the spatial derivatives of the surrounding fields but by an integration of adjacent fields along the perimeter of the cell. This allows for partial or deformed cells, thus better approximating the

**Figure 3.** Normalized amplitude and phase of the far-field radiation pattern in the *E* plane for a 50 mm × 47 mm flat

Three different step approximations, shown in Figure 4, were used to model the sloping sides, depending upon the angle of the bend. *For fine adjustments*, the ratio of the vertical to horizon‐ tal spatial increments is adjusted to yield the desired slope angle. This has an additional benefit of simplifying the implementation of the contour method when applied to field components adjacent to the sloped surfaces. Because each cell is truncated in the same way, the contour corrections proceeded without the need to calculate intercepts of the actual slope line at each cell.

drooped surfaces.

microstrip on a 150 mm square ground plane.

284 Advancement in Microstrip Antennas with Recent Applications

**Figure 5.** *E* plane (x-z plane) elevation phase for a 55 mm × 47 mm microstrip with a 45° bend, centered on a 83 mm × 75 mm ground plane.

Before progressing to the double-bend antenna, a test was conducted to verify that the FDTD model maintained continuity at the point where the stepped approximation changed from a 2:1 to a 1:1 ratio. To accomplish this, we modeled a 40 mm × 50 mm bent microstrip antenna near the change of an angle of 550 using both ratios. A comparison of the input impedance and

substrate, *εr* = 2.2, and a 300

radiation pattern by manipulating the droop parameters.

droop angle. Results obtained for the real and imaginary parts of

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the input impedance and far-field radiations patterns, as shown in Figs. 9 and 10, respectively, display good agreement between simulated and measured results. The excellent agreement demonstrated thus far between the amplitude and phase of the simulated and measured farfield radiation patterns prompted further exploration of the possibility of controlling the

**Figure 8.** *E* plane pattern and impedance comparison for 50 mm × 40 mm microstrip (ε*<sup>r</sup>* = 2) with a 55° bend, calculat‐

ed for a 1:1 and 2:1 step approximation of the sloped sides.

**Figure 6.** *H* plane elevation pattern for a 55 mm × 47 mm microstrip with a 45° bend, 10 mm from each short side, magnitude (top) and phase (bottom).

**Figure 7.** Measured and calculated input impedance for a 47 mm × 55 mm microstrip on a 45° bent ground plane (33 mm flat top with 11 mm bent over each angle).

elevation patterns depicted in Figure 8 show good agreement between results obtained from each approximation.

To duplicate the bend on the two remaining sides to achieve the full drooped structure, we constructed two drooped antennas and used them to verify the performance of the completed model. The first antenna, shown in Figure 9, consists of a 62 mm × 62 mm patch, a 40 mm square elevated section, printed on a 1.5 mm thick substrate, *εr* = 4.2, and a 600 droop angle. The second is a 64 mm × 64 mm patch, a 50 mm square elevated section, printed on a 3 mm substrate, *εr* = 2.2, and a 300 droop angle. Results obtained for the real and imaginary parts of the input impedance and far-field radiations patterns, as shown in Figs. 9 and 10, respectively, display good agreement between simulated and measured results. The excellent agreement demonstrated thus far between the amplitude and phase of the simulated and measured farfield radiation patterns prompted further exploration of the possibility of controlling the radiation pattern by manipulating the droop parameters.

**Figure 8.** *E* plane pattern and impedance comparison for 50 mm × 40 mm microstrip (ε*<sup>r</sup>* = 2) with a 55° bend, calculat‐ ed for a 1:1 and 2:1 step approximation of the sloped sides.

elevation patterns depicted in Figure 8 show good agreement between results obtained from

**Figure 7.** Measured and calculated input impedance for a 47 mm × 55 mm microstrip on a 45° bent ground plane (33

**Figure 6.** *H* plane elevation pattern for a 55 mm × 47 mm microstrip with a 45° bend, 10 mm from each short side,

To duplicate the bend on the two remaining sides to achieve the full drooped structure, we constructed two drooped antennas and used them to verify the performance of the completed model. The first antenna, shown in Figure 9, consists of a 62 mm × 62 mm patch, a 40 mm

The second is a 64 mm × 64 mm patch, a 50 mm square elevated section, printed on a 3 mm

droop angle.

square elevated section, printed on a 1.5 mm thick substrate, *εr* = 4.2, and a 600

each approximation.

mm flat top with 11 mm bent over each angle).

magnitude (top) and phase (bottom).

286 Advancement in Microstrip Antennas with Recent Applications

**3. Parametric analysis of the drooped microstrip antennas**

reduction.

anechoic chamber measurements in a 50

for sub-centimeter static geodetic positions [43].

**3.1. Drooped microstrip with a downward bend**

to an ideal hemisphere in a 50

10, 30, or 50 mm2

Further parametric analyses were conducted to determine the range of structural variations that can be utilized to optimize the performance of the drooped microstrip antennas. The design process was complicated due to the several interacting parameters that must be considered in order to provide adequate low-angle coverage, uniform phase response, and polarization purity. Parameters investigated include the angle and location of the bend, length of the ground plane skirt, thickness and dielectric constant of the substrate. The results are assessed on the basis of their impact on antenna gain at bore sight, phase performance in the upper hemisphere, pattern beamwidth, cross polarization rejection, and near horizon gain

In all cases to be presented, the radiation patterns were obtained at the resonant frequency of the dominant mode. The amplitude and phase of the radiation patterns were obtained from

These were then analyzed to determine the 3 dB beamwidth and near horizon gain reduction with respect to bore sight (zenith). In order to determine the absolute gain, we integrated the calculated patterns over the upper and lower hemispheres with a 50 step in azimuth and elevation. All field calculations were then referenced to the resulting isotropic power density.

Whereas an enormous volume of literature is available on patch antennas for GPS applications [2]-[20], a close scrutiny revealed that the design objectives in the majority of these studies and the performance characterization were based entirely on the amplitude of the co- and crosspolarized radiation patterns. Only few have considered the phase response as a figure of merit in the design process and/or in the analysis or measurements [28], [31]-[35], [43]. The phase response directly weighs the arriving signals and produces a phase-shaping effect, which depends on the angle of arrival of the satellite signals. The calibration of the phase response provides invaluable information regarding the level of accuracy one can ultimately achieve

The measured upper hemispherical phase response of the antennas under test was matched

ideal hemisphere was adjusted to minimize the RMS error between the measured and ideal phase [43]. The origin of this hemisphere is defined as the "center of best fit" or "phase center," and the difference between the measured and ideal phase is defined as the "phase residual" or "phase error." The RMS value of the phase error is used as a figure of merit to describe the

For the initial test, a 40 mm × 40 mm patch was placed on three different ground planes. These, in turn, were bent at three different distances from the center, forming a flat square top that was

. Simulations were carried out for bend angles, ranging from 0 to 900

phase distortion introduced by the antennas considered in the rest of this chapter.

Figure 11 depicts the parameters and geometries used for the initial set of simulations.

× 50 grid, using equal solid angle weighting. The position of the

in a 150

step.

grid, following the procedure described in [43].

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289

× 50

**Figure 9.** Measured and calculated input impedance for a 60° double bend microstrip.

**Figure 10.** Measured and calculated elevation patterns for 64 mm square microstrip with a 300 bend, ε*r* = 2.2, *E* plane (*x-z*) top and *H* plane (*y-z*) bottom (1550 MHz).

### **3. Parametric analysis of the drooped microstrip antennas**

Further parametric analyses were conducted to determine the range of structural variations that can be utilized to optimize the performance of the drooped microstrip antennas. The design process was complicated due to the several interacting parameters that must be considered in order to provide adequate low-angle coverage, uniform phase response, and polarization purity. Parameters investigated include the angle and location of the bend, length of the ground plane skirt, thickness and dielectric constant of the substrate. The results are assessed on the basis of their impact on antenna gain at bore sight, phase performance in the upper hemisphere, pattern beamwidth, cross polarization rejection, and near horizon gain reduction.

In all cases to be presented, the radiation patterns were obtained at the resonant frequency of the dominant mode. The amplitude and phase of the radiation patterns were obtained from anechoic chamber measurements in a 50 × 50 grid, following the procedure described in [43]. These were then analyzed to determine the 3 dB beamwidth and near horizon gain reduction with respect to bore sight (zenith). In order to determine the absolute gain, we integrated the calculated patterns over the upper and lower hemispheres with a 50 step in azimuth and elevation. All field calculations were then referenced to the resulting isotropic power density.

Whereas an enormous volume of literature is available on patch antennas for GPS applications [2]-[20], a close scrutiny revealed that the design objectives in the majority of these studies and the performance characterization were based entirely on the amplitude of the co- and crosspolarized radiation patterns. Only few have considered the phase response as a figure of merit in the design process and/or in the analysis or measurements [28], [31]-[35], [43]. The phase response directly weighs the arriving signals and produces a phase-shaping effect, which depends on the angle of arrival of the satellite signals. The calibration of the phase response provides invaluable information regarding the level of accuracy one can ultimately achieve for sub-centimeter static geodetic positions [43].

The measured upper hemispherical phase response of the antennas under test was matched to an ideal hemisphere in a 50 × 50 grid, using equal solid angle weighting. The position of the ideal hemisphere was adjusted to minimize the RMS error between the measured and ideal phase [43]. The origin of this hemisphere is defined as the "center of best fit" or "phase center," and the difference between the measured and ideal phase is defined as the "phase residual" or "phase error." The RMS value of the phase error is used as a figure of merit to describe the phase distortion introduced by the antennas considered in the rest of this chapter.

#### **3.1. Drooped microstrip with a downward bend**

**Figure 9.** Measured and calculated input impedance for a 60° double bend microstrip.

288 Advancement in Microstrip Antennas with Recent Applications

**Figure 10.** Measured and calculated elevation patterns for 64 mm square microstrip with a 300 bend, ε*r* = 2.2, *E* plane

(*x-z*) top and *H* plane (*y-z*) bottom (1550 MHz).

For the initial test, a 40 mm × 40 mm patch was placed on three different ground planes. These, in turn, were bent at three different distances from the center, forming a flat square top that was 10, 30, or 50 mm2 . Simulations were carried out for bend angles, ranging from 0 to 900 in a 150 step. Figure 11 depicts the parameters and geometries used for the initial set of simulations.

**Figure 11.** Initial structural variations included three bend locations at 5, 10, and 25 mm from the patch center, droop angles ranging from 0° to 90° and 3 ground planes having flat (unbent) dimensions of 60, 80, 100 mm. Three sub‐ strate materials with relative permittivity of 2.2, 4.2, and 10.

Three dielectric constants were used in the course of the simulations to examine the effects of different substrate materials; values of 2.2 and 4.2 were selected because of the availability of substrates to construct the verification cases, while an *ε<sup>r</sup>* of 10 was chosen as an example of a ceramic substrate frequently used in industry. Copper tape applied to bulk Teflon, one eighth inch thick, was used to form several fixed and adjustable structures. A standard polyesterbased circuit board was used to construct others.

A sequence of elevation patterns is presented in Figure 12, which displays the progression of the radiated fields with the size and bend angle of the ground plane. Not unexpectedly, smaller ground planes with larger bends allow more energy to escape off the back, until it appears that the main beam is 1800 from the bore sight direction. This is made obvious by plotting the polar patterns for *Eθ*, using a linearly polarized excitation. With a circularly polarized excitation, backward radiated energy appears predominately in the cross polarized component, making the effect less noticeable. Notably, all subsequent results for gain, beamwidth, or phase are presented while exciting circular polarization as this would be the normal operating mode of the antenna.

**Figure 12.** Elevation patterns for bend angle variations of 0° to 90° with 60 (left), 80 (center), and 100 (right) mm ground planes. *E*θ component, ε*r* = 2.2, all with 30 mm top. All patterns are normalized to 0 dB maximum, 10 dB/divi‐

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Next, a prototype was constructed with adjustable bend plates, and the measured beamwidths were compared against simulated results. For construction simplicity, the antenna was excited using linear polarization with only two sloped sides. In Figure 13, the measured and computed

beamwidths show a slight improvement with increasing bend angle.

sion.

**Figure 11.** Initial structural variations included three bend locations at 5, 10, and 25 mm from the patch center, droop angles ranging from 0° to 90° and 3 ground planes having flat (unbent) dimensions of 60, 80, 100 mm. Three sub‐

Three dielectric constants were used in the course of the simulations to examine the effects of different substrate materials; values of 2.2 and 4.2 were selected because of the availability of substrates to construct the verification cases, while an *ε<sup>r</sup>* of 10 was chosen as an example of a ceramic substrate frequently used in industry. Copper tape applied to bulk Teflon, one eighth inch thick, was used to form several fixed and adjustable structures. A standard polyester-

A sequence of elevation patterns is presented in Figure 12, which displays the progression of the radiated fields with the size and bend angle of the ground plane. Not unexpectedly, smaller ground planes with larger bends allow more energy to escape off the back, until it appears that

patterns for *Eθ*, using a linearly polarized excitation. With a circularly polarized excitation, backward radiated energy appears predominately in the cross polarized component, making the effect less noticeable. Notably, all subsequent results for gain, beamwidth, or phase are presented while exciting circular polarization as this would be the normal operating mode of

from the bore sight direction. This is made obvious by plotting the polar

strate materials with relative permittivity of 2.2, 4.2, and 10.

290 Advancement in Microstrip Antennas with Recent Applications

based circuit board was used to construct others.

the main beam is 1800

the antenna.

**Figure 12.** Elevation patterns for bend angle variations of 0° to 90° with 60 (left), 80 (center), and 100 (right) mm ground planes. *E*θ component, ε*r* = 2.2, all with 30 mm top. All patterns are normalized to 0 dB maximum, 10 dB/divi‐ sion.

Next, a prototype was constructed with adjustable bend plates, and the measured beamwidths were compared against simulated results. For construction simplicity, the antenna was excited using linear polarization with only two sloped sides. In Figure 13, the measured and computed beamwidths show a slight improvement with increasing bend angle.

**<sup>ε</sup>***r=2.2* Boresight gain

7.2 7.1 6.9 6.6 5.5 5.2 0.1

7.2 7.0 7.1 6.9 6.8 7.1 6.4

**<sup>ε</sup>***r=4.2* Boresight gain

4.7 4.5 4.8 3.4 0.9 -1.1 -27

4.7 4.5 4.6 4.9 3.5 3.0 1.0

(dBi)

5.8 5.7 5.9 5.2 4.3 1.5 -28

5.8 6.0 5.9 5.8 4.4 4.3 1.5

Bend angle (30 mm top)

Bend angle (50 mm top)

Bend angle (10 mm top)

Bend angle (30 mm top)

(dBi)

7.5 7.5 7.4 7.1 6.7 6.4 2.3

7.5 7.7 7.5 7.2 7.2 7.2 6.9

**Table 1.** Results for the drooped microstrip, substrate permittivity 2.2.

6.2 6.0 6.0 6.5 6.1 3.3 -29

6.2 6.0 6.0 5.9 5.3 4.8 2.4

7.7 7.8 7.4 7.0 6.5 6.2 3.78

> 7.7 7.9 7.4 7.1 7.0 6.8 6.9

3 dB Beamwidth (deg)

3 dB Beamwidth (deg)

Width-mm 60 80 100 60 80 100 60 80 100 60 80 100



Width-mm 60 80 100 60 80 100 60 80 100 60 80 100



Near horizon gain Roll-off (dB)

Drooped Microstrip Antennas for GPS Marine and Aerospace Navigation



Near horizon gain Roll-off (dB)

> -13.5 -10.8 -11.0 -9.0 -5.7 0.8 -

> -13.5 -13.8 -13.3 -13.4 -11.4 -10.9 -8.2



1.6 2.2 1.5 2.0 3.6 12.8 -

> 1.6 1.2 0.9 1.2 1.6 0.9 1.6



2.9 2.4 1.5 1.9 2.3 1.6 8.2

RMS phase error (deg)

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2.6 3.3 1.8 1.6 6.5 4.4 2.7

2.6 3.3 1.8 1.6 6.5 4.4 2.7

RMS phase error (deg)

> 1.2 3.6 1.0 1.5 2.7 2.9 -

> 1.2 1.1 0.5 0.8 1.9 0.9 1.5

1.1 4.3 1.0 1.2 2.6 3.3 -

1.1 0.9 0.5 0.7 1.3 0.8 1.4

2.0 3.4 1.7 2.0 2.6 2.8 2.8 293

2.0 3.4 1.7 2.0 2.6 2.8 2.8

**Figure 13.** Measured and computed 3 dB beamwidth for an adjustable bend microstrip antenna.

Variations in bore sight gain, 3 dB beamwidth, near-horizon gain roll-off, and RMS phase error as a result of increasing the bend angle are summarized in Tables 1 to 3. The 10 mm case shows a sharp gain reduction at extreme bend angles. The results reveal a relatively minor improve‐ ment in the pattern beamwidth, even for large bend angles and varying ground plane sizes for the higher dielectric substrates. Indeed, the radiation patterns above the horizon remain virtually unchanged for bend angles up to 600 .


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**Table 1.** Results for the drooped microstrip, substrate permittivity 2.2.

**Figure 13.** Measured and computed 3 dB beamwidth for an adjustable bend microstrip antenna.

virtually unchanged for bend angles up to 600

292 Advancement in Microstrip Antennas with Recent Applications

(dBi)

7.5 7.5 7.1 6.5 5.6 3.3 -25

7.7 7.7 7.1 6.4 6.2 5.4 -26

**<sup>ε</sup>***r=2.2* Boresight gain

7.2 7.1 6.9 5.5 0.9 0.8 -24

Bend angle (10 mm top)

Variations in bore sight gain, 3 dB beamwidth, near-horizon gain roll-off, and RMS phase error as a result of increasing the bend angle are summarized in Tables 1 to 3. The 10 mm case shows a sharp gain reduction at extreme bend angles. The results reveal a relatively minor improve‐ ment in the pattern beamwidth, even for large bend angles and varying ground plane sizes for the higher dielectric substrates. Indeed, the radiation patterns above the horizon remain

.

Width-mm 60 80 100 60 80 100 60 80 100 60 80 100


Near horizon gain Roll-off (dB)

> -12. -11. -10. -9.1 -9.5 -9.3 23


3.0 2.5 1.5 2.2 4.3 12. 11.9

RMS phase error (deg)

> 2.6 3.3 1.8 1.6 6.5 4.4 2.7

2.0 3.4 1.7 2.0 2.6 2.8 2.8

3 dB Beamwidth (deg)



The phase of the elevation cuts shown in Figure 14 shows a remarkable change in the belowhorizon phase. For small bend angles, the phase diminishes from the bore sight value when approaching the horizon, while at higher bends the phase increases from the bore sight value.

above the horizon. Bending of the structure in this manner could provide an additional beamwidth and, more importantly, the phase stability necessary to achieve a design specifi‐

**Figure 14.** Elevation phase patterns for different bend angles (00, 150, 300, 600, and 700), with a 100 mm ground plane

Another consideration for GPS antennas is the cross polarization behavior. Odd reflections from nearby objects tend to be orthogonally polarized. Hence, it is important that the antenna be able to reject these, particularly near the horizon. To demonstrate the effect of bending the structure on the polarization performance, we examined the ratio of the right- to the left-hand

**Figure 15.** Cross polarization rejection (Defined as the ratio of the cross-polarized to the co-polarized components) for

different bend angles (00, 150, 300, 600, and 900); top, ε*r* = 4.2, bottom, ε*r* = 2.2.

cation, particularly if further modification of the substrate is not feasible.

of bend, a region exists where the elevation phase remains relatively constant

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295

At about 150

to 300

and a 30 mm flat (ε*r* = 4.2).

**Table 2.** Results for the drooped microstrip, substrate permittivity 4.2.


**Table 3.** Results for the drooped microstrip, substrate permittivity 10.

The phase of the elevation cuts shown in Figure 14 shows a remarkable change in the belowhorizon phase. For small bend angles, the phase diminishes from the bore sight value when approaching the horizon, while at higher bends the phase increases from the bore sight value. At about 150 to 300 of bend, a region exists where the elevation phase remains relatively constant above the horizon. Bending of the structure in this manner could provide an additional beamwidth and, more importantly, the phase stability necessary to achieve a design specifi‐ cation, particularly if further modification of the substrate is not feasible.

**<sup>ε</sup>***r=4.2* Boresight gain

4.8 4.9 5.3 5.0 4.7 5.2 4.4

**<sup>ε</sup>***r=10* Boresight gain

3.1 3.4 3.8 2.5 3.3 2.4 -38

3.1 3.8 3.8 2.9 2.7 2.3 1.5

3.1 2.9 3.2 3.4 3.0 3.7 2.0

(dBi)

4.7 4.4 4.4 3.6 2.6 3.0 -40

4.7 4.5 5.0 4.1 3.6 2.9 1.4

4.7 4.6 4.9 4.5 3.7 3.6 2.6

Bend angle (50 mm top)

Bend angle (10 mm top)

Bend angle (30 mm top)

Bend angle (50 mm top)

(dBi)

294 Advancement in Microstrip Antennas with Recent Applications

5.8 5.9 6.0 5.8 5.6 5.5 5.2 6.2 6.3 6.2 6.0 5.8 5.4 5.4

**Table 2.** Results for the drooped microstrip, substrate permittivity 4.2.

5.3 5.2 5.1 4.5 3.6 3.5 -43

5.3 4.9 5.4 4.7 4.3 3.5 1.5

5.3 5.4 5.5 4.8 4.9 4.5 3.2

**Table 3.** Results for the drooped microstrip, substrate permittivity 10.

3 dB Beamwidth (deg)

3 dB Beamwidth (deg)

Width-mm 60 80 100 60 80 100 60 80 100 60 80 100




Width-mm 60 80 100 60 80 100 60 80 100 60 80 100


Near horizon gain Roll-off (dB)

> -13.5 -13.6 -13.6 -13.4 -12.8 -12.5 -12.6

Near horizon gain Roll-off (dB)

> -11.5 -10.4 -10.7 -9.5 -8.3 -8.5 -

> -11.5 -11.3 -12.1 -10.3 -11.1 -8.6 -6.1

> -11.5 -10.7 -12.4 -10.7 -8.9 -9.1 -9.2




0.80 0.40 0.25 0.39 0.56 0.24 0.29

0.80 0.34 0.46 0.37 0.51 0.28 2.1


RMS phase error (deg)

> 0.9 0.7 0.5 0.6 0.6 0.4 0.6

RMS phase error (deg)

> 0.49 1.0 0.46 0.41 0.80 0.72 -

> 0.49 0.46 0.33 0.32 1.2 0.21 0.31

> 0.49 0.70 0.90 0.56 0.51 2.2 2.4

0.53 0.87 0.42 0.38 0.86 0.97 -

0.53 0.48 0.36 0.30 1.2 0.18 0.34

.53 0.80 0.32 0.41 0.90 1.5 2.2

1.2 0.9 0.6 0.6 1.0 0.5 0.6

**Figure 14.** Elevation phase patterns for different bend angles (00, 150, 300, 600, and 700), with a 100 mm ground plane and a 30 mm flat (ε*r* = 4.2).

Another consideration for GPS antennas is the cross polarization behavior. Odd reflections from nearby objects tend to be orthogonally polarized. Hence, it is important that the antenna be able to reject these, particularly near the horizon. To demonstrate the effect of bending the structure on the polarization performance, we examined the ratio of the right- to the left-hand

**Figure 15.** Cross polarization rejection (Defined as the ratio of the cross-polarized to the co-polarized components) for different bend angles (00, 150, 300, 600, and 900); top, ε*r* = 4.2, bottom, ε*r* = 2.2.

circular components for several bend angles. Almost without exception, the bend degraded the cross polarization rejection near the horizon as shown in Figure 15, making the antenna more susceptible to spurious signals.

shows the gain and the 3-dB beamwidth for the upward bend cases. As seen, a noticeable beam broadening is evident at the higher bend angles with 3-dB beamwidths up to 60% greater than the equivalent flat case. One can observe, however, a distinct reduction in the beamwidth for the initial small bend angles, particularly for larger ground planes. The beamwidth did not

Drooped Microstrip Antennas for GPS Marine and Aerospace Navigation

A prototype antenna was constructed and tested to allow comparison against experimental results. The antenna is identical to the one shown in Figure 13; only two variable upward bends were used in this case. For measurement purposes, a linearly polarized excitation was used. The initial dip in the beamwidth did not occur in the simulated and measured results shown in Fig. 18.

.

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297

recover to that of the equivalent downward case until the bend angle exceeded 600

**Figure 18.** *E* plane elevation pattern and beamwidth behavior of the adjustable upward bend antenna.

at the 30 mm position. The results are summarized in Tables 4 and 5.

elevation fell within 300

phase curves above 900

nation near the horizon.

With the completion of the initial set of simulations and measurements, we pursued the model with additional parameter variations. Bend locations at the 10 and 50 mm positions were tested for the *εr* = 2.2 substrate. Also, the *εr* = 4.2 substrate was examined when the bend was positioned

A changing upward bend produced little effect on the phase in the upper hemisphere. All

Like the downward bend, the RMS phase error over the hemisphere did not vary substantially with bend angle, although a slightly greater differentiation is evident between different ground plane sizes. The elevation gain and cross polarization rejection for various upward bend angles are displayed in Figs. 19 and 20, respectively. When compared to Figure 15, it is apparent that the rejection is better by 2 dB near the horizon for upward bends, but the opposite is true below the horizon. In both cases, bending the ground plane reduces the cross-polarization discrimi‐

of each other over the full range of bend angles.

**Figure 16.** Geometry of the upward bend antenna.

#### **3.2. Drooped microstrip with an upward bend**

The modification of the FDTD model to accommodate upward bends was accomplished by interchanging the positions of the ground plane and the element. The structure would then be upside down in the computational space but would have an upward bend. Initially, we considered the antenna shown in Figure16, which has a 30 mm flat top on a substrate with *ε<sup>r</sup>* = 2.2. Three ground plane sizes were analyzed with bend angles varied up to 900 . Figure 17

**Figure 17.** Gain and beam width variation with upward bend angle for three ground plane sizes, ε*<sup>r</sup>* = 2.2. (Bend form‐ ing a 30 mm flat center section).

shows the gain and the 3-dB beamwidth for the upward bend cases. As seen, a noticeable beam broadening is evident at the higher bend angles with 3-dB beamwidths up to 60% greater than the equivalent flat case. One can observe, however, a distinct reduction in the beamwidth for the initial small bend angles, particularly for larger ground planes. The beamwidth did not recover to that of the equivalent downward case until the bend angle exceeded 600 .

circular components for several bend angles. Almost without exception, the bend degraded the cross polarization rejection near the horizon as shown in Figure 15, making the antenna

The modification of the FDTD model to accommodate upward bends was accomplished by interchanging the positions of the ground plane and the element. The structure would then be upside down in the computational space but would have an upward bend. Initially, we considered the antenna shown in Figure16, which has a 30 mm flat top on a substrate with *ε<sup>r</sup>*

**Figure 17.** Gain and beam width variation with upward bend angle for three ground plane sizes, ε*<sup>r</sup>* = 2.2. (Bend form‐

. Figure 17

= 2.2. Three ground plane sizes were analyzed with bend angles varied up to 900

more susceptible to spurious signals.

296 Advancement in Microstrip Antennas with Recent Applications

**Figure 16.** Geometry of the upward bend antenna.

ing a 30 mm flat center section).

**3.2. Drooped microstrip with an upward bend**

A prototype antenna was constructed and tested to allow comparison against experimental results. The antenna is identical to the one shown in Figure 13; only two variable upward bends were used in this case. For measurement purposes, a linearly polarized excitation was used. The initial dip in the beamwidth did not occur in the simulated and measured results shown in Fig. 18.

**Figure 18.** *E* plane elevation pattern and beamwidth behavior of the adjustable upward bend antenna.

With the completion of the initial set of simulations and measurements, we pursued the model with additional parameter variations. Bend locations at the 10 and 50 mm positions were tested for the *εr* = 2.2 substrate. Also, the *εr* = 4.2 substrate was examined when the bend was positioned at the 30 mm position. The results are summarized in Tables 4 and 5.

A changing upward bend produced little effect on the phase in the upper hemisphere. All phase curves above 900 elevation fell within 300 of each other over the full range of bend angles. Like the downward bend, the RMS phase error over the hemisphere did not vary substantially with bend angle, although a slightly greater differentiation is evident between different ground plane sizes. The elevation gain and cross polarization rejection for various upward bend angles are displayed in Figs. 19 and 20, respectively. When compared to Figure 15, it is apparent that the rejection is better by 2 dB near the horizon for upward bends, but the opposite is true below the horizon. In both cases, bending the ground plane reduces the cross-polarization discrimi‐ nation near the horizon.


Bend angle (deg.)

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299

**Figure 19.** Gain variation with upward bend angle for three ground plane sizes, εr = 4.2. (Bend forming a 30 mm flat

**Figure 20.** Cross polarization rejection (Defined as the ratio of the cross-polarized to the co-polarized components) for

In this chapter, we have presented numerical simulations and experimental measurements to analyze downward and upward drooped microstrip antennas with the intent of modifying the radiation pattern of the basic planar patch to accommodate the coverage requirements of GPS marine navigation and positioning. Magnitude and phase of the simulated and measured far-field radiation patterns are presented to reveal the tradeoffs in performance between patch geometry, ground plane size, and orientation. Results reported for the wide range of structural variations applied to the base antenna along with changes in the substrate material should be

It has been found that an accurate and stable phase center can be obtained over the entire hemisphere for moderate upward bends. Numerical simulations and measurements demon‐ strate that the 3-dB beamwidth of the flat microstrip patch can be increased by at least 15% and 60% for the downward and upward bends, respectively. The phase stability demonstrated by the slightly bent structures may be viewed as advantageous in cases where circumstances require distorting the element but where a significant alteration in the pattern is not desired. These were accomplished, however, at the expense of some loss of the low-profile character

different bend angles (00, 150, 450, 600, and 900), 100 mm ground plane, ε*r* = 2.2.

valuable to designers seeking to achieve a specific coverage performance.

center section).

**4. Concluding remarks**

of the antenna.

**Table 4.** Results for the drooped microstrip, substrate permittivity 2.2.


**Table 5.** Results for the drooped microstrip, substrate permittivity 4.2.

**Figure 19.** Gain variation with upward bend angle for three ground plane sizes, εr = 4.2. (Bend forming a 30 mm flat center section).

**Figure 20.** Cross polarization rejection (Defined as the ratio of the cross-polarized to the co-polarized components) for different bend angles (00, 150, 450, 600, and 900), 100 mm ground plane, ε*r* = 2.2.

#### **4. Concluding remarks**

**<sup>ε</sup>***r=2.2* Boresight gain

7.2 4.0 4.7 5.1 4.1 2.5 -1.5

7.2 7.7 7.1 6.6 6.0 4.9 5.1

7.2 7.3 7.0 7.3 7.1 6.9 6.5

**<sup>ε</sup>r=4.2** Bore-sight gain

4.7 6.6 5.1 5.6 4.5 3.4 4.7

(dBi)

5.8 7.0 5.4 6.0 4.9 4.2 4.7

Bend angle (10 mm top)

Bend angle (30 mm top)

Bend angle (50 mm top)

Bend angle (30 mm top)

(dBi)

298 Advancement in Microstrip Antennas with Recent Applications

7.5 6.2 6.5 3.5 -.8 -4.4 -6.1

7.5 8.4 7.6 7.0 6.2 4.9 0.6

7.5 7.6 7.4 7.5 6.8 6.8 5.8

7.7 6.4 1.3 3.5 2.6 -.42 -6.4

7.7 8.7 7.7 7.3 6.0 4.9 3.7

7.7 7.9 7.9 7.8 7.1 6.9 5.1

**Table 4.** Results for the drooped microstrip, substrate permittivity 2.2.

6.2 7.5 5.6 6.4 5.1 4.3 4.4

**Table 5.** Results for the drooped microstrip, substrate permittivity 4.2.

3 dB Beamwidth (deg)

3 dB Beamwidth (deg)

Width-mm 60 80 100 60 80 100 60 80 100 60 80 100


Width-mm 60 80 100 60 80 100 60 80 100 60 80 100




Near horizon gain Roll-off (dB)

> -12. -11. -10. -7.6 -6.7 5.8 -6.6

> -12. -13. -12. -9.8 -7.8 -5.8 -2.1

> -12. -13. -12. -11. -9.4 -5.4 -7.2

Near horizon gain Roll-off (dB)

> -14. -9.4 -8.6 -7.4 -7.3 -6.1 -6.0





3.0 6.2 3.6 4.5 6.6 35.8 11.0

> 3.0 1.8 2.8 2.3 2.2 4.4 6.8

> 3.0 1.2 1.6 1.1 1.0 2.5 5.6

RMS phase error (deg)

> 2.6 6.8 3.5 3.3 5.2 6.5 69.9

> > 2.6 1.6 2.4 1.5 1.3 2.8 4.7

2.6 1.5 2.4 1.5 1.1 2.9 12.6

RMS phase error (deg)

> 1.2 1.0 3.8 2.0 3.9 2.7 2.1

1.1 1.7 4.7 2.6 3.8 1.6 1.4

2.0 9.1 5.5 3.3 4.6 11.2 80.1

> 2.0 3.9 4.0 2.0 1.5 1.9 1.5

2.0 2.5 3.4 1.9 1.3 2.0 10.3

> In this chapter, we have presented numerical simulations and experimental measurements to analyze downward and upward drooped microstrip antennas with the intent of modifying the radiation pattern of the basic planar patch to accommodate the coverage requirements of GPS marine navigation and positioning. Magnitude and phase of the simulated and measured far-field radiation patterns are presented to reveal the tradeoffs in performance between patch geometry, ground plane size, and orientation. Results reported for the wide range of structural variations applied to the base antenna along with changes in the substrate material should be valuable to designers seeking to achieve a specific coverage performance.

> It has been found that an accurate and stable phase center can be obtained over the entire hemisphere for moderate upward bends. Numerical simulations and measurements demon‐ strate that the 3-dB beamwidth of the flat microstrip patch can be increased by at least 15% and 60% for the downward and upward bends, respectively. The phase stability demonstrated by the slightly bent structures may be viewed as advantageous in cases where circumstances require distorting the element but where a significant alteration in the pattern is not desired. These were accomplished, however, at the expense of some loss of the low-profile character of the antenna.

Although not dramatically affected, the cross polarization discrimination of the bent antennas is reduced by 3 dB at the horizon compared to the equivalent flat case. Calculations of the RMS error in the spherical phase fit over the upper hemisphere showed little change with bend angle or position. In general, there is a tradeoff in achieving broad-beam pattern coverage, and maintaining high cross polarization discrimination.

[5] Ying, Z, & Kildal, P. S. Improvement of Dipole, Helix, Spiral, Microstrip Patch and Aperture Antennas with Ground Planes by Using Corrugated Soft Surfaces. *IEE Proc.*

Drooped Microstrip Antennas for GPS Marine and Aerospace Navigation

http://dx.doi.org/10.5772/55002

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[6] Pozar, D. M, Duffy, S. M, & Dual-band, A. Circularly Polarized Aperture-Coupled Stacked Microstrip Antenna for Global Positioning Satellite. *IEEE Trans. Antennas*

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[9] Lee, S, Woo, J, Ryu, M, & Shin, H. Corrugated Circular Microstrip Patch Antennas

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The crossed dipole source, when used with the pedestal ground plane, demonstrated signifi‐ cant pattern improvements which inspired our interest in the drooped microstrip structure. It should be noted, however, that the crossed dipole is fundamentally different from the microstrip antenna, being itself a stand-alone radiator which operates in the presence of secondary sources created by the ground plane image. For the microstrip antenna, the ground plane is an integral part of the structure. The interior field distribution of the fundamental mode remains essentially unchanged with ground plane manipulation, and pattern modifi‐ cation can come about only by repositioning of the radiating edges in space.

### **Author details**

Ken G. Clark1 , Hussain M. Al-Rizzo2\*, James M. Tranquilla1 , Haider Khaleel3 and Ayman Abbosh2

\*Address all correspondence to: hmalrizzo@ualr.edu

1 EMR Microwave Technology Corporation, 64 Alison Blvd., Fredericton, NB, Canada

2 Systems Engineering Department, Donaghey College of Engineering and Information Technology, University of Arkansas at Little Rock, USA

3 Department of Engineering Science, Sonoma State University, Rohnert Park, CA, USA

#### **References**


[5] Ying, Z, & Kildal, P. S. Improvement of Dipole, Helix, Spiral, Microstrip Patch and Aperture Antennas with Ground Planes by Using Corrugated Soft Surfaces. *IEE Proc. Microw. Antennas Propagat*., Jun. (1996). , 143(3), 244-248.

Although not dramatically affected, the cross polarization discrimination of the bent antennas is reduced by 3 dB at the horizon compared to the equivalent flat case. Calculations of the RMS error in the spherical phase fit over the upper hemisphere showed little change with bend angle or position. In general, there is a tradeoff in achieving broad-beam pattern coverage, and

The crossed dipole source, when used with the pedestal ground plane, demonstrated signifi‐ cant pattern improvements which inspired our interest in the drooped microstrip structure. It should be noted, however, that the crossed dipole is fundamentally different from the microstrip antenna, being itself a stand-alone radiator which operates in the presence of secondary sources created by the ground plane image. For the microstrip antenna, the ground plane is an integral part of the structure. The interior field distribution of the fundamental mode remains essentially unchanged with ground plane manipulation, and pattern modifi‐

, Haider Khaleel3

and

cation can come about only by repositioning of the radiating edges in space.

, Hussain M. Al-Rizzo2\*, James M. Tranquilla1

1 EMR Microwave Technology Corporation, 64 Alison Blvd., Fredericton, NB, Canada

3 Department of Engineering Science, Sonoma State University, Rohnert Park, CA, USA

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2 Systems Engineering Department, Donaghey College of Engineering and Information

maintaining high cross polarization discrimination.

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\*Address all correspondence to: hmalrizzo@ualr.edu

Technology, University of Arkansas at Little Rock, USA

**Author details**

Ken G. Clark1

**References**

Ayman Abbosh2


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220-223.


**Chapter 13**

**Wearable Antennas for Medical Applications**

Microstrip antennas are widely employed in communication system and seekers. Microstrip antennas posse's attractive features such as low profile, flexible, light weight, small volume and low production cost. In addition, the benefit of a compact low cost feed network is attained by integrating the RF frontend with the radiating elements on the same substrate. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, the effect of human body on the electrical performance of wearable antennas at 434 MHz is not presented [8-13]. RF transmission properties of human tissues have been investigated in several articles [8-9]. Several wearable antennas have been presented in the last decade [10-14]. A review of wearable and body mounted antennas designed and developed for various applications at different frequency bands over the last decade can be found in [10]. In [11] meander wearable antennas in close proximity of a human body are presented in the frequency range between 800 MHz and 2700 MHz. In [12] a textile antenna performance in the vicinity of the human body is presented at 2.4 GHz. In [13] the effect of human body on wearable 100 MHz portable radio antennas is studied. In [13] the authors concluded that wearable antennas need to be shorter by 15% to 25% from the antenna length in free-space. Measurement of the antenna gain in [13] shows that a wide dipole (116 x 10 cm) has -13dBi gain. The antennas presented in [10-13] were developed mostly for cellular applications. Requirements and the frequency range

for medical applications are different from those for cellular applications

In this chapter, a new class of wideband compact wearable microstrip antennas for medical applications is presented. Numerical results with and without the presence of the human body are discussed. The antennas VSWR is better than 2:1at 434 MHz + 5%. The antenna beam width is around 100º. The antennas gain is around 0 to 4 dBi. The antenna resonant frequency is shifted by 5% if the air spacing between the antenna and the human body is increased from 0

> © 2013 Sabban; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use,

© 2013 Sabban; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

distribution, and reproduction in any medium, provided the original work is properly cited.

Additional information is available at the end of the chapter

Albert Sabban

**1. Introduction**

mm to 5 mm.

http://dx.doi.org/10.5772/54663

### **Wearable Antennas for Medical Applications**

### Albert Sabban

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/54663

### **1. Introduction**

Microstrip antennas are widely employed in communication system and seekers. Microstrip antennas posse's attractive features such as low profile, flexible, light weight, small volume and low production cost. In addition, the benefit of a compact low cost feed network is attained by integrating the RF frontend with the radiating elements on the same substrate. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, the effect of human body on the electrical performance of wearable antennas at 434 MHz is not presented [8-13]. RF transmission properties of human tissues have been investigated in several articles [8-9]. Several wearable antennas have been presented in the last decade [10-14]. A review of wearable and body mounted antennas designed and developed for various applications at different frequency bands over the last decade can be found in [10]. In [11] meander wearable antennas in close proximity of a human body are presented in the frequency range between 800 MHz and 2700 MHz. In [12] a textile antenna performance in the vicinity of the human body is presented at 2.4 GHz. In [13] the effect of human body on wearable 100 MHz portable radio antennas is studied. In [13] the authors concluded that wearable antennas need to be shorter by 15% to 25% from the antenna length in free-space. Measurement of the antenna gain in [13] shows that a wide dipole (116 x 10 cm) has -13dBi gain. The antennas presented in [10-13] were developed mostly for cellular applications. Requirements and the frequency range for medical applications are different from those for cellular applications

In this chapter, a new class of wideband compact wearable microstrip antennas for medical applications is presented. Numerical results with and without the presence of the human body are discussed. The antennas VSWR is better than 2:1at 434 MHz + 5%. The antenna beam width is around 100º. The antennas gain is around 0 to 4 dBi. The antenna resonant frequency is shifted by 5% if the air spacing between the antenna and the human body is increased from 0 mm to 5 mm.

© 2013 Sabban; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Sabban; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

#### **2. Dually polarized 434 MHz printed antenna**

A new compact microstrip loaded dipole antennas has been designed to provide horizontal polarization. The antenna dimensions have been optimized to operate on the human body by employing Agilent Advanced Design System (ADS) software [16]. The antenna consists of two layers. The first layer consists of RO3035 0.8 mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8 mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, thinner antennas are flexible. Thicker antennas have been designed with wider bandwidth. The printed slot antenna provides a vertical polarization. In several medical systems the required polarization may be vertical or horizontal. The proposed antenna is dually polarized. The printed dipole and the slot antenna provide dual orthogonal polariza‐ tions. The dimensions of the dual polarized antenna presented in Figure 1are 26 x 6 x 0.16 cm. The antenna may be used as a wearable antenna on a human body. The antenna may be attached to the patient shirt, patient stomach, or in the back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 2 dBi. The computed S11 and S22 parameters are presented in Figure 2. Figure 3 presents the antenna measured S11 parame‐ ters. The computed radiation patterns are shown in Figure 4. The co-polar radiation pattern belongs to the yz plane. The cross-polar radiation pattern belongs to the xz plane. The antenna cross polarized field strength may be adjusted by varying the slot feed location. The dimen‐ sions of the folded dually polarized antenna presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

Figure 1: Printed dually polarized antenna, 26 x 6 x 0.16 cm.

presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been

2

computed S11. The computed radiation pattern is shown in Fig 10.

A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and computed S11. The computed radiation pattern is

2

computed S11. The computed radiation pattern is shown in Fig 10.

The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The

Figure 1: Printed dually polarized antenna, 26 x 6 x 0.16 cm.

Figure 1: Printed dually polarized antenna, 26 x 6 x 0.16 cm.

Slot feed

Slot feed

Figure 2: Computed S11 and S22 results

Figure 2: Computed S11 and S22 results

S11 & S22

S11 & S22

Dipole

Dipole

Coupling stubs

Coupling stubs

Slot

 Dipole Feed

 Dipole Feed

Slot

Figure 3: Measured S11 on human body

Figure 3: Measured S11 on human body

3. New Loop Antenna with Ground Plane

3. New Loop Antenna with Ground Plane

measured results were compared to a known reference antenna.

**3. New loop antenna with ground plane**

results were compared to a known reference antenna.

results were compared to a known reference antenna.

**Figure 3.** Measured S11 on human body

shown in Fig 10.

**Figure 2.** Computed S11 and S22 results

S11

S11

presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

21cm

21cm

4cm

4cm

307

http://dx.doi.org/10.5772/54663

Wearable Antennas for Medical Applications

The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured

The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured

 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and

 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and

The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured

 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and

**Figure 1.** Printed dually polarized antenna, 26 x 6 x 0.16 cm.

Figure 2: Computed S11 and S22 results

S11 & S22

Figure 3: Measured S11 on human body

3. New Loop Antenna with Ground Plane

results were compared to a known reference antenna.

S11

2

computed S11. The computed radiation pattern is shown in Fig 10.

4cm

presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

21cm

21cm

Slot feed

Figure 1: Printed dually polarized antenna, 26 x 6 x 0.16 cm.

**Figure 2.** Computed S11 and S22 results S11 & S22

Slot

 Dipole Feed

Dipole

Slot

Figure 2: Computed S11 and S22 results

Figure 2: Computed S11 and S22 results

Dipole

Coupling stubs

Coupling stubs

**2. Dually polarized 434 MHz printed antenna**

306 Advancement in Microstrip Antennas with Recent Applications

human body electrical characteristics.

Slot

 Dipole Feed

2

computed S11. The computed radiation pattern is shown in Fig 10.

Figure 1: Printed dually polarized antenna, 26 x 6 x 0.16 cm.

Slot feed

Figure 2: Computed S11 and S22 results

S11 & S22

Dipole

**Figure 1.** Printed dually polarized antenna, 26 x 6 x 0.16 cm.

Coupling stubs

Figure 3: Measured S11 on human body

3. New Loop Antenna with Ground Plane

results were compared to a known reference antenna.

S11

A new compact microstrip loaded dipole antennas has been designed to provide horizontal polarization. The antenna dimensions have been optimized to operate on the human body by employing Agilent Advanced Design System (ADS) software [16]. The antenna consists of two layers. The first layer consists of RO3035 0.8 mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8 mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, thinner antennas are flexible. Thicker antennas have been designed with wider bandwidth. The printed slot antenna provides a vertical polarization. In several medical systems the required polarization may be vertical or horizontal. The proposed antenna is dually polarized. The printed dipole and the slot antenna provide dual orthogonal polariza‐ tions. The dimensions of the dual polarized antenna presented in Figure 1are 26 x 6 x 0.16 cm. The antenna may be used as a wearable antenna on a human body. The antenna may be attached to the patient shirt, patient stomach, or in the back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 2 dBi. The computed S11 and S22 parameters are presented in Figure 2. Figure 3 presents the antenna measured S11 parame‐ ters. The computed radiation patterns are shown in Figure 4. The co-polar radiation pattern belongs to the yz plane. The cross-polar radiation pattern belongs to the xz plane. The antenna cross polarized field strength may be adjusted by varying the slot feed location. The dimen‐ sions of the folded dually polarized antenna presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the

> presented in Figure 5 are 7 x 5 x 0.16 cm. Figure 6 presents the antenna computed S11 and S22 parameters. The computed radiation patterns of the folded dipole are shown in Figure 7. The antennas radiation characteristics on human body have been measured by using a phantom. The phantom electrical characteristics represent the human body electrical characteristics.

> > 21cm

4cm

The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured

 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and

characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured Figure 3: Measured S11 on human body **Figure 3.** Measured S11 on human body

3. New Loop Antenna with Ground Plane A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured results were compared to a known reference antenna. 3. New Loop Antenna with Ground Plane The phantom has a cylindrical shape with a 40cm diameter and a length of 1.5m. The phantom contains a mix of 55% water 44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation characteristics. S11 and S12 parameters were measured directly on human body by using a network analyzer. The measured results were compared to a known reference antenna.

44% sugar and 1% salt. The antenna under test was placed on the phantom during the measurements of the antennas radiation

 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop

#### antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and **3. New loop antenna with ground plane**

 2 A new loop antenna with ground plane has been designed on Kapton substrates with thickness of 0.25mm and 0.4mm. The antenna without ground plane is shown in Figure 8. The loop antenna VSWR without the tuning capacitor was 4:1. This loop antenna may be tuned by adding a capacitor or varactor as shown in Figure 8. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10.

2

computed S11. The computed radiation pattern is shown in Fig 10.

computed S11. The computed radiation pattern is shown in Fig 10.

results were compared to a known reference antenna.

**Figure 4.** Antenna Radiation patterns

Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body. Linear Polarization -40 -30 -20 Mag. [dB]E\_cross

E\_co


0

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor, better than the loop antenna without ground plane. The computed 3D radiation pattern is shown in Fig 11. Figure 4: Antenna Radiation patterns Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. -100 -50 0 50 100 -50 THETA E

Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body.

better than the loop antenna without ground plane. The computed 3D radiation pattern is shown in Fig 11.

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor,

<sup>θ</sup><sup>r</sup>

<sup>x</sup> <sup>φ</sup>

z

y

3

Beam width 3dB Gain dBi VSWR

Linear Polarization

E\_cross


THETA

The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig 11. Computed S11 of the Loop

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the antenna shown in Figure 1) for different belt thickness, shirt thickness and air spacing between the antennas and human body.

Loop no GND 100° 0 4:1 Loop with GND 100° 0 2:1

Figure 6: Folded antenna Computed S11 and S22 results

4

Figure 8: Tunable loop antenna without ground plane

Antenna with a tuning capacitor is given in Figure 12.

**4. Antenna S11 Variation as Function of Distance from Body** 

S11**&** S22

Figure 7: Folded antenna Radiation patterns

Antenna with no tuning





Mag. [dB]

E


0

E\_co

Table 1. Comparison of Loop Antennas

**20pF Capacitor/ Varactor**

**50mm** 

capacitor

**Figure 7.** Folded antenna Radiation patterns

**Figure 6.** Folded antenna Computed S11 and S22 results

 

**Feed lines** 

Figure 5: Folded dual polarized antenna, 7x5x0.16cm.

5.5cm

Figure 4: Antenna Radiation patterns


Mag. [dB]

0

E\_co


E

Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body.

<sup>θ</sup><sup>r</sup>

<sup>x</sup> <sup>φ</sup>

z

y

309

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor,

Wearable Antennas for Medical Applications

http://dx.doi.org/10.5772/54663

better than the loop antenna without ground plane. The computed 3D radiation pattern is shown in Fig 11.

4cm

Linear Polarization

E\_cross


THETA

Figure 5: Folded dual polarized antenna, 7x5x0.16cm.

3

Figure 6: Folded antenna Computed S11 and S22 results

S11**&** S22

**Figure 5.** Folded dual polarized antenna, 7x5x0.16cm.

<sup>θ</sup><sup>r</sup>

<sup>x</sup> <sup>φ</sup>

z

y

Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body.

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor,

better than the loop antenna without ground plane. The computed 3D radiation pattern is shown in Fig 11.

Linear Polarization

E\_cross


THETA

Figure 5: Folded dual polarized antenna, 7x5x0.16cm.

5.5cm

Figure 4: Antenna Radiation patterns


Mag. [dB]

0

E\_co


E

**Figure 6.** Folded antenna Computed S11 and S22 results

**Figure 4.** Antenna Radiation patterns

308 Advancement in Microstrip Antennas with Recent Applications

Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. Tuning the antenna allow us to work in a wider bandwidth.

E\_co

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor, better than the loop antenna without ground plane.

> Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. Tuning the antenna allow us to work in a wider bandwidth. Figure 9 presents the loop antenna computed S11on human body.

<sup>θ</sup><sup>r</sup>

<sup>x</sup> <sup>φ</sup>

z

y

There is good agreement between measured and computed S11. The computed radiation pattern is shown in Fig 10. Table I compares the electrical performance of a loop antenna with ground plane with a loop antenna without ground plane. There is a good agreement between measured and computed results of antenna parameters on human body. The results presented in Table I indicates that the loop antenna with ground plane is matched to the human body environment, without the tuning capacitor,

better than the loop antenna without ground plane. The computed 3D radiation pattern is shown in Fig 11.

4cm

Linear Polarization

E\_cross


THETA

Figure 4: Antenna Radiation patterns

3

Figure 6: Folded antenna Computed S11 and S22 results

S11**&** S22

**Figure 5.** Folded dual polarized antenna, 7x5x0.16cm.

Figure 5: Folded dual polarized antenna, 7x5x0.16cm.

5.5cm

Figure 9 presents the loop antenna computed S11on human body.

E

Mag. [dB]


0


The computed 3D radiation pattern is shown in Fig 11.

Loop no GND 100° 0 4:1 Loop with GND 100° 0 2:1

Beam width 3dB Gain dBi VSWR

The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig 11. Computed S11 of the Loop

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the antenna shown in Figure 1) for different belt thickness, shirt thickness and air spacing between the antennas and human body.

3

Figure 6: Folded antenna Computed S11 and S22 results

Antenna with no tuning **Figure 7.** Folded antenna Radiation patterns

capacitor

Figure 7: Folded antenna Radiation patterns

Table 1. Comparison of Loop Antennas

**20pF Capacitor/ Varactor**

**50mm** 

 

**Feed lines** 

4

Figure 8: Tunable loop antenna without ground plane

Antenna with a tuning capacitor is given in Figure 12.

**4. Antenna S11 Variation as Function of Distance from Body** 




0

E\_co

Beam width 3dB Gain dBi VSWR

Linear Polarization

E\_cross

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The

thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body

5

5

air spacing up to 5mm between the antennas and patient body.

5

air spacing up to 5mm between the antennas and patient body.

air spacing up to 5mm between the antennas and patient body.

Figure 11: New Loop Antenna 3D Radiation pattern

Figure 10: New Loop Antenna Radiation pattern on human body

Figure 10: New Loop Antenna Radiation pattern on human body

One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shown in Figure 5) has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for

<sup>x</sup> <sup>φ</sup>

One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shown in Figure 5) has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for

<sup>x</sup> <sup>φ</sup>

<sup>θ</sup><sup>r</sup>

z

z

z

<sup>θ</sup><sup>r</sup>

y

Wearable Antennas for Medical Applications

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311

<sup>x</sup> <sup>φ</sup>

<sup>θ</sup><sup>r</sup>

y

One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shown in Figure 5) has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for

y

Figure 11: New Loop Antenna 3D Radiation pattern

Figure 11: New Loop Antenna 3D Radiation pattern

Figure 10: New Loop Antenna Radiation pattern on human body

E

E

E

**Figure 10.** New Loop Antenna Radiation pattern on human body

S11

**Figure 9.** Computed S11 of new Loop Antenna

S11

S11

**Figure 11.** New Loop Antenna 3D Radiation pattern

Figure 9: Computed S11 of new Loop Antenna

Figure 9: Computed S11 of new Loop Antenna

Figure 9: Computed S11 of new Loop Antenna

Loop no GND 100° 0 4:1 Loop with GND 100° 0 2:1

Antenna with no tuning

Table 1. Comparison of Loop Antennas

capacitor

**Table 1.** Comparison of Loop Antennas Figure 7: Folded antenna Radiation patterns 

 The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig 11. Computed S11 of the Loop **Figure 8.** Tunable loop antenna without ground plane

Antenna with a tuning capacitor is given in Figure 12. **4. Antenna S11 Variation as Function of Distance from Body**  The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig 11. Computed S11 of the Loop Antenna with a tuning capacitor is given in Figure 12.

Figure 8: Tunable loop antenna without ground plane

#### dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt **4. Antenna S11 variation as function of distance from body**

 4 tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the antenna shown in Figure 1) for different belt thickness, shirt thickness and air spacing between the antennas and human body. The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the antenna shown in Figure 1) for different belt thickness, shirt thickness and air spacing between the antennas and human body.

Figure 9: Computed S11 of new Loop Antenna **Figure 9.** Computed S11 of new Loop Antenna S11

S11

E

Antenna with no tuning capacitor Beam width 3dB Gain dBi VSWR





Mag. [dB]

E


0

E\_co

Antenna with no tuning

Table 1. Comparison of Loop Antennas

**20pF Capacitor/ Varactor**

**50mm** 

capacitor

**Table 1.** Comparison of Loop Antennas

310 Advancement in Microstrip Antennas with Recent Applications

 

**Feed lines** 

**4. Antenna S11 variation as function of distance from body**

**Figure 8.** Tunable loop antenna without ground plane

air spacing between the antennas and human body.

Loop no GND 100° 0 4:1 Loop with GND 100° 0 2:1

Figure 7: Folded antenna Radiation patterns

Beam width 3dB Gain dBi VSWR

Linear Polarization

E\_cross


THETA

The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig 11. Computed S11 of the Loop

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt

antenna shown in Figure 1) for different belt thickness, shirt thickness and air spacing between the antennas and human body.

Loop no GND 100° 0 4:1 Loop with GND 100° 0 2:1

4

Figure 8: Tunable loop antenna without ground plane

Antenna with a tuning capacitor is given in Figure 12.

The computed 3D radiation pattern and the coordinate used in this chapter are shown in Fig

The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. The patient body thickness was varied from 15mm to 300mm. The dielectric constant of the body was varied from 40 to 50. The antenna was placed inside a belt with thickness between 2 to 4mm with dielectric constant from 2 to 4. The air layer between the belt and the patient shirt may vary from 0mm to 8mm. The shirt thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the antenna shown in Figure 1) for different belt thickness, shirt thickness and

11. Computed S11 of the Loop Antenna with a tuning capacitor is given in Figure 12.

**4. Antenna S11 Variation as Function of Distance from Body** 

Figure 9: Computed S11 of new Loop Antenna

thickness was varied from 0.5mm to 1mm. The dielectric constant of the shirt was varied from 2 to 4. Properties of human body tissues are listed in Table II see [8]. These properties were employed in the antenna design. Figure 15 presents S11 results (of the **Figure 10.** New Loop Antenna Radiation pattern on human body E

Figure 10: New Loop Antenna Radiation pattern on human body

5

air spacing up to 5mm between the antennas and patient body.

patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body.

One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shown in Figure 5) has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for

y

z

<sup>θ</sup><sup>r</sup>

y

between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than

patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for

5

5

air spacing up to 5mm between the antennas and patient body.

air spacing up to 5mm between the antennas and patient body.

Figure 10: New Loop Antenna Radiation pattern on human body

Explanation of Figure 16 is given in Table 3. If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for 2:1 when there is no air spacing between the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shown in Figure 5) has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and Figure 11: New Loop Antenna 3D Radiation pattern **Figure 11.** New Loop Antenna 3D Radiation pattern


**Table 2.** Properties of human body tissues

 

> 

 Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor **Figure 12.** Computed S11 of Loop Antenna, without ground plane, with a tuning capacitor **S11** 

Skin <sup>σ</sup> ε 41.6 40.43 Stomach <sup>σ</sup> 0.67 0.73 **Figure 13.** Radiation pattern of Loop Antenna without ground on human body

**Belt Sensor** 

Muscle

**ε=2-4**

**ε=2-4**

 **Ground** 

Lung <sup>σ</sup>

Table 2. Properties of human body tissues

**Shirt ε=2-4**

 **Ground** 

**Belt Sensor** 

**Air** 

**Shirt ε=2-4**

**Air** 

ε

ε

ε

Colon, Muscle σ ε 0.98 63.6 1.06 61.9 Lung <sup>σ</sup> ε 0.27 38.4 0.27 38.4 Table 2. Properties of human body tissues Skin <sup>σ</sup> ε 0.57 41.6 0.6 40.43 Stomach <sup>σ</sup> ε 0.67 42.9 0.73 41.41 Colon, σ 0.98 1.06 One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between

**0.5-0.8mm** 

63.6

0.27 38.4

**0.0-2mm** 

**0.5-0.8mm** 

**0.0-2mm** 

**3-4 mm** 

**3-4 mm** 

0.57

42.9

Tissue Property 434 MHz 600 MHz

0.6

Figure 13: Radiation pattern of Loop Antenna without ground on human body

the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shownin Figure 5) has V.S.W.Rbetterthan 2.0:1 for air spacingupto 5mmbetween the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacingbetweenthe sensors andthehumanbodyis increasedfrom0mmto5mmthe antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better

ε

ε

σ ε

ε

Tissue Property 434 MHz 600 MHz

0.57 41.6

0.67 42.9

0.98 63.6

0.27 38.4

**0.5-0.8mm** 

**0.0-2mm** 

**3-4 mm** 

Figure 13: Radiation pattern of Loop Antenna without ground on human body

0.6 40.43

0.73 41.41

1.06 61.9

0.27 38.4

Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor

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313

Wearable Antennas for Medical Applications

6

Figure 14: Analyzed structure for Impedance calculations

o **Free Space** 

**Air 0.0-5mm** 

**Body ε=40-50 15-300mm** 

than 2.0:1 for air spacing up to 5mm between the antennas and patient body.

Lung <sup>σ</sup>

Table 2. Properties of human body tissues

**Shirt ε=2-4**

**Air** 

Skin <sup>σ</sup>

Stomach <sup>σ</sup>

Colon, Muscle

**Belt Sensor** 

**ε=2-4**

 **Ground** 

  **S11** 

E

**Figure 15.** S11 results for different antenna positions relative to the human body

**Figure 14.** Analyzed structure for Impedance calculations

Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor

41.41

61.9

0.27 38.4

6

o **Free Space** 

Figure 14: Analyzed structure for Impedance calculations

6

Figure 14: Analyzed structure for Impedance calculations

**Body ε=40-50 15-300mm** 

o **Free Space** 

**Air 0.0-5mm** 

**Air 0.0-5mm** 

**Body ε=40-50 15-300mm** 

Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor

the antenna and the patient body. Results shown in Figure 16 indicates that the folded antenna (the antenna shownin Figure 5) has V.S.W.Rbetterthan 2.0:1 for air spacingupto 5mmbetween the antennas and patient body. Figure 16 presents S11 results of the folded antenna results for different position relative to the human body. Explanation of Figure 16 is given in Table 3. If the air spacingbetweenthe sensors andthehumanbodyis increasedfrom0mmto5mmthe antenna resonant frequency is shifted by 5%. The loop antenna with ground plane has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body. Figure 13: Radiation pattern of Loop Antenna without ground on human body Tissue Property 434 MHz 600 MHz Skin <sup>σ</sup> ε 0.57 41.6 0.6 40.43 Stomach <sup>σ</sup> ε 0.67 42.9 0.73 41.41 Colon, Muscle σ ε 0.98 63.6 1.06 61.9 Lung <sup>σ</sup> 0.27 0.27

38.4

38.4

ε

Table 2. Properties of human body tissues


Figure 14: Analyzed structure for Impedance calculations **Figure 14.** Analyzed structure for Impedance calculations

 

E

**S11** 

Tissue Property 434 MHz 600 MHz

0.57 41.6

0.67 42.9

0.98 63.6

0.27 38.4

Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor

Figure 12: Computed S11 of Loop Antenna, without ground plane, with A tuning capacitor

Figure 13: Radiation pattern of Loop Antenna without ground on human body

Figure 13: Radiation pattern of Loop Antenna without ground on human body

0.6 40.43

0.73 41.41

1.06 61.9

0.6 40.43

0.73 41.41

1.06 61.9

0.27 38.4

0.27 38.4

Tissue Property 434 MHz 600 MHz

0.57 41.6

0.67 42.9

Tissue Property 434 MHz 600 MHz

0.98 63.6

One may conclude from results shown in Figure 15 that the antenna has V.S.W.R better than 2.5:1 for air spacing up to 8mm between the antennas and patient body. For frequencies ranging from 415MHz to 445MHz the antenna has V.S.W.R better than 2:1 when there is no air spacing between

0.57 41.6

0.67 42.9

0.98 63.6

0.27 38.4

0.27 38.4

**0.5-0.8mm** 

**0.0-2mm** 

**0.5-0.8mm** 

**0.0-2mm** 

**3-4 mm** 

**3-4 mm** 

Skin <sup>σ</sup>

Stomach <sup>σ</sup>

**Figure 13.** Radiation pattern of Loop Antenna without ground on human body

Lung <sup>σ</sup>

Stomach <sup>σ</sup>

Skin <sup>σ</sup>

Table 2. Properties of human body tissues

Lung <sup>σ</sup>

Table 2. Properties of human body tissues

**Shirt ε=2-4**

 **Ground** 

**Belt Sensor** 

**Air** 

**Shirt ε=2-4**

**Air** 

Colon, Muscle

**Belt Sensor** 

Colon, Muscle

**ε=2-4**

**ε=2-4**

 **Ground** 

ε

ε

σ ε

ε

ε

σ ε

ε

ε

0.6 40.43

0.73 41.41

> 1.06 61.9

> 0.27 38.4

ε

ε

ε

ε

**Figure 12.** Computed S11 of Loop Antenna, without ground plane, with a tuning capacitor

6

o **Free Space** 

Figure 14: Analyzed structure for Impedance calculations

6

Figure 14: Analyzed structure for Impedance calculations

**Body ε=40-50 15-300mm** 

o **Free Space** 

**Air 0.0-5mm** 

**Air 0.0-5mm** 

**Body ε=40-50 15-300mm** 

Skin <sup>σ</sup>

312 Advancement in Microstrip Antennas with Recent Applications

Stomach <sup>σ</sup>

Colon, Muscle <sup>σ</sup>

 

> 

**S11** 

**S11** 

E

E

**Table 2.** Properties of human body tissues

Lung <sup>σ</sup>

**Figure 15.** S11 results for different antenna positions relative to the human body

If the air spacing between the sensors and the human body is increased from 0mm to 5mm the computed antenna resonant frequency is shifted by 2%. However, if the air spacing between the sensors and the human body is increased up to 5mm the measured loop antenna resonant frequency is shifted by 5%. Explanation of Figure 17 is given in Table 4.

powerful,butbecauseofrapidfall-offwithdistance,theantennadonotradiateenergytoinfinite distances,butinsteadtheradiatedpowerremaintrappedintheregionneartotheantenna.Thus, the near-fields only transfer energy to close distances from the receivers. The receiving and transmitting antennas are magnetically coupled. Change in current flow through one wire induces a voltage across the ends of the other wire through electromagnetic induction. The amount of inductive coupling between two conductors is measured by their mutual induc‐ tance. In these applications we have to refer to the near field and not to the far field radiation.

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Figure 16: Folded antenna S11 results for different antenna position relative to the human body

Figure 16: Folded antenna S11 results for different antenna position relative to the human body

Figure 17: Loop antenna S11 results for different antenna position relative to the human body

Head

8

Cables

Wearable Diversity antennas

**Figure 17.** Loop antenna S11 results for different antenna position relative to the human body

Wearable Diversity antennas

Cables

Belt

Belt

Head

Figure 17: Loop antenna S11 results for different antenna position relative to the human body

S11

S11

**Figure 16.** Folded antenna S11 results for different antenna position relative to the human body

recorder

 Loop or Printed Antenna

S11

recorder

Figure 18: Printed Wearable antenna

Figure 18: Printed Wearable antenna

8

 Loop or Printed Antenna

S11


**Table 3.** Explanation of Figure 16

#### **5. Wearable antenna**

An application of the proposed antenna is shown in Figure 18. Three to four folded dipole or loop antennas may be assembled in a belt and attached to the patient stomach. The cable from each antenna is connected to a recorder. The received signal is routed to a switching matrix. The signal with the highest level is selected during the medical test. The antennas receive a signal that is transmitted from various positions in the human body. Folded antennas may be also attached on the patient back in order to improve the level of the received signal from different locations in the human body. Figure 19 and Figure 20 show various antenna locations on the back and front of the human body for different medical applications.


**Table 4.** Explanation of Figure 17

transmittingandreceivingantennas is less than2D²/λ.Dis the largestdimensionofthe radiator. Intheseapplications theamplitudeofthe electromagnetic fieldclose totheantennamaybequite powerful,butbecauseofrapidfall-offwithdistance,theantennadonotradiateenergytoinfinite distances,butinsteadtheradiatedpowerremaintrappedintheregionneartotheantenna.Thus, the near-fields only transfer energy to close distances from the receivers. The receiving and transmitting antennas are magnetically coupled. Change in current flow through one wire induces a voltage across the ends of the other wire through electromagnetic induction. The amount of inductive coupling between two conductors is measured by their mutual induc‐ tance. In these applications we have to refer to the near field and not to the far field radiation.

If the air spacing between the sensors and the human body is increased from 0mm to 5mm the computed antenna resonant frequency is shifted by 2%. However, if the air spacing between the sensors and the human body is increased up to 5mm the measured loop antenna resonant

An application of the proposed antenna is shown in Figure 18. Three to four folded dipole or loop antennas may be assembled in a belt and attached to the patient stomach. The cable from each antenna is connected to a recorder. The received signal is routed to a switching matrix. The signal with the highest level is selected during the medical test. The antennas receive a signal that is transmitted from various positions in the human body. Folded antennas may be also attached on the patient back in order to improve the level of the received signal from different locations in the human body. Figure 19 and Figure 20 show various antenna locations

transmittingandreceivingantennas is less than2D²/λ.Dis the largestdimensionofthe radiator. Intheseapplications theamplitudeofthe electromagnetic fieldclose totheantennamaybequite

on the back and front of the human body for different medical applications.

frequency is shifted by 5%. Explanation of Figure 17 is given in Table 4.

Plot colure Sensor position

**Table 3.** Explanation of Figure 16

**5. Wearable antenna**

Plot colure Sensor position

**Table 4.** Explanation of Figure 17

Red Body 15mm air spacing 0mm Blue Air spacing 5mm Body 15mm Pink Body 40mm air spacing 0mm Green Body 30mm air spacing 0mm Sky Body 15mm Air spacing 2mm Purple Body 15mm Air spacing 4mm

Red Shirt thickness 0.5mm Blue Shirt thickness 1mm Pink Air spacing 2mm Green Air spacing 4mm Sky Air spacing 1mm Purple Air spacing 5mm

314 Advancement in Microstrip Antennas with Recent Applications

Figure 16: Folded antenna S11 results for different antenna position relative to the human body

Figure 16: Folded antenna S11 results for different antenna position relative to the human body

**Figure 16.** Folded antenna S11 results for different antenna position relative to the human body S11

 Loop or Printed Head **Figure 17.** Loop antenna S11 results for different antenna position relative to the human body

Figure 18: Printed Wearable antenna

8

recorder

 Loop or Printed Antenna

recorder

Figure 18: Printed Wearable antenna

Antenna

8

Cables

Wearable Diversity antennas

Wearable Diversity antennas

Cables

Belt

Belt

Head

Figure 17: Loop antenna S11 results for different antenna position relative to the human body

**Figure 18.** Printed Wearable antenna

Figure 19: Printed Patch Antenna locations for various medical applications

In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm. The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm.

GND

In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm.

In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm. The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm.

GND

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of FR4 0.25mm dielectric substrate. The second layer consists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is

The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in

 In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the

Slot Dipole

Feed Lines

A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consists

shown in Figure 22. The antenna dimensions are 5x5x0.05cm.

Figure 21. Microstrip Antennas for medical Applications

5cm

Figure 25. The computed radiation pattern is shown in Figure 26.

10

Figure 22. Printed Compact dual polarized antenna

antenna resonant frequency is shifted by 5%.

**7. Helix Antenna Performance on Human Body**

**Figure 20.** a. Microstrip Antennas for medical Applications b. Backside of the antennas6. Compact dual polarized

A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consistsof FR4 0.25mm dielectric substrate. The second layer con‐ sists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is shown in Figure 22. The antenna dimensions are

The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm.

Figure 19: Printed Patch Antenna locations for various medical applications

Antennas Antennas

9

  5cm

**6. Compact Dual Polarized Printed Antenna**

printed antenna

5x5x0.05cm.

Figure 20. a. Microstrip Antennas for medical Applications b. Backside of the antennas

Folded Dual polarized Antenna

Loop Antenna With GND

**Figure 21.** Microstrip Antennas for medical Applications

Loop Antenna With GND

A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consists

9

**6. Compact Dual Polarized Printed Antenna**

Figure 20. a. Microstrip Antennas for medical Applications b. Backside of the antennas

Loop Antenna With GND

**Figure 19.** Printed Patch Antenna locations for various medical applications

The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in

 In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the

Slot Dipole

Feed Lines

In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm. The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm. Figure 19: Printed Patch Antenna locations for various medical applications In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is Antennas Antennas

presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm. The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm.

**6. Compact Dual Polarized Printed Antenna** A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consists **Figure 20.** a. Microstrip Antennas for medical Applications b. Backside of the antennas6. Compact dual polarized printed antenna

 9 A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consistsof FR4 0.25mm dielectric substrate. The second layer con‐ sists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is shown in Figure 22. The antenna dimensions are 5x5x0.05cm. of FR4 0.25mm dielectric substrate. The second layer consists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is

shown in Figure 22. The antenna dimensions are 5x5x0.05cm.

Figure 25. The computed radiation pattern is shown in Figure 26.

5cm

10

Figure 22. Printed Compact dual polarized antenna

antenna resonant frequency is shifted by 5%.

**7. Helix Antenna Performance on Human Body**

Figure 21. Microstrip Antennas for medical Applications **Figure 21.** Microstrip Antennas for medical Applications

  5cm

**Figure 18.** Printed Wearable antenna

316 Advancement in Microstrip Antennas with Recent Applications

9

**6. Compact Dual Polarized Printed Antenna**

Figure 20. a. Microstrip Antennas for medical Applications b. Backside of the antennas

Loop Antenna With GND

Figure 19: Printed Patch Antenna locations for various medical applications

Antennas Antennas

**Figure 19.** Printed Patch Antenna locations for various medical applications

In Figure 20 and 21 several microstrip antennas for medical applications at 434MHz are shown. The Backside of the antennas is presented in Figure 20.b. The diameter of the loop antenna presented in Figure 21 is 50 mm. The dimensions of the folded dipole antenna are 7x6x0.16cm. The dimensions of the compact folded dipole presented in Figure 21 are 5x5x0.5cm.

GND

A new compact microstrip loaded dipole antennas has been designed. The antenna consists of two layers. The first layer consists

Folded Dual polarized Antenna

The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in Figure 25. The computed radiation pattern is shown in Figure 26. Figure 21. Microstrip Antennas for medical Applications The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around Loop Antenna With GND

Figure 25. The computed radiation pattern is shown in Figure 26.

shown in Figure 22. The antenna dimensions are 5x5x0.05cm.

of FR4 0.25mm dielectric substrate. The second layer consists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is

10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in

 In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper

thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna

11

11

0

THETA

0

20

20

40

40

60

60

80

80

100

100

Figure 26. Antenna Radiation pattern

0THETA

Figure 26. Antenna Radiation pattern

Figure 23. Computed S11 results of compact antenna

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Figure 23. Computed S11 results of compact antenna

Figure 24. Measured S11 on human body

Figure 24. Measured S11 on human body

Figure 25. Compact Antenna 3D Radiation pattern

Figure 25. Compact Antenna 3D Radiation pattern

E\_co E\_cross

E\_co E\_cross

Linear Polarization

Linear Polarization

S11

S11

**Figure 23.** Computed S11 results of compact antenna

**Figure 24.** Measured S11 on human body

S11

S11




Mag. [dB]

Mag. [dB]

0


**E**

**Figure 25.** Compact Antenna 3D Radiation pattern

**E**


Figure 22. Printed Compact dual polarized antenna

**7. Helix Antenna Performance on Human Body Figure 22.** Printed Compact dual polarized antenna

 

#### under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm **6. Helix antenna performance on human body**

 10 dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the antenna resonant frequency is shifted by 5%. In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the antenna resonant frequency is shifted by 5%.

Figure 23. Computed S11 results of compact antenna

Figure 23. Computed S11 results of compact antenna

 **Figure 23.** Computed S11 results of compact antenna S11

The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in Figure 25. The

Figure 21. Microstrip Antennas for medical Applications

5cm

Figure 25. The computed radiation pattern is shown in Figure 26.

shown in Figure 22. The antenna dimensions are 5x5x0.05cm.

of FR4 0.25mm dielectric substrate. The second layer consists of Kapton 0.25mm dielectric substrate. The substrate thickness determines the antenna bandwidth. However, with thinner substrate we may achieve better flexibility. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized antenna is

The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 0dBi. The computed S11 parameters are presented in Figure 23. Figure 24 presents the antenna measured S11 parameters. The antenna cross-polarized field strength may be adjusted by varying the slot feed location. The computed 3D radiation pattern of the antenna is shown in

> Slot Dipole

Feed Lines

a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the

10

In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas a helix antenna has been designed. A helix antenna with 9 turns is shown in figure 27. The backside of the circuit is copper under the microstrip matching stubs. However, in the helix antenna area there is no ground plane. The antenna has been designed to operate on human body. A matching microstrip line network has been designed on RO4003 substrate with 0.8mm thickness. The helix antenna has VSWR better than 3:1 at the frequency range from 440MHz to 460MHz. The antenna dimensions are 4x4x0.6cm. Figure 28 presents the measured S11 parameters on human body. The computed E and H radiation plane of the helix antenna is shown in Figure 29. The helix antenna input impedance variation as function of distance from the body is very sensitive. If the air spacing between the helix antenna and the human body is increased from 0mm to 2mm the antenna resonant frequency

Figure 22. Printed Compact dual polarized antenna

antenna resonant frequency is shifted by 5%.

**7. Helix Antenna Performance on Human Body**

computed radiation pattern is shown in Figure 26.

318 Advancement in Microstrip Antennas with Recent Applications

5cm

Folded Dual polarized Antenna

Loop Antenna With GND

 

**6. Helix antenna performance on human body**

**Figure 22.** Printed Compact dual polarized antenna

is shifted by 5%.

 In order to compare the variation of the new antenna input impedance as function of distance from the body to other antennas Figure 24. Measured S11 on human body **Figure 24.** Measured S11 on human body

11

11

0

THETA

0

20

20

40

40

60

60

80

80

100

100

Figure 26. Antenna Radiation pattern

Figure 26. Antenna Radiation pattern





**E**


S11

S11

Figure 25. Compact Antenna 3D Radiation pattern

Figure 24. Measured S11 on human body

Figure 23. Computed S11 results of compact antenna

**Figure 26.** Antenna Radiation pattern

 11 However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted only by 5%. However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the

antenna resonant frequency is shifted only by 5%.

Figure 26. Antenna Radiation pattern

Figure 27. Helix Antenna for medical Applications

Figure 27. Helix Antenna for medical Applications

**Figure 27.** Helix Antenna for medical Applications

12

12

**8.1 Dually Polarized Tunable Printed Antenna** 

Figure29. E and H plane radiation pattern of The Helix antenna

z

**8.1 Dually Polarized Tunable Printed Antenna** 

Figure29. E and H plane radiation pattern of The Helix antenna

y

8. **Wearable Tunable Printed Antennas for Medical Applications**

z

8. **Wearable Tunable Printed Antennas for Medical Applications**

y

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body. 12

**8.1 Dually Polarized Tunable Printed Antenna** 

**7. Wearable tunable printed antennas for medical applications**

Figure29. E and H plane radiation pattern of The Helix antenna

z

8. **Wearable Tunable Printed Antennas for Medical Applications**

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm dielectric substrate. The substrate thickness affects the antenna band width. The printed slot antenna provides a vertical polarization. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dimensions of the dual polarized antenna are 26x6x0.16cm. Also tunable compact folded dual polarized antennas have been designed. The dimensions of the compact antennas are 5x5x0.05cm.Varactors are connected to the antenna feed lines as shown in Figure 30. The voltage controlled varactors are used to control the antenna resonant frequency. The varactor bias voltage may be varied automatically to set the antenna resonant frequency at different locations on the human body. The antenna may be used as a wearable antenna on a human body. The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agree‐ ment between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around

y

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

 A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm

However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the

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321

antenna resonant frequency is shifted only by 5%.

Helix **Tuning stubs Microsrip Line** 

Figure 27. Helix Antenna for medical Applications

Figure 28: Measured S11 on human body

**Figure 29.** E and H plane radiation pattern of The Helix antenna

**7.1. Dually polarized tunable printed antenna**

S11

2dBi.

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

 A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm

 A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm

**Figure 28.** Measured S11 on human body

Figure 28: Measured S11 on human body

However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the

8. **Wearable Tunable Printed Antennas for Medical Applications**

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of

**Figure 29.** E and H plane radiation pattern of The Helix antenna

Figure 28: Measured S11 on human body

S11

antenna resonant frequency is shifted only by 5%.

Helix **Tuning stubs Microsrip Line** 

Figure 27. Helix Antenna for medical Applications

11

Helix **Tuning stubs Microsrip Line** 

12

12

**8.1 Dually Polarized Tunable Printed Antenna** 

Figure29. E and H plane radiation pattern of The Helix antenna

z

**8.1 Dually Polarized Tunable Printed Antenna** 

Figure29. E and H plane radiation pattern of The Helix antenna

y

8. **Wearable Tunable Printed Antennas for Medical Applications**

z

8. **Wearable Tunable Printed Antennas for Medical Applications**

y

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

 A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm

 A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm

However, if the air spacing between the new dual polarized antenna and the human body is

antenna resonant frequency is shifted only by 5%.

Figure 27. Helix Antenna for medical Applications

Figure 27. Helix Antenna for medical Applications

Figure 28: Measured S11 on human body

Figure 28: Measured S11 on human body

S11

**Figure 28.** Measured S11 on human body

**Figure 27.** Helix Antenna for medical Applications

S11

0

THETA

20

40

60

80

100

However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the

However, if the air spacing between the new dual polarized antenna and the human body is increased from 0mm to 5mm the

Figure 26. Antenna Radiation pattern

increased from 0mm to 5mm the antenna resonant frequency is shifted only by 5%.

antenna resonant frequency is shifted only by 5%.

Helix **Tuning stubs Microsrip Line** 

Figure 23. Computed S11 results of compact antenna

Figure 24. Measured S11 on human body

Figure 25. Compact Antenna 3D Radiation pattern

E\_co E\_cross

Linear Polarization

S11

S11

320 Advancement in Microstrip Antennas with Recent Applications

**Figure 26.** Antenna Radiation pattern



Mag. [dB]

0


**E**

#### **8.1 Dually Polarized Tunable Printed Antenna 7. Wearable tunable printed antennas for medical applications**

Figure29. E and H plane radiation pattern of The Helix antenna

 12 two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm A new class of wideband tunable wearable microstrip antennas for medical applications is presented in this section. The antennas VSWR is better than 2:1at 434MHz+5%. The antenna beam width is around 100º. The antennas gain is around 0 to 2dBi. A voltage controlled varactor is used to control the antenna resonant frequency at different locations on the human body.

#### **7.1. Dually polarized tunable printed antenna**

A compact tunable microstrip dipole antenna has been designed to provide horizontal polarization. The antenna consists of two layers. The first layer consists of RO3035 0.8mm dielectric substrate. The second layer consists of RT-Duroid 5880 0.8mm dielectric substrate. The substrate thickness affects the antenna band width. The printed slot antenna provides a vertical polarization. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dimensions of the dual polarized antenna are 26x6x0.16cm. Also tunable compact folded dual polarized antennas have been designed. The dimensions of the compact antennas are 5x5x0.05cm.Varactors are connected to the antenna feed lines as shown in Figure 30. The voltage controlled varactors are used to control the antenna resonant frequency. The varactor bias voltage may be varied automatically to set the antenna resonant frequency at different locations on the human body. The antenna may be used as a wearable antenna on a human body. The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agree‐ ment between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 2dBi.

dielectric substrate. The substrate thickness affects the antenna band width. The printed slot antenna provides a vertical polarization. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dimensions of the dual polarized antenna are 26x6x0.16cm. Also tunable compact folded dual polarized antennas have been designed. The dimensions of the compact antennas are 5x5x0.05cm.Varactors are connected to the antenna feed lines as shown in Figure 30. The voltage controlled varactors are used to control the antenna resonant frequency. The varactor bias voltage may be varied automatically

on a human body. The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 2dBi.

> Figure 31 presents the antenna measured S11 parameters without a varactor. Figure 32 presents the antenna S11 parameters as function of different varactor capacitances. Figure 33 presents the tunable antenna resonant frequency as function of the varactor capacitance. The antenna resonant frequency varies around 5% for capacitances up to 2.5pF. The antenna beam width is 100º. The antenna cross polarized field strength may be adjusted by varying the slot feed

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Figure 31 presents the antenna measured S11 parameters without a varactor. Figure 32 presents the antenna S11 parameters as function of different varactor capacitances. Figure 33 presents the tunable antenna resonant frequency as function of the varactor capacitance. The antenna resonant frequency varies around 5% for capacitances up to 2.5pF. The antenna beam width is 100º. The

432 427 423 418 413

The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. Properties of human body tissues are listed in Table 2 see [8]. Figure 34 presents S11 results for different belt and shirt thickness, and air spacing between the antennas and human body.

**Resonant Frequency (MHz)**

location.

Figure 30.Dual polarized tunable antenna, 26x6x0.16 cm.

Figure 31.Measured S11 on human body

8

**Figure 34.** S11 results for different antenna positions

Figure 33.Resonant frequency as function of varactor capacitance

Figure 32.The Tunable S11 parameter as function of varactor capacitance

antenna cross polarized field strength may be adjusted by varying the slot feed location.

**7.2. Antenna S11 varitaion as function of distance from body** 

**7.2. Antenna S11 varitaion as function of distance from body**

**Figure 33.** Resonant frequency as function of varactor capacitance

0 0.5 1 1.5 2 2.5 3

**Capacitor (pF)**

Readout

Readout

Readout

 Figure 30. Dual polarized tunable antenna, 26x6x0.16 cm. **Figure 30.** Dual polarized tunable antenna, 26x6x0.16 cm.

**Figure 31.** Measured S11 on human body

**Figure 32.** The Tunable S11 parameter as function of varactor capacitance

Readout

Readout

Readout

bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antenna gain is around 2dBi. Figure 31 presents the antenna measured S11 parameters without a varactor. Figure 32 presents the antenna S11 parameters as function of different varactor capacitances. Figure 33 presents the tunable antenna resonant frequency as function of the varactor capacitance. The antenna resonant frequency varies around 5% for capacitances up to 2.5pF. The antenna beam width is 100º. The antenna cross polarized field strength may be adjusted by varying the slot feed location. Figure 32.The Tunable S11 parameter as function of varactor capacitance Figure 31 presents the antenna measured S11 parameters without a varactor. Figure 32 presents the antenna S11 parameters as function of different varactor capacitances. Figure 33 presents the tunable antenna resonant frequency as function of the varactor capacitance. The antenna resonant frequency varies around 5% for capacitances up to 2.5pF. The antenna beam width is 100º. The

Figure 33.Resonant frequency as function of varactor capacitance **Figure 33.** Resonant frequency as function of varactor capacitance

**7.2. Antenna S11 varitaion as function of distance from body** 

Figure 30.Dual polarized tunable antenna, 26x6x0.16 cm.

Figure 31.Measured S11 on human body

#### **7.2. Antenna S11 varitaion as function of distance from body**

antenna cross polarized field strength may be adjusted by varying the slot feed location.

 8 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. Properties of human body tissues are listed in Table 2 see [8]. Figure 34 presents S11 results for different belt and shirt thickness, and air spacing between the antennas and human body.

**Figure 34.** S11 results for different antenna positions

13

m2

m5

m3

2pF Capacitor

dB(dpant\_1h2pf \_shirt..S(1,1))=-40.434 418.0MHz

1.5pF Capacitor

m4

m1

m5 f req=

400 410 420 430 440 450 460 470 480 490 500

1pF Capacitor

2.5pF Capacitor

freq, MHz

Figure 32. The Tunable S11 parameter as function of varactor capacitance

dB(dpant\_1h1pf 5\_shirt..S(1,1))=-44.255 423.0MHz

> m4 f req=

m1 f req=

400 420 440 460 480 500

m1 freq=

freq, MHz

The antenna cross polarized field strength may be adjusted by varying the slot feed location.

dB(dpant\_1h2pf 5\_shirt..S(1,1))=-38.901 413.0MHz

dB(dpant\_1h\_shirt..S(1,1))=-27.153 432.0MHz m2 freq= dB(S(1,1))=-40.357 427.0MHz

Figure 31 presents the antenna measured S11 parameters without a varactor. Figure 32 presents the antenna S11 parameters as function of different varactor capacitances. Figure 33 presents the tunable antenna resonant frequency as function of the varactor capacitance. The antenna resonant frequency varies around 5% for capacitances up to 2.5pF. The antenna beam width is 100º.

Figure 30. Dual polarized tunable antenna, 26x6x0.16 cm.

Slot feed

m2

m2 freq=

dB(S(1,1))=-10.034 460.0MHz

dB(S(1,1))=-27.153 432.0MHz

Slot

Figure 31. Measured S11 on human body

dB(S(1,1))=-14.764 410.0MHz

m3

m3 freq=

m1

Dipole

Coupling stubs

 Dipole Feed

Varactors

0

**Figure 30.** Dual polarized tunable antenna, 26x6x0.16 cm.

322 Advancement in Microstrip Antennas with Recent Applications

**Figure 31.** Measured S11 on human body

dB(S(1,1))




dB(S(1,1))

dB(dpant\_1h\_shirt..S(1,1))

**Figure 32.** The Tunable S11 parameter as function of varactor capacitance

dB(dpant\_1h2pf\_shirt..S(1,1))

dB(dpant\_1h2pf5\_shirt..S(1,1))

dB(dpant\_1h1pf5\_shirt..S(1,1))

0

m3 f req=


dielectric substrate. The substrate thickness affects the antenna band width. The printed slot antenna provides a vertical polarization. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dimensions of the dual polarized antenna are 26x6x0.16cm. Also tunable compact folded dual polarized antennas have been designed. The dimensions of the compact antennas are 5x5x0.05cm.Varactors are connected to the antenna feed lines as shown in Figure 30. The voltage controlled varactors are used to control the antenna resonant frequency. The varactor bias voltage may be varied automatically to set the antenna resonant frequency at different locations on the human body**.** The antenna may be used as a wearable antenna on a human body. The antenna may be attached to the patient shirt in the patient stomach or back zone. The antenna has been analyzed by using Agilent ADS software. There is a good agreement between measured and computed results. The antenna

0 0.5 1 1.5 2 2.5 3

**Capacitor (pF)**

If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. There is good agreement between measured and calculated results. The voltage controlled varactor may be used to tune the antenna resonant frequency due to different antenna locations on a human body. Figure 35 presents several compacttunableAntennas formedicalApplications.Avoltage controlledvaractormaybeused also to tune the loop antenna resonant frequency at different antenna locations on the body. Figure 34. S11 results for different antenna positions If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. There is good agreement between measured and calculated results. The voltage controlled varactor may be used to tune the antenna resonant frequency due to different antenna locations on a human body. Figure 35 presents several compact

frequency at different antenna locations on the body.

Figure 33. Resonant frequency as function of varactor capacitance

8.2. ANTENNA S11 VARITAION AS FUNCTION OF DISTANCE FROM BODY

432 427 423 418 413

**Resonant Frequency (MHz)**

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. Properties of human body tissues are listed in Table 2 see [8]. Figure

tunable Antennas for medical Applications. A voltage controlled varactor may be used also to tune the loop antenna resonant

Figure 37 presents the C-V curves of varactors MA46H070 to MA46H074

Figure 37. C-V curves of varactors MA46H070 to MA46H076

Figure 38 presents a compact tunable antenna with a varactor. Figure 39. presents measured S11 as function of varactor bias voltage. We may conclude that varactors may be used to compen‐ sate variations in the antenna resonant frequency at different locations on the human body.

Figure 45. RFID printed antenna, 8.4x6.4x0.16cm.

.

Figure 38. Tunable antenna with a varactor

**Figure 38.** Tunable antenna with a varactor

Varactor

**Figure 37.** C-V curves of varactors MA46H070 to MA46H076

16

**RFID** (*R*adio *F*requency *Id*entification) is an electronic method of exchanging data over radio frequency waves. There are three major components in RFID system: Transponder (Tag), Antenna and a Controller. The RFID tag, antenna and controller may be

300 320 340 360 380 400 420 440 460 480 500

m4 m1 freq=

m4 freq=

7V

dB(S(1,1))=-29.253 384.0MHz

dB(FOLDED\_VARC\_7V..S(1,1))=-27.767 396.0MHz

m3 freq=

dB(FOLDED\_VARC\_XV..S(1,1))=-31.592 386.0MHz

m1

m3

m2

dB(FOLDED\_NOVARC..S(1,1))=-13.968

freq, MHz

Figure 39. Measured S11 as function of varactor bias voltage

**9. Compact Wearable RFID Antennas** 





dB(S(1,1))

dB(FOLDED\_NOVARC..S(1,1))

dB(FOLDED\_VARC\_XV..S(1,1))

dB(FOLDED\_VARC\_7V..S(1,1))


0

m2 freq=

375.0MHz

8V

9V NO VARACTOR 2

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Tuning varactors are voltage variable capacitors designed to provide electronic tuning of microwave components. Varactors are manufactured on silicon and gallium arsenide substrates. Gallium arsenide varactors offer higher Q and may be used at higher frequencies than silicon varactors. Hyperabrupt varactors provide nearly linear variation of frequency with applied control

1

34 presents S11 results for different belt and shirt thickness, and air spacing between the antennas and human body.

**9** 1 Place Table 2 in one page Place Table 2 in one page not in 2 pages

8.3. VARACTORS **Figure 35.** Tunable Antennas for medical Applications 

**Page No. Line** 

#### **7.3. Varactors**

 14 Tuning varactors are voltage variable capacitors designed to provide electronic tuning of microwave components. Varactors are manufactured on silicon and gallium arsenide sub‐ strates. Gallium arsenide varactors offer higher Q and may be used at higher frequencies than silicon varactors. Hyperabrupt varactors provide nearly linear variation of frequency with applied control voltage. However abrupt varactors provide inverse fourth root frequency dependence. MACOM offers several gallium arsenide hyperabrupt varactors such as MA46 series. Figure 36 presents the C-V curves of varactors MA46505 to MA46506. **25** 14 Figure 45 is not visible Correct Figure 45, attached new Figure 48 **27** 1 Figure 48 is not visible Correct Figure 48, attached new Figure 48 **29** 1 than 2D²/ λ. In one line than 2D²/λ. **30** 22 Author detail is not completed Author detail is not completed, attached new **<sup>30</sup>**24 References ( " "), not printed in all references up to the end Attached new references List **<sup>31</sup>**31 References ( " "), not printed in all references up

to the end Attached new references List

**20** 17 Figure 36 Attached new Figure 36 **21** 2 Figure 37 Attached new Figure 37

Figure 36. Varactor capacitance as function of bias voltage

**Figure 36.** Varactor capacitance as function of bias voltage

#### Figure 37 presents the C-V curves of varactors MA46H070 to MA46H074

 Figure 37. C-V curves of varactors MA46H070 to MA46H076 **Figure 37.** C-V curves of varactors MA46H070 to MA46H076

If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. There is good agreement between measured and calculated results. The voltage controlled varactor may be used to tune the antenna resonant frequency due to different antenna locations on a human body. Figure 35 presents several compacttunableAntennas formedicalApplications.Avoltage controlledvaractormaybeused also to tune the loop antenna resonant frequency at different antenna locations on the body.

Figure 34. S11 results for different antenna positions

frequency at different antenna locations on the body.

324 Advancement in Microstrip Antennas with Recent Applications

0 0.5 1 1.5 2 2.5 3

**Capacitor (pF)**

Figure 33. Resonant frequency as function of varactor capacitance

8.2. ANTENNA S11 VARITAION AS FUNCTION OF DISTANCE FROM BODY

432 427 423 418 413

**Resonant Frequency (MHz)**

 The Antennas input impedance variation as function of distance from the body had been computed by employing ADS software. The analyzed structure is presented in Figure 14. Properties of human body tissues are listed in Table 2 see [8]. Figure

If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. There is good agreement between measured and calculated results. The voltage controlled varactor may be used to tune the antenna resonant frequency due to different antenna locations on a human body. Figure 35 presents several compact tunable Antennas for medical Applications. A voltage controlled varactor may be used also to tune the loop antenna resonant

Tuning varactors are voltage variable capacitors designed to provide electronic tuning of microwave components. Varactors are manufactured on silicon and gallium arsenide substrates. Gallium arsenide varactors offer higher Q and may be used at higher frequencies than silicon varactors. Hyperabrupt varactors provide nearly linear variation of frequency with applied control

1

34 presents S11 results for different belt and shirt thickness, and air spacing between the antennas and human body.

14

**25** 14 Figure 45 is not visible Correct Figure 45, attached new Figure 48 **27** 1 Figure 48 is not visible Correct Figure 48, attached new Figure 48

**30** 22 Author detail is not completed Author detail is not completed, attached new

to the end Attached new references List

to the end Attached new references List

In one line than 2D²/λ.

Tuning varactors are voltage variable capacitors designed to provide electronic tuning of microwave components. Varactors are manufactured on silicon and gallium arsenide sub‐ strates. Gallium arsenide varactors offer higher Q and may be used at higher frequencies than silicon varactors. Hyperabrupt varactors provide nearly linear variation of frequency with applied control voltage. However abrupt varactors provide inverse fourth root frequency dependence. MACOM offers several gallium arsenide hyperabrupt varactors such as MA46

**20** 17 Figure 36 Attached new Figure 36 **21** 2 Figure 37 Attached new Figure 37

Figure 35. Tunable Antennas for medical Applications

**8** 12 , with A tuning capacitor , with a tuning capacitor

Loop Antenna With GND

**No. Delete Replace with** 

**9** 1 Place Table 2 in one page Place Table 2 in one page not in 2 pages

Varactors

8.3. VARACTORS

**29** 1 than 2D²/ λ.

**Figure 36.** Varactor capacitance as function of bias voltage

series. Figure 36 presents the C-V curves of varactors MA46505 to MA46506.

**<sup>30</sup>**24 References ( " "), not printed in all references up

**<sup>31</sup>**31 References ( " "), not printed in all references up

Figure 36. Varactor capacitance as function of bias voltage

**Figure 35.** Tunable Antennas for medical Applications

**Page No. Line** 

**7.3. Varactors**

. Figure 38 presents a compact tunable antenna with a varactor. Figure 39. presents measured S11 as function of varactor bias voltage. We may conclude that varactors may be used to compen‐ sate variations in the antenna resonant frequency at different locations on the human body.

**RFID** (*R*adio *F*requency *Id*entification) is an electronic method of exchanging data over radio frequency waves. There are three major components in RFID system: Transponder (Tag), Antenna and a Controller. The RFID tag, antenna and controller may be

300 320 340 360 380 400 420 440 460 480 500

m4 m1 freq=

m4 freq=

7V

dB(S(1,1))=-29.253 384.0MHz

dB(FOLDED\_VARC\_7V..S(1,1))=-27.767 396.0MHz

m3 freq=

dB(FOLDED\_VARC\_XV..S(1,1))=-31.592 386.0MHz

m1

m3

m2

dB(FOLDED\_NOVARC..S(1,1))=-13.968

freq, MHz

**Figure 38.** Tunable antenna with a varactor

16

Figure 39. Measured S11 as function of varactor bias voltage

**9. Compact Wearable RFID Antennas** 





dB(S(1,1))

dB(FOLDED\_NOVARC..S(1,1))

dB(FOLDED\_VARC\_XV..S(1,1))

dB(FOLDED\_VARC\_7V..S(1,1))


0

m2 freq=

375.0MHz

8V

9V NO VARACTOR

0.8mm dielectric substrate. The second layer consists of Kapton 0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The antenna may be attached to the customer shirt in the customer stomach or back zone. The antenna has been analyzed by using

are crucial in the development of RIFD systems.

are crucial in the development of RIFD systems.

Agilent ADS software.

Agilent ADS software.

Dipole

Dipole

**9.1 Dual Polarized 13.5MHz Compact Printed Antenna**

**9.1 Dual Polarized 13.5MHz Compact Printed Antenna**

more effective at penetrating non-metal objects especially objects with high water content.

animals. The proposed antennas may be attached to cars, trucks, containers and other various objects.

animals. The proposed antennas may be attached to cars, trucks, containers and other various objects.

more effective at penetrating non-metal objects especially objects with high water content.

Slot

**RFID System** 

Slot

**RFID System** 

assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are

RF transmission properties of human tissues have been investigated in several papers [8-9]. The effect of human body on the antenna performance is investigated in this chapter. The proposed antennas may be used as wearable antennas on persons or

A new class of wideband compact printed and microstrip antennas for RFID applications is presented in this chapter.

assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are

RF transmission properties of human tissues have been investigated in several papers [8-9]. The effect of human body on the antenna performance is investigated in this chapter. The proposed antennas may be used as wearable antennas on persons or

Wearable Antennas for Medical Applications

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327

 One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact printed antennas

A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4 0.8mm dielectric substrate. The second layer consists of Kapton 0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The antenna may be attached to the customer shirt in the customer stomach or back zone. The antenna has been analyzed by using

 One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact printed antennas

A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4 0.8mm dielectric substrate. The second layer consists of Kapton 0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The antenna may be attached to the customer shirt in the customer stomach or back zone. The antenna has been analyzed by using

> Dipole Feed Slot Feed

Dipole Feed Slot Feed

A new class of wideband compact printed and microstrip antennas for RFID applications is presented in this chapter.

17

0 5 10 15 20 25 30

dB(S(1,1))=-26.371 7.500MHz

m2 m3

> m1 freq=

> m3 freq=

dB(S(1,1))=-19.991 17.00MHz

dB(S(1,1))=-26.371 7.500MHz

dB(S(1,1))=-19.991 17.00MHz

0 5 10 15 20 25 30

freq, MHz

m2 m3

Figure 40. Printed Compact dual polarized antenna, 64x64x1.6mm

m1 freq=

m3 freq=

freq, MHz

17

The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is around 160º. The computed S11 parameters are presented in Figure

Figure 41. Computed S11 results

S11



Figure 41. Computed S11 results


dB(S(2,2))

dB(S(1,1))

dB(S(1,1))

dB(S(2,2))


**Figure 40.** Printed Compact dual polarized antenna, 64x64x1.6mm

m2 freq=



S11

Figure 40. Printed Compact dual polarized antenna, 64x64x1.6mm

m1

m1

dB(S(1,1))=-21.886 13.50MHz

dB(S(1,1))=-21.886 13.50MHz

> m2 freq=

Agilent ADS software.

 

> 

**Figure 41.** Computed S11 results

**Figure 39.** Measured S11 as function of varactor bias voltage

#### **8. Compact wearable RFID antennas**

**RFID** (**R**adio **F**requency **Id**entification) is an electronic method of exchanging data over radio frequency waves. There are three major components in RFID system: Transponder (Tag), Antenna and a Controller. The RFID tag, antenna and controller may be assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are more effective at penetrating non-metal objects especially objects with high water content.

A new class of wideband compact printed and microstrip antennas for RFID applications is presented in this chapter.

RF transmission properties of human tissues have been investigated in several papers [8-9]. The effect of human body on the antenna performance is investigated in this chapter. The proposed antennas may be used as wearable antennas on persons or animals. The proposed antennas may be attached to cars, trucks, containers and other various objects.

#### **8.1. Dual polarized 13.5MHz compact printed antenna**

One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact printed antennas are crucial in the development of RIFD systems.

A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4

Readout

Readout

Readout

assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are

RF transmission properties of human tissues have been investigated in several papers [8-9]. The effect of human body on the

assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are

antenna performance is investigated in this chapter. The proposed antennas may be used as wearable antennas on persons or

polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The antenna may be attached to the customer shirt in the customer stomach or back zone. The antenna has been analyzed by using

0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The

A new class of wideband compact printed and microstrip antennas for RFID applications is presented in this chapter.

Readout

Readout

Readout

0.8mm dielectric substrate. The second layer consists of Kapton 0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal polarizations. The dual polarized RFID antenna is shown in Figure 40. The antenna dimensions are 6.4x6.4x0.16cm. The antenna may be attached to the customer shirt in the customer stomach or back zone. The antenna has been analyzed by using Agilent ADS software. **9.1 Dual Polarized 13.5MHz Compact Printed Antenna** One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact printed antennas are crucial in the development of RIFD systems. A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4 0.8mm dielectric substrate. The second layer consists of Kapton 0.8mm dielectric substrate. The substrate thickness determines the antenna bandwidth. A printed slot antenna provides a vertical polarization. The proposed antenna is dual polarized. The printed dipole and the slot antenna provide dual orthogonal animals. The proposed antennas may be attached to cars, trucks, containers and other various objects. **9.1 Dual Polarized 13.5MHz Compact Printed Antenna** One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact printed antennas are crucial in the development of RIFD systems. A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4 0.8mm dielectric substrate. The second layer consists of Kapton

more effective at penetrating non-metal objects especially objects with high water content.

 Figure 40. Printed Compact dual polarized antenna, 64x64x1.6mm -14 **Figure 40.** Printed Compact dual polarized antenna, 64x64x1.6mm 


m2 freq=

Figure 41. Computed S11 results

13.50MHz

Figure 40. Printed Compact dual polarized antenna, 64x64x1.6mm

dB(S(2,2))

 

Agilent ADS software.

 17 **Figure 41.** Computed S11 results

300 320 340 360 380 400 420 440 460 480 500

dB(S(1,1))=-29.253 384.0MHz

dB(FOLDED\_VARC\_7V..S(1,1))=-27.767 396.0MHz

m3 freq=

dB(FOLDED\_VARC\_XV..S(1,1))=-31.592 386.0MHz

m1

**RFID** (**R**adio **F**requency **Id**entification) is an electronic method of exchanging data over radio frequency waves. There are three major components in RFID system: Transponder (Tag), Antenna and a Controller. The RFID tag, antenna and controller may be assembled on the same board. Microstrip antennas are widely presented in books and papers in the last decade [1-7]. However, compact wearable printed antennas are not widely used at 13.5MHz RIFD systems. HF tags work best at close range but are more effective at penetrating non-metal objects

A new class of wideband compact printed and microstrip antennas for RFID applications is

RF transmission properties of human tissues have been investigated in several papers [8-9]. The effect of human body on the antenna performance is investigated in this chapter. The proposed antennas may be used as wearable antennas on persons or animals. The proposed

One of the most critical elements of any RFID system is the electrical performance of its antenna. The antenna is the main component for transferring energy from the transmitter to the passive RFID tags, receiving the transponder's replying signal and avoiding in-band interference from electrical noise and other nearby RFID components. Low profile compact

A new compact microstrip loaded dipole antennas has been designed at 13.5MHz to provide horizontal polarization. The antenna consists of two layers. The first layer consists of FR4

antennas may be attached to cars, trucks, containers and other various objects.

8V

9V NO VARACTOR

m3

m4 m1 freq=

m4 freq=

7V

m2

dB(FOLDED\_NOVARC..S(1,1))=-13.968 375.0MHz

freq, MHz


dB(S(1,1))

dB(FOLDED\_NOVARC..S(1,1))

dB(FOLDED\_VARC\_XV..S(1,1))

dB(FOLDED\_VARC\_7V..S(1,1))

326 Advancement in Microstrip Antennas with Recent Applications

**Figure 39.** Measured S11 as function of varactor bias voltage

**8. Compact wearable RFID antennas**

especially objects with high water content.

**8.1. Dual polarized 13.5MHz compact printed antenna**

printed antennas are crucial in the development of RIFD systems.

presented in this chapter.

0

m2 freq=


 17 The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is around 160º. The computed S11 parameters are presented in Figure 41. There is a good agreement between measured and computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure 43. The computed 3D radiation pattern of the antenna is shown in Figure 44.

#### **8.2. Varying the antenna feed network** The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is

Several designs with different feed network have been developed. A compact antenna with different feed network is shown in Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5 compares the electrical performance of a loop antenna with the compact dual polarized antenna. computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure 43. The computed 3D radiation pattern of the antenna is shown in Figure 44. **9.2 Varying the Antenna Feed Network**  Several designs with different feed network have been developed. A compact antenna with different feed network is shown in Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is around 160º. The computed S11 parameters are presented in Figure 41. There is a good agreement between measured and computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure 43. The computed 3D radiation pattern of the antenna is shown in Figure 44. **9.2 Varying the Antenna Feed Network**  

around 160º. The computed S11 parameters are presented in Figure 41. There is a good agreement between measured and

a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5

**Figure 44.** Antenna Radiation pattern

Agilent ADS software.

**8.3. RFID wearable loop antennas**

impedance of a square four turn loop at 13.5MHz is

This matching network has narrow bandwidth.

.

**Figure 45.** RFID printed antenna, 8.4x6.4x0.16cm.

RFID loop antennas are widely used. Several RFID loop antennas are presented in [15]. RFID loop antennas have low efficiency and narrow bandwidth. As an example the measured

0.47 + j107.5 Ω. A matching network is used to match the antenna to 50 Ω. The matching network consists of a 56pF shunt capacitor, 1kΩ shunt resistor and another 56pF capacitor.

A transmitting antenna is placed 20cm from a square four turn loop antenna. 0dBm CW signal is applied to the transmitting antenna. The measured power level at the output of the loop antenna is -50dBm. A square four turn loop antenna has been designed at 13.5MHz by using

Figure 45. RFID printed antenna, 8.4x6.4x0.16cm.

Figure 37. C-V curves of varactors MA46H070 to MA46H076

Wearable Antennas for Medical Applications

http://dx.doi.org/10.5772/54663

329

2

 Several designs with different feed network have been developed. A compact antenna with different feed network is shown in Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5

compares the electrical performance of a loop antenna with the compact dual polarized antenna.

100

 Figure 42. Measured S11 on human body **Figure 42.** Measured S11 on human body S11

Figure 42. Measured S11 on human body

Linear Polarization

18

18

Figure 44. 3D Antenna Radiation pattern

Figure 44. 3D Antenna Radiation pattern

Figure 43. Antenna Radiation pattern

**Figure 43.** Antenna Radiation pattern

 Several designs with different feed network have been developed. A compact antenna with different feed network is shown in **Figure 44.** Antenna Radiation pattern

41. There is a good agreement between measured and computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure

Several designs with different feed network have been developed. A compact antenna with different feed network is shown in Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5 compares the electrical performance of a loop antenna with the compact dual

**9.2 Varying the Antenna Feed Network** 

pattern of the antenna is shown in Figure 44.

**9.2 Varying the Antenna Feed Network** 

pattern of the antenna is shown in Figure 44.

Figure 42. Measured S11 on human body

Figure 42. Measured S11 on human body

Linear Polarization

E\_co E\_cross

E\_co E\_cross

Linear Polarization

0

THETA

20

0

THETA

20

40

40

60

60

80

80

100

100

Figure 43. Antenna Radiation pattern

Figure 43. Antenna Radiation pattern



Mag. [dB]

0

E

Mag. [dB]

E



0



Figure 44. 3D Antenna Radiation pattern

Figure 44. 3D Antenna Radiation pattern

 The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is around 160º. The computed S11 parameters are presented in Figure 41. There is a good agreement between measured and computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure 43. The computed 3D radiation

 The antenna S11parameter is better than -21dB at 13.5MHz. The antenna gain is around -10dBi. The antenna beam width is around 160º. The computed S11 parameters are presented in Figure 41. There is a good agreement between measured and computed results. Figure 42 presents the antenna measured S11 parameters. The antenna cross- polarized field strength may be adjusted by varying the slot feed location. The computed radiation pattern is shown in Figure 43. The computed 3D radiation

Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is

compares the electrical performance of a loop antenna with the compact dual polarized antenna.

compares the electrical performance of a loop antenna with the compact dual polarized antenna.

18

18

43. The computed 3D radiation pattern of the antenna is shown in Figure 44.

**8.2. Varying the antenna feed network**

328 Advancement in Microstrip Antennas with Recent Applications

S11

**Figure 42.** Measured S11 on human body

**Figure 43.** Antenna Radiation pattern

S11

polarized antenna.

#### a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5 Several designs with different feed network have been developed. A compact antenna with different feed network is shown in Figure 45. The antenna dimensions are 8.4x6.4x0.16cm. Figure 46 presents the antenna computed S11on human body. There is **8.3. RFID wearable loop antennas**

a good agreement between measured and computed results. The computed radiation pattern is shown in Fig 47. Table 5 RFID loop antennas are widely used. Several RFID loop antennas are presented in [15]. RFID loop antennas have low efficiency and narrow bandwidth. As an example the measured impedance of a square four turn loop at 13.5MHz is

> 0.47 + j107.5 Ω. A matching network is used to match the antenna to 50 Ω. The matching network consists of a 56pF shunt capacitor, 1kΩ shunt resistor and another 56pF capacitor. This matching network has narrow bandwidth.

> A transmitting antenna is placed 20cm from a square four turn loop antenna. 0dBm CW signal is applied to the transmitting antenna. The measured power level at the output of the loop antenna is -50dBm. A square four turn loop antenna has been designed at 13.5MHz by using Agilent ADS software. Figure 37. C-V curves of varactors MA46H070 to MA46H076


Figure 45. RFID printed antenna, 8.4x6.4x0.16cm.

**Figure 45.** RFID printed antenna, 8.4x6.4x0.16cm.

.

.

Figure 45. RFID printed antenna, 8.4x6.4x0.16cm.

Figure 46. RFID Antenna Computed S11and S22 results

**9.3 RFID Wearable Loop Antennas** 

designed at 13.5MHz by using Agilent ADS software.

RFID loop antennas are widely used. Several RFID loop antennas are presented in [15]. RFID loop antennas have low

0.47 + j107.5 Ω. A matching network is used to match the antenna to 50 Ω. The matching network consists of a 56pF shunt

A transmitting antenna is placed 20cm from a square four turn loop antenna. 0dBm CW signal is applied to the transmitting antenna. The measured power level at the output of the loop antenna is -50dBm. A square four turn loop antenna has been

efficiency and narrow bandwidth. As an example the measured impedance of a square four turn loop at 13.5MHz is

capacitor, 1kΩ shunt resistor and another 56pF capacitor. This matching network has narrow bandwidth.

The antenna is printed on a FR4 substrate. The antenna dimensions are 32x52.4x0.25mm. The antenna layout is shown in figure **Figure 46.** RFID Antenna Computed S11and S22 results

48. S11 results of the printed loop antenna are shown in figure 49. The antenna S11 parameter is better than -9.5dB without an external matching network. The computed radiation pattern is shown in Figure 50. The computed radiation pattern takes into account an infinite ground plane. Antenna Beam width° 3dB Gain dBi VSWR Loop Antenna 140 -25 2:1 Microstrip Antenna 160 -10 1.2:1 The antenna is printed on a FR4 substrate. The antenna dimensions are 32x52.4x0.25mm. The antenna layout is shown in figure 48. S11 results of the printed loop antenna are shown in figure 49. The antenna S11 parameter is better than -9.5dB without an external matching network. The computed radiation pattern is shown in Figure 50. The computed radiation pattern takes into account an infinite ground plane.

TABLE 5. COMPARISON OF LOOP ANTENNA AND MICROSTRIP ANTENNA PARAMETERS

3

Figure 48. A square four turn loop antenna

Figure 48. A square four turn loop antenna

the electrical engineering department.

 **References** 




Mag. [dB]



**Figure 48.** A square four turn loop antenna

1983.

S11

**Figure 49.** Loop Antenna Computed S11 results

2006.

 **A. Sabban** (M'87-SM'94) received the B.Sc and M.Sc degrees in electrical engineering from Tel Aviv University, Israel in 1976 and 1986 respectively. He received the Ph.D.

Figure 47. Compact Antenna, 8.4x6.4x0.16cm, Radiation pattern



Mag. [dB]

E

0


Linear Polarization

E\_co E\_cross

0

THETA

20

40

60

80

100

Wearable Antennas for Medical Applications

http://dx.doi.org/10.5772/54663

331

degree in electrical engineering from Colorado University at Boulder, USA, in 1991. Dr. A. Sabban reasearch interests are microwave and

In 1976 he joined the armament development authority RAFAEL in Israel. In RAFAEL he worked as a senior researcher, group leader and project leader in the electromagnetic department till 2007. In 2007 he retired from RAFAEL. From 2008 to 2010 he worked as an RF Specialist and project leader in Hitech companies. From 2010 to date he is a senior lecturer and researcher in Ort Braude College in Israel in

[2] A. Sabban and K.C. Gupta, "Characterization of Radiation Loss from Microstrip Discontinuities Using a Multiport Network Modeling

[3] A. Sabban," A New Wideband Stacked Microstrip Antenna", I.E.E.E Antenna and Propagation Symp., Houston, Texas, U.S.A, June

[5] R. Kastner, E. Heyman, A. Sabban, "Spectral Domain Iterative Analysis of Single and Double-Layered Microstrip Antennas Using the Conjugate Gradient Algorithm", I.E.E.E Trans. on Antennas and Propagation, Vol. 36, No. 9, Sept. 1988, pp. 1204-1212. [6] A. Sabban, "Wideband Microstrip Antenna Arrays", I.E.E.E Antenna and Propagation Symposium MELCOM, Tel-Aviv,1981. [7] A. Sabban, "Microstrip Antenna Arrays", Microstrip Antennas, Nasimuddin Nasimuddin (Ed.), ISBN: 978-953-307-247-0, InTech,

[8] Lawrence C. Chirwa\*, Paul A. Hammond, Scott Roy, and David R. S. Cumming*,* "Electromagnetic Radiation from Ingested Sources in the Human Intestine between 150 MHz and 1.2 GHz", IEEE Transaction on Biomedical eng., VOL. 50, NO. 4, April 2003, pp 484-492. [9] D.Werber, A. Schwentner, E. M. Biebl, "Investigation of RF transmission properties of human tissues", Adv. Radio Sci., 4, 357–360,

[10] Gupta, B., Sankaralingam S., Dhar, S.,"Development of wearable and implantable antennas in the last decade", Microwave Symposium

[11] Thalmann T., Popovic Z., Notaros B.M, Mosig, J.R.," Investigation and design of a multi-band wearable antenna", 3rd European

[12] Salonen, P., Rahmat-Samii, Y., Kivikoski, M.," Wearable antennas in the vicinity of human body", IEEE Antennas and Propagation

[13] Kellomaki T., Heikkinen J., Kivikoski, M., " Wearable antennas for FM reception", First European Conference on Antennas and

[14] A. Sabban, "Wideband printed antennas for medical applications" APMC 2009 Conference**,** Singapore, 12/2009.

[4] A. Sabban, E. Navon " A MM-Waves Microstrip Antenna Array", I.E.E.E Symposium, Tel-Aviv, March 1983.

MHz

http://www.intechopen.com/articles/show/title/microstrip-antenna-arrays , pp..361-384, 2011.

antenna engineering. He published over 60 research chapters and hold a patent in the antenna area.

S11

[1] J.R. James, P.S Hall and C. Wood, "Microstrip Antenna Theory and Design",1981.

2 4 6 8 10 12 14 16 18 20

Approach", I.E.E.E Trans. on M.T.T, Vol. 39,No. 4,April 1991, pp. 705-712.

Frequency

20

The Microstrip Antenna input impedance variation as function of distance from the body has been computed by employing ADS software. The analyzed structure is presented in Figure 14 Properties of human body tissues are listed in Table 2 see [8]. These properties were used in the antenna design. S11 parameters for different human body thicknesses have been computed. We may note that the differences in the results for body thickness of 15mm to 100mm are negligi‐ ble. S11 parameters for different position relative to the human body have been computed. If the airspacingbetweentheantennaandthehumanbodyisincreasedfrom0mmto10mmtheantenna

(MMS), 2010 Mediterranean 2010 , Page(s): 251 – 267.

Figure 49. Loop Antenna Computed S11 results

Propagation, EuCAP 2006 , pp. 1-6.

S11 parameters may change by less than 1%. The VSWR is better than 1.5:1.

Society International Symposium, 2004. Vol.1 pp. 467 – 470.

Conference on Antennas and Propagation, EuCAP 2009. Pp. 462 – 465.


**Table 5.** Comparison of Loop Antenna and Microstrip antenna parameters

Figure 47. Compact Antenna, 8.4x6.4x0.16cm, Radiation pattern

Figure 48. A square four turn loop antenna



S11


Mag. [dB]



20

2 4 6 8 10 12 14 16 18 20

S11

Frequency

MHz

Figure 49. Loop Antenna Computed S11 results

**Figure 47.** Compact Antenna, 8.4x6.4x0.16cm, Radiation pattern

Linear Polarization

E\_co E\_cross

0

20

40

60

80

100

Figure 48. A square four turn loop antenna **Figure 48.** A square four turn loop antenna

19

Linear Polarization

E\_co E\_cross

0

THETA

20

40

60

80

100

Figure 47. Compact Antenna, 8.4x6.4x0.16cm, Radiation pattern

Figure 48. A square four turn loop antenna



S11


Mag. [dB]



20

2 4 6 8 10 12 14 16 18 20

S11

Frequency

MHz

Figure 49. Loop Antenna Computed S11 results

Antenna Beam width° 3dB Gain dBi VSWR Loop Antenna 140 -25 2:1 Microstrip Antenna 160 -10 1.2:1

Antenna Beam width° 3dB Gain dBi VSWR Loop Antenna 140 -25 2:1 Microstrip Antenna 160 -10 1.2:1

The antenna is printed on a FR4 substrate. The antenna dimensions are 32x52.4x0.25mm. The antenna layout is shown in figure 48. S11 results of the printed loop antenna are shown in figure 49. The antenna S11 parameter is better than -9.5dB without an external matching network. The computed radiation pattern is shown in Figure 50. The computed radiation

2 4 6 8 10 12 14 16 18 20

dB(S(1,1))=-17.949 13.50MHz

m2 freq= m1 freq=

m2

dB(S(1,1))=-24.986 6.000MHz

freq, MHz

TABLE 5. COMPARISON OF LOOP ANTENNA AND MICROSTRIP ANTENNA PARAMETERS

**9.3 RFID Wearable Loop Antennas** 

designed at 13.5MHz by using Agilent ADS software.

Figure 45. RFID printed antenna, 8.4x6.4x0.16cm.

m1

Figure 46. RFID Antenna Computed S11and S22 results

account an infinite ground plane.



S11and S22


dB(S(1,1))

dB(S(2,2))

**Figure 46.** RFID Antenna Computed S11and S22 results

pattern takes into account an infinite ground plane.

**Table 5.** Comparison of Loop Antenna and Microstrip antenna parameters



Mag. [dB]

E

**Figure 47.** Compact Antenna, 8.4x6.4x0.16cm, Radiation pattern

0



.

330 Advancement in Microstrip Antennas with Recent Applications

RFID loop antennas are widely used. Several RFID loop antennas are presented in [15]. RFID loop antennas have low

0.47 + j107.5 Ω. A matching network is used to match the antenna to 50 Ω. The matching network consists of a 56pF shunt

A transmitting antenna is placed 20cm from a square four turn loop antenna. 0dBm CW signal is applied to the transmitting antenna. The measured power level at the output of the loop antenna is -50dBm. A square four turn loop antenna has been

The antenna is printed on a FR4 substrate. The antenna dimensions are 32x52.4x0.25mm. The antenna layout is shown in figure 48. S11 results of the printed loop antenna are shown in figure 49. The antenna S11 parameter is better than -9.5dB without an external matching network. The computed radiation pattern is shown in Figure 50. The computed radiation pattern takes into

efficiency and narrow bandwidth. As an example the measured impedance of a square four turn loop at 13.5MHz is

capacitor, 1kΩ shunt resistor and another 56pF capacitor. This matching network has narrow bandwidth.

[3] A. Sabban," A New Wideband Stacked Microstrip Antenna", I.E.E.E Antenna and Propagation Symp., Houston, Texas, U.S.A, June

[5] R. Kastner, E. Heyman, A. Sabban, "Spectral Domain Iterative Analysis of Single and Double-Layered Microstrip Antennas Using the

[13] Kellomaki T., Heikkinen J., Kivikoski, M., " Wearable antennas for FM reception", First European Conference on Antennas and

[14] A. Sabban, "Wideband printed antennas for medical applications" APMC 2009 Conference**,** Singapore, 12/2009.

 **A. Sabban** (M'87-SM'94) received the B.Sc

Figure 48. A square four turn loop antenna



Mag. [dB]

E

0


1983. [4] A. Sabban, E. Navon " A MM-Waves Microstrip Antenna Array", I.E.E.E Symposium, Tel-Aviv, March 1983. Figure 49. Loop Antenna Computed S11 results **Figure 49.** Loop Antenna Computed S11 results

Conjugate Gradient Algorithm", I.E.E.E Trans. on Antennas and Propagation, Vol. 36, No. 9, Sept. 1988, pp. 1204-1212. [6] A. Sabban, "Wideband Microstrip Antenna Arrays", I.E.E.E Antenna and Propagation Symposium MELCOM, Tel-Aviv,1981. [7] A. Sabban, "Microstrip Antenna Arrays", Microstrip Antennas, Nasimuddin Nasimuddin (Ed.), ISBN: 978-953-307-247-0, InTech, http://www.intechopen.com/articles/show/title/microstrip-antenna-arrays , pp..361-384, 2011. [8] Lawrence C. Chirwa\*, Paul A. Hammond, Scott Roy, and David R. S. Cumming*,* "Electromagnetic Radiation from Ingested Sources in the Human Intestine between 150 MHz and 1.2 GHz", IEEE Transaction on Biomedical eng., VOL. 50, NO. 4, April 2003, pp 484-492. [9] D.Werber, A. Schwentner, E. M. Biebl, "Investigation of RF transmission properties of human tissues", Adv. Radio Sci., 4, 357–360, 2006. [10] Gupta, B., Sankaralingam S., Dhar, S.,"Development of wearable and implantable antennas in the last decade", Microwave Symposium (MMS), 2010 Mediterranean 2010 , Page(s): 251 – 267. [11] Thalmann T., Popovic Z., Notaros B.M, Mosig, J.R.," Investigation and design of a multi-band wearable antenna", 3rd European Conference on Antennas and Propagation, EuCAP 2009. Pp. 462 – 465. [12] Salonen, P., Rahmat-Samii, Y., Kivikoski, M.," Wearable antennas in the vicinity of human body", IEEE Antennas and Propagation Society International Symposium, 2004. Vol.1 pp. 467 – 470. 20 The Microstrip Antenna input impedance variation as function of distance from the body has been computed by employing ADS software. The analyzed structure is presented in Figure 14 Properties of human body tissues are listed in Table 2 see [8]. These properties were used in the antenna design. S11 parameters for different human body thicknesses have been computed. We may note that the differences in the results for body thickness of 15mm to 100mm are negligi‐ ble. S11 parameters for different position relative to the human body have been computed. If the airspacingbetweentheantennaandthehumanbodyisincreasedfrom0mmto10mmtheantenna S11 parameters may change by less than 1%. The VSWR is better than 1.5:1.

Propagation, EuCAP 2006 , pp. 1-6.

The Microstrip Antenna input impedance variation as function of distance from the body has been computed by employing ADS software. The analyzed structure is presented in Figure 14 Properties of human body tissues are listed in Table 2 see [8]. These

relative to the human body have been computed. If the air spacing between the antenna and the human body is increased from

In RFID systems the distance between the transmitting and receiving antennas is less than 2D²/λ, where D is the largest dimension of the antenna. The receiving and transmitting antennas are magnetically coupled. In these applications we refer to the near field and not

Wearable Antennas for Medical Applications

http://dx.doi.org/10.5772/54663

333

Figure 52 and Figure 53 present compact printed antenna for RFID applications. The presented

This chapter presents wideband microstrip antennas with high efficiency for medical appli‐ cations. The antenna dimensions may vary from 26x6x0.16cm to 5x5x0.05cm according to the medical system specification. The antennas bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antennas gain varies from 0 to 4dBi. The antenna S11 results for different belt thickness, shirt thickness and air spacing between the antennas and human body are presented in this chapter. If the air spacing between the new

antennas may be assembled in a belt and attached to the patient stomach or back.

to the far field radiation pattern.

**Figure 52.** New Microstrip Antenna for RFID applications

**Figure 53.** Loop Antenna for RFID applications

**9. Conclusions**

the customer stomach. The antennas may be employed as transmitting or as receiving antennas. The antennas may receive or

0mm to 10mm the antenna S11 parameters may change by less than 1%. The VSWR is better than 1.5:1.

**Figure 50.** Loop Antenna Radiation patterns for an infinite ground plane

recorder

RFID Antenna

#### An application of the proposed antenna is shown in Figure 51. The RFID antennas may be assembled in a belt and attached to **8.4. Proposed antenna applications**

transmit information to medical systems. Loop or Printed An application of the proposed antenna is shown in Figure 51. The RFID antennas may be assembled in a belt and attached to the customer stomach. The antennas may be employed as transmitting or as receiving antennas. The antennas may receive or transmit information to medical systems.

> Wearable Diversity antennas

**9.4 PROPOSED ANTENNA APPLICATIONS** 

Figure 50. Loop Antenna Radiation patterns for an infinite ground plane

21

**Figure 51.** Wearable RFID antenna

In RFID systems the distance between the transmitting and receiving antennas is less than 2D²/λ, where D is the largest dimension of the antenna. The receiving and transmitting antennas are magnetically coupled. In these applications we refer to the near field and not to the far field radiation pattern.

the customer stomach. The antennas may be employed as transmitting or as receiving antennas. The antennas may receive or **Figure 52.** New Microstrip Antenna for RFID applications

Figure 52 and Figure 53 present compact printed antenna for RFID applications. The presented antennas may be assembled in a belt and attached to the patient stomach or back.

**Figure 53.** Loop Antenna for RFID applications

#### **9. Conclusions**

21

Figure 52. New Microstrip Antenna for RFID applications

recorder

The Microstrip Antenna input impedance variation as function of distance from the body has been computed by employing ADS software. The analyzed structure is presented in Figure 14 Properties of human body tissues are listed in Table 2 see [8]. These properties were used in the antenna design. S11 parameters for different human body thicknesses have been computed. We may note that the differences in the results for body thickness of 15mm to 100mm are negligible. S11 parameters for different position relative to the human body have been computed. If the air spacing between the antenna and the human body is increased from

An application of the proposed antenna is shown in Figure 51. The RFID antennas may be assembled in a belt and attached to

 In RFID systems the distance between the transmitting and receiving antennas is less than 2D²/λ, where D is the largest dimension of the antenna. The receiving and transmitting antennas are magnetically coupled. In these applications we refer to

0mm to 10mm the antenna S11 parameters may change by less than 1%. The VSWR is better than 1.5:1.

Figure 50. Loop Antenna Radiation patterns for an infinite ground plane

50

100

150

200

**9.4 PROPOSED ANTENNA APPLICATIONS** 

An application of the proposed antenna is shown in Figure 51. The RFID antennas may be assembled in a belt and attached to the customer stomach. The antennas may be employed as transmitting or as receiving antennas. The antennas may receive or transmit information to

0

THETA


E\_co E\_cross

Wearable Diversity antennas

Belt

Wearable Diversity antennas

Belt

transmit information to medical systems.

 Loop or Printed RFID Antenna



Mag. [dB]

332 Advancement in Microstrip Antennas with Recent Applications

E

**8.4. Proposed antenna applications**

medical systems.

**Figure 51.** Wearable RFID antenna

0


**Figure 50.** Loop Antenna Radiation patterns for an infinite ground plane


recorder

Figure 51. Wearable RFID antenna

the near field and not to the far field radiation pattern.

 Loop or Printed RFID Antenna

> This chapter presents wideband microstrip antennas with high efficiency for medical appli‐ cations. The antenna dimensions may vary from 26x6x0.16cm to 5x5x0.05cm according to the medical system specification. The antennas bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antennas gain varies from 0 to 4dBi. The antenna S11 results for different belt thickness, shirt thickness and air spacing between the antennas and human body are presented in this chapter. If the air spacing between the new

dual polarized antenna and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. However, if the air spacing between the helix antenna and the human body is increased only from 0mm to 2mm the antenna resonant frequency is shifted by 5%. The effect of the antenna location on the human body should be considered in the antenna design process. The proposed antenna may be used in Medicare RF systems.

[4] Sabban, E. Navon " A MM-Waves Microstrip Antenna Array", I.E.E.E Symposium,

Wearable Antennas for Medical Applications

http://dx.doi.org/10.5772/54663

335

[5] R. Kastner, E. Heyman, A. Sabban, "Spectral Domain Iterative Analysis of Single and Double-Layered Microstrip Antennas Using the Conjugate Gradient Algorithm", I.E.E.E Trans. on Antennas and Propagation, Vol. 36, No. 9, Sept. 1988, pp. 1204-1212.

[6] Sabban, "Wideband Microstrip Antenna Arrays", I.E.E.E Antenna and Propagation

[7] Sabban, "Microstrip Antenna Arrays", Microstrip Antennas, Nasimuddin Nasimud‐ din (Ed.), ISBN: 978-953-307-247-0, InTech, http://www.intechopen.com/articles/

[8] Lawrence C. Chirwa\*, Paul A. Hammond, Scott Roy, and David R. S. Cumming, "Electromagnetic Radiation from Ingested Sources in the Human Intestine between 150 MHz and 1.2 GHz", IEEE Transaction on Biomedical eng., VOL. 50, NO. 4, April

[9] D.Werber, A. Schwentner, E. M. Biebl, "Investigation of RF transmission properties of

[10] Gupta, B., Sankaralingam S., Dhar, S.,"Development of wearable and implantable an‐ tennas in the last decade", Microwave Symposium (MMS), 2010 Mediterranean 2010 ,

[11] Thalmann T., Popovic Z., Notaros B.M, Mosig, J.R.," Investigation and design of a multi-band wearable antenna", 3rd European Conference on Antennas and Propaga‐

[12] Salonen, P., Rahmat-Samii, Y., Kivikoski, M.," Wearable antennas in the vicinity of human body", IEEE Antennas and Propagation Society International Symposium,

[13] Kellomaki T., Heikkinen J., Kivikoski, M., " Wearable antennas for FM reception", First European Conference on Antennas and Propagation, EuCAP 2006 , pp. 1-6. [14] Sabban, "Wideband printed antennas for medical applications" APMC 2009 Confer‐

[15] Youbok Lee, "Antenna Circuit Design for RFID Applications", Microchip Technology

[16] ADS software, Agilent http://www.home.agilent.com/agilent/product.jspx?

cc=IL&lc=eng&ckey=1297113&nid=-34346.0.00&id=1297113

Tel-Aviv, March 1983.

2003, pp 484-492.

Page(s): 251 – 267.

tion, EuCAP 2009. Pp. 462 – 465.

2004. Vol.1 pp. 467 – 470.

ence, Singapore, 12/2009.

Inc., Microchip AN 710c.

Symposium MELCOM, Tel-Aviv,1981.

show/title/microstrip-antenna-arrays , pp..361-384, 2011.

human tissues", Adv. Radio Sci., 4, 357–360, 2006.

A wideband tunable microstrip antennas with high efficiency for medical applications has been presented in this chapter. The antenna dimensions may vary from 26x6x0.16cm to 5x5x0.05cm according to the medical system specification. The antennas bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antennas gain varies from 0 to 2dBi. If the air spacing between the dual polarized antenna and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. A varactor is employed to compensate variations in the antenna resonant frequency at different locations on the human body.

This chapter presents also wideband compact printed antennas, Microstrip and Loop anten‐ nas, for RFID applications. The antenna beam width is around 160º. The antenna gain is around -10dBdBi. The proposed antennas may be used as wearable antennas on persons or animals. The proposed antennas may be attached to cars, trucks and other various objects. If the air spacing between the antenna and the human body is increased from 0mm to 10mm the antenna S11 parameters may change by less than 1%. The antenna VSWR is better than 1.5:1 for all tested environments.

#### **Author details**

Albert Sabban1,2,3\*

1 Ort Braude College, Karmiel, Israel

2 Tel Aviv University, Israel

3 Colorado University, Boulder, USA

#### **References**


[4] Sabban, E. Navon " A MM-Waves Microstrip Antenna Array", I.E.E.E Symposium, Tel-Aviv, March 1983.

dual polarized antenna and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. However, if the air spacing between the helix antenna and the human body is increased only from 0mm to 2mm the antenna resonant frequency is shifted by 5%. The effect of the antenna location on the human body should be considered in the antenna design process. The proposed antenna may be used in Medicare RF systems.

A wideband tunable microstrip antennas with high efficiency for medical applications has been presented in this chapter. The antenna dimensions may vary from 26x6x0.16cm to 5x5x0.05cm according to the medical system specification. The antennas bandwidth is around 10% for VSWR better than 2:1. The antenna beam width is around 100º. The antennas gain varies from 0 to 2dBi. If the air spacing between the dual polarized antenna and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%. A varactor is employed to compensate variations in the antenna resonant frequency at different

This chapter presents also wideband compact printed antennas, Microstrip and Loop anten‐ nas, for RFID applications. The antenna beam width is around 160º. The antenna gain is around -10dBdBi. The proposed antennas may be used as wearable antennas on persons or animals. The proposed antennas may be attached to cars, trucks and other various objects. If the air spacing between the antenna and the human body is increased from 0mm to 10mm the antenna S11 parameters may change by less than 1%. The antenna VSWR is better than 1.5:1 for all tested

[1] J.R. James, P.S Hall and C. Wood, "Microstrip Antenna Theory and Design",1981.

[2] Sabban and K.C. Gupta, "Characterization of Radiation Loss from Microstrip Discon‐ tinuities Using a Multiport Network Modeling Approach", I.E.E.E Trans. on M.T.T,

[3] Sabban," A New Wideband Stacked Microstrip Antenna", I.E.E.E Antenna and Prop‐

locations on the human body.

334 Advancement in Microstrip Antennas with Recent Applications

environments.

**Author details**

Albert Sabban1,2,3\*

**References**

1 Ort Braude College, Karmiel, Israel

3 Colorado University, Boulder, USA

Vol. 39,No. 4,April 1991, pp. 705-712.

agation Symp., Houston, Texas, U.S.A, June 1983.

2 Tel Aviv University, Israel


**Chapter 14**

**Provisional chapter**

**Reconfigurable Microstrip Antennas for Cognitive**

**Reconfigurable Microstrip Antennas for Cognitive**

An increasing demand for radio spectrum has resulted from the emergence of feature-rich and high-data-rate wireless applications. The spectrum is scarce, and the current radio spectrum regulations make its use inefficient. This necessitates the development of new

According to the current spectrum allocation regulations, specific bands are assigned to particular services, and only licensed users are granted access to licensed bands. Cognitive radio (CR) is expected to revolutionize the way spectrum is allocated. In a CR network, the intelligent radio part allows unlicensed users (secondary users) to access spectrum bands

Two approaches to sharing spectrum between primary and secondary users have been considered: spectrum underlay and spectrum overlay. In the underlay approach, secondary users should operate below the noise floor of primary users, and thus severe constraints are imposed on their transmission power. Ultra-wideband (UWB) technology is very suitable as the enabling technology for this approach. In spectrum overlay CR, secondary users search

In this chapter, we report and discuss antenna designs for overlay and underlay CR. We start by studying techniques employed in the design of UWB antennas. This is done in Section 3. Such antennas are used for underlay CR, but also for channel sensing in overlay CR. We then move in Section 4 to antennas that allow the use of UWB in overlay CR. These are basically UWB antennas, but have the ability to selectively induce frequency notches in the bands of primary services, thus preventing any interference to them and giving the UWB transmitters used by the secondary users the chance to increase their power, and hence to achieve long-distance communication. In Section 5, we investigate the design of antennas for overlay CR. In this scheme, an antenna should be able to monitor the spectrum (sensing),

> ©2012 Al-Husseini et al., licensee InTech. This is an open access chapter distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Al-Husseini et al.; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

> © 2012 Al-Husseini et al.; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

for unused frequency bands, called white spaces, and use them for communication.

dynamic spectrum allocation policies to better exploit the existing spectrum.

licensed to primary users, while avoiding interference with them.

Mohammed Al-Husseini, Karim Y. Kabalan, Ali El-Hajj and Christos G. Christodoulou

Mohammed Al-Husseini, Karim Y. Kabalan, Ali El-Hajj and Christos G. Christodoulou

Additional information is available at the end of the chapter

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/53430

**1. Introduction**

**Radio**

**Radio**

**Provisional chapter**

#### **Reconfigurable Microstrip Antennas for Cognitive Radio Reconfigurable Microstrip Antennas for Cognitive Radio**

Mohammed Al-Husseini, Karim Y. Kabalan, Ali El-Hajj and Christos G. Christodoulou Mohammed Al-Husseini, Karim Y. Kabalan, Ali El-Hajj and Christos G. Christodoulou

Additional information is available at the end of the chapter

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/53430

#### **1. Introduction**

An increasing demand for radio spectrum has resulted from the emergence of feature-rich and high-data-rate wireless applications. The spectrum is scarce, and the current radio spectrum regulations make its use inefficient. This necessitates the development of new dynamic spectrum allocation policies to better exploit the existing spectrum.

According to the current spectrum allocation regulations, specific bands are assigned to particular services, and only licensed users are granted access to licensed bands. Cognitive radio (CR) is expected to revolutionize the way spectrum is allocated. In a CR network, the intelligent radio part allows unlicensed users (secondary users) to access spectrum bands licensed to primary users, while avoiding interference with them.

Two approaches to sharing spectrum between primary and secondary users have been considered: spectrum underlay and spectrum overlay. In the underlay approach, secondary users should operate below the noise floor of primary users, and thus severe constraints are imposed on their transmission power. Ultra-wideband (UWB) technology is very suitable as the enabling technology for this approach. In spectrum overlay CR, secondary users search for unused frequency bands, called white spaces, and use them for communication.

In this chapter, we report and discuss antenna designs for overlay and underlay CR. We start by studying techniques employed in the design of UWB antennas. This is done in Section 3. Such antennas are used for underlay CR, but also for channel sensing in overlay CR. We then move in Section 4 to antennas that allow the use of UWB in overlay CR. These are basically UWB antennas, but have the ability to selectively induce frequency notches in the bands of primary services, thus preventing any interference to them and giving the UWB transmitters used by the secondary users the chance to increase their power, and hence to achieve long-distance communication. In Section 5, we investigate the design of antennas for overlay CR. In this scheme, an antenna should be able to monitor the spectrum (sensing),

Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2013 Al-Husseini et al.; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2012 Al-Husseini et al.; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

©2012 Al-Husseini et al., licensee InTech. This is an open access chapter distributed under the terms of the

and communicate over a chosen white space (communication). For the latter operation, the antenna must be frequency-reconfigurable. Single- and dual-port antennas for overlay CR can be designed. In the dual-port case, one port has UWB frequency response and is used for channel sensing, and the second port, which is frequency-reconfigurable, is used for communicating. In the more challenging single-port design, the same port can have UWB response for sensing, and can be reconfigured for tunable narrowband operation when required to communicate over a white space.

choose the suitable operating frequency (frequency agility), and the ability to adapt the modulation/coding schemes and transmit power as needed. The self-organized capability has to do with the possession of a good spectrum management scheme, a good mobility and connection management, and the ability to to support security functions in dynamic

Reconfigurable Microstrip Antennas for Cognitive Radio

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339

Dynamic spectrum access (DSA) represents the opposite direction of the current static spectrum management policy. It is broadly categorized under three models: the dynamic exclusive use model, the open sharing model, and the hierarchical access model. The

In the dynamic exclusive use model, the spectrum bands are still licensed to services for exclusive use, as in the current spectrum regulation policy, but flexibility is introduced to improve spectrum efficiency. Two approaches have been proposed under this model: spectrum property rights and dynamic spectrum allocation. The first approach, the spectrum property rights, allows licensees to sell and trade spectrum and to freely choose technology. In the second approach, the dynamic spectrum allocation, the aim is to improve spectrum efficiency through dynamic spectrum assignment by exploiting the spatial and temporal

The open sharing model employs open sharing among peer users as the basis for managing a spectral region. Supporters of this model rely on the huge success of wireless services

A hierarchical access structure with primary and secondary users is adopted by the third model. Here, the spectrum licensed to primary users is open to secondary users while limiting interference to the primary users. Two approaches to spectrum sharing between primary and secondary users have been considered: spectrum underlay and spectrum

In the underlay approach, secondary users should operate below the noise floor of primary users, and thus severe constraints are imposed on their transmission power. One way to achieve this is to spread the transmitted signals of secondary users over an ultra-wide frequency band (UWB), leading to a short-range high data rate with extremely low

environments.

**2.2. Spectrum sharing approaches**

**Figure 1.** Dynamic spectrum access models [3]

traffic statistics of different services.

operating in the ISM band.

overlay.

taxonomy of DSA is illustrated in Fig. 1 [3].

#### **2. Dynamic spectrum access and cognitive radio**

The increasing demand for wireless connectivity and current crowding of licensed and unlicensed spectra necessitate a new communication paradigm to exploit the existing spectrum in better ways. The current approach for spectrum allocation is based on assigning a specific band to a particular service. The FCC Spectrum Policy Task Force [1] reported vast temporal and geographic variations in the usage of allocated spectrum with utilization ranging from 15 to 85% in the bands below 3 GHz. In the frequency range above 3 GHz the bands are even more poorly utilized. In other words, a large portion of the assigned spectrum is used sporadically, leading to an under utilization of a significant amount of spectrum. This inefficiency arises from the inflexibility of the regulatory and licensing process, which typically assigns the complete rights to a frequency band to a primary user. This approach makes it extremely difficult to recycle these bands once they are allocated, even if these users poorly utilize this valuable resource. A solution to this inefficiency, which has been highly successful in the ISM (2.4 GHz), the U-NII (5–6 GHz), and microwave (57–64 GHz) bands, is to make spectra available on an unlicensed basis. However, in order to obtain spectra for unlicensed operation, new sharing concepts have been introduced to allow use by secondary users under the requirement that they limit their interference to pre-existing primary users.

#### **2.1. Cognitive radio**

Cognitive radio (CR) technology is key enabling technology which provides the capability to share the wireless channel with the licensed users in an opportunistic way. CRs are foreseen to be able to provide the high bandwidth to mobile users via heterogeneous wireless architectures and dynamic spectrum access techniques.

In order to share the spectrum with licensed users without interfering with them, and meet the diverse quality of service requirements of applications, each CR user in a CR network must [2]:


To fulfill these functions of spectrum sensing, spectrum decision, spectrum sharing and spectrum mobility, a CR has to be cognitive, reconfigurable and self-organized. An example of the cognitive capability is the CR's ability to sense the spectrum and detect spectrum holes (also called white spaces), which are those frequency bands not used by the licensed users. The reconfigurable capability can be summarized by the ability to dynamically choose the suitable operating frequency (frequency agility), and the ability to adapt the modulation/coding schemes and transmit power as needed. The self-organized capability has to do with the possession of a good spectrum management scheme, a good mobility and connection management, and the ability to to support security functions in dynamic environments.

#### **2.2. Spectrum sharing approaches**

2 Advancements in Microstrip and Printed Antennas

required to communicate over a white space.

**2.1. Cognitive radio**

must [2]:

mobility.

**2. Dynamic spectrum access and cognitive radio**

architectures and dynamic spectrum access techniques.

and communicate over a chosen white space (communication). For the latter operation, the antenna must be frequency-reconfigurable. Single- and dual-port antennas for overlay CR can be designed. In the dual-port case, one port has UWB frequency response and is used for channel sensing, and the second port, which is frequency-reconfigurable, is used for communicating. In the more challenging single-port design, the same port can have UWB response for sensing, and can be reconfigured for tunable narrowband operation when

The increasing demand for wireless connectivity and current crowding of licensed and unlicensed spectra necessitate a new communication paradigm to exploit the existing spectrum in better ways. The current approach for spectrum allocation is based on assigning a specific band to a particular service. The FCC Spectrum Policy Task Force [1] reported vast temporal and geographic variations in the usage of allocated spectrum with utilization ranging from 15 to 85% in the bands below 3 GHz. In the frequency range above 3 GHz the bands are even more poorly utilized. In other words, a large portion of the assigned spectrum is used sporadically, leading to an under utilization of a significant amount of spectrum. This inefficiency arises from the inflexibility of the regulatory and licensing process, which typically assigns the complete rights to a frequency band to a primary user. This approach makes it extremely difficult to recycle these bands once they are allocated, even if these users poorly utilize this valuable resource. A solution to this inefficiency, which has been highly successful in the ISM (2.4 GHz), the U-NII (5–6 GHz), and microwave (57–64 GHz) bands, is to make spectra available on an unlicensed basis. However, in order to obtain spectra for unlicensed operation, new sharing concepts have been introduced to allow use by secondary users under the requirement that they limit their interference to pre-existing primary users.

Cognitive radio (CR) technology is key enabling technology which provides the capability to share the wireless channel with the licensed users in an opportunistic way. CRs are foreseen to be able to provide the high bandwidth to mobile users via heterogeneous wireless

In order to share the spectrum with licensed users without interfering with them, and meet the diverse quality of service requirements of applications, each CR user in a CR network

• Determine the portion of spectrum that is available, which is known as Spectrum sensing.

• Coordinate access to this channel with other users, which is known as Spectrum sharing. • Vacate the channel when a licensed user is detected, which is referred as Spectrum

To fulfill these functions of spectrum sensing, spectrum decision, spectrum sharing and spectrum mobility, a CR has to be cognitive, reconfigurable and self-organized. An example of the cognitive capability is the CR's ability to sense the spectrum and detect spectrum holes (also called white spaces), which are those frequency bands not used by the licensed users. The reconfigurable capability can be summarized by the ability to dynamically

• Select the best available channel, which is called Spectrum decision.

Dynamic spectrum access (DSA) represents the opposite direction of the current static spectrum management policy. It is broadly categorized under three models: the dynamic exclusive use model, the open sharing model, and the hierarchical access model. The taxonomy of DSA is illustrated in Fig. 1 [3].

**Figure 1.** Dynamic spectrum access models [3]

In the dynamic exclusive use model, the spectrum bands are still licensed to services for exclusive use, as in the current spectrum regulation policy, but flexibility is introduced to improve spectrum efficiency. Two approaches have been proposed under this model: spectrum property rights and dynamic spectrum allocation. The first approach, the spectrum property rights, allows licensees to sell and trade spectrum and to freely choose technology. In the second approach, the dynamic spectrum allocation, the aim is to improve spectrum efficiency through dynamic spectrum assignment by exploiting the spatial and temporal traffic statistics of different services.

The open sharing model employs open sharing among peer users as the basis for managing a spectral region. Supporters of this model rely on the huge success of wireless services operating in the ISM band.

A hierarchical access structure with primary and secondary users is adopted by the third model. Here, the spectrum licensed to primary users is open to secondary users while limiting interference to the primary users. Two approaches to spectrum sharing between primary and secondary users have been considered: spectrum underlay and spectrum overlay.

In the underlay approach, secondary users should operate below the noise floor of primary users, and thus severe constraints are imposed on their transmission power. One way to achieve this is to spread the transmitted signals of secondary users over an ultra-wide frequency band (UWB), leading to a short-range high data rate with extremely low

In general, the guidelines to design UWB antennas include:

two of them will be described in this Section.

configuration of this antenna is shown in Fig. 3.

**3.1. Combination of UWB techniques**

discussed in 3.2,

size.

techniques.

current flow, and as a result to better wideband characteristics,

connections, inset feed, or slits under the feed in the ground plane,

• The proper selection of the patch shape. Round shapes and round edges lead to smoother

Reconfigurable Microstrip Antennas for Cognitive Radio

http://dx.doi.org/10.5772/53430

341

• The good design of the ground plane. Partial ground planes, and ground planes with specially designed slots, play a major role in obtaining UWB response. This property is

• The matching between the feed line and the patch. This is achieved using either tapered

• The use of fractal shapes, which are known for their self-repetitive characteristic, used to obtain multi- and wide-band operation, and their space-filling property, which leads to increasing the electrical length of the antenna without tampering with its overall physical

To investigate the above guidelines, several UWB antennas have been designed [10–14]. Only

The UWB design presented in [12] features a microstrip feed line with two 45◦ bends and a tapered section for size reduction and matching, respectively. The ground plane is partial and comprises a rectangular part and a trapezoidal part. The patch is a half ellipse with the cut made along the minor axis. Four slots whose location and size relate to a modified Sierpinski carpet, with the ellipse as the basic shape, are incorporated into the patch. The

(a) (b)

**Figure 3.** (a) Configuration and (b) photo of the UWB antenna in [12]. The antenna combines several bandwidth enhancement

**Figure 2.** Underlay (a) and overlay (b) spectrum sharing approaches

transmission power (less than -42 dBm/MHz in the 3.1–10.6 GHz band). Assuming that primary users transmit all the time (worst case scenario), this approach does not rely on detection and exploitation of spectrum white space.

The spectrum overlay approach, also termed opportunistic spectrum access or OSA, imposes restrictions on when and where secondary users may transmit rather on their transmission power. In this approach, secondary users avoid higher priority users through the use of spectrum sensing and adaptive allocation. They identify and exploit the spectrum holes defined in space, time, and frequency.

The underlay and overlay approaches in the hierarchical model are illustrated in Fig. 2. They can be employed simultaneously for further spectrum efficiency improvement. Furthermore, the hierarchical model is more compatible with current spectrum management policies and legacy wireless systems as compared to the other two models.

#### **3. UWB antennas**

UWB antennas are required for underlay CR, and for sensing in overlay CR. UWB antennas were originally meant to radiate very short pulses over short distances. They have been used in medical applications, GPRs, and other short-range communications requiring high throughputs. The literature is rich with articles pertaining to the design of UWB antennas [4–9]. For example, the authors in [4] present a UWB knight's helm shape antenna fabricated on an FR4 board with a double slotted rectangular patch tapered from a 50-Ω feed line, and a partial ground plane flushed with the feed line. Three techniques are applied for good impedance matching over the UWB range: 1) the dual slots on the rectangular patch, 2) the tapered connection between the rectangular patch and the feed line, and 3) a partial ground plane flushed with the feed line. Consistent omnidirectional radiation patterns and a small group delay characterize this UWB antenna.

In general, the guidelines to design UWB antennas include:

4 Advancements in Microstrip and Printed Antennas

**Figure 2.** Underlay (a) and overlay (b) spectrum sharing approaches

detection and exploitation of spectrum white space.

legacy wireless systems as compared to the other two models.

defined in space, time, and frequency.

group delay characterize this UWB antenna.

**3. UWB antennas**

transmission power (less than -42 dBm/MHz in the 3.1–10.6 GHz band). Assuming that primary users transmit all the time (worst case scenario), this approach does not rely on

The spectrum overlay approach, also termed opportunistic spectrum access or OSA, imposes restrictions on when and where secondary users may transmit rather on their transmission power. In this approach, secondary users avoid higher priority users through the use of spectrum sensing and adaptive allocation. They identify and exploit the spectrum holes

The underlay and overlay approaches in the hierarchical model are illustrated in Fig. 2. They can be employed simultaneously for further spectrum efficiency improvement. Furthermore, the hierarchical model is more compatible with current spectrum management policies and

UWB antennas are required for underlay CR, and for sensing in overlay CR. UWB antennas were originally meant to radiate very short pulses over short distances. They have been used in medical applications, GPRs, and other short-range communications requiring high throughputs. The literature is rich with articles pertaining to the design of UWB antennas [4–9]. For example, the authors in [4] present a UWB knight's helm shape antenna fabricated on an FR4 board with a double slotted rectangular patch tapered from a 50-Ω feed line, and a partial ground plane flushed with the feed line. Three techniques are applied for good impedance matching over the UWB range: 1) the dual slots on the rectangular patch, 2) the tapered connection between the rectangular patch and the feed line, and 3) a partial ground plane flushed with the feed line. Consistent omnidirectional radiation patterns and a small


To investigate the above guidelines, several UWB antennas have been designed [10–14]. Only two of them will be described in this Section.

#### **3.1. Combination of UWB techniques**

The UWB design presented in [12] features a microstrip feed line with two 45◦ bends and a tapered section for size reduction and matching, respectively. The ground plane is partial and comprises a rectangular part and a trapezoidal part. The patch is a half ellipse with the cut made along the minor axis. Four slots whose location and size relate to a modified Sierpinski carpet, with the ellipse as the basic shape, are incorporated into the patch. The configuration of this antenna is shown in Fig. 3.

**Figure 3.** (a) Configuration and (b) photo of the UWB antenna in [12]. The antenna combines several bandwidth enhancement techniques.

**Figure 4.** Reflection coefficient of the UWB antenna in Fig. 3

Four techniques are applied for good impedance matching over the UWB range: 1) the specially selected patch shape, 2) the tapered connection between the patch and the feed line, 3) the optimized partial ground plane, and 4) the slots whose design is based on the knowledge of fractal shapes. As a result, this antenna has an impedance bandwidth over the 2–11 GHz range, as shown in Fig. 4, and thus can operate in the bands used for UMTS, WLAN, WiMAX, and UWB applications. Consistent omnidirectional radiation patterns, and good gain and efficiency values characterize this UWB antenna. The radiation patterns are shown in Fig. 5.

**Figure 6.** Design with parametrized ground plane slot [14]

**Figure 7.** Reflection coefficient of the antenna in Fig. 6 for different *Rg* values

**4. Antennas with reconfigurable band rejection**

in the measured patterns of Fig. 10, taken at 4 GHz.

optimal designs are shown in Fig. 8.

The effect of changing the parameter *Rg* on the reflection coefficient is shown in Fig. 7. The results show that a slot of a specific size (*Rg* = 14.5 mm) results in a UWB response, so does a partial rectangular ground plane (corresponds to *Rg* = ∞). The configurations of these two

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(a) (b)

The measured and computed reflection coefficients of the two optimal designs are given in Fig. 9. Since the studied antenna is the type of a printed monopole, both optimal cases (with ground slot or partial rectangular ground) have omnidirectional radiation patterns, as shown

UWB technology is usually associated with the CR underlay mode [17]. It can, however, be implemented in the overlay mode. The difference between the two modes is the amount of transmitted power. In the underlay mode, UWB has a considerably restricted power, which is spread over a wide frequency band. In the overlay mode, however, the transmitted power can be much higher. It actually can be increased to a level that is comparable to the power

**Figure 5.** Patterns of the antenna in Fig. 3 in the X−Z plane (dotted line) and Y−Z plane (solid line) for (a) 2.1 GHz, (b) 2.4 GHz, (c) 3.5 GHz, and (d) 5.1 GHz

#### **3.2. Effect of ground plane**

The effect of the ground plane on the performance of UWB antennas is studied in [14–16]. The design in [14] is a coplanar-waveguide-fed antenna based on an egg-shaped conductor, and is taken as an example. The shape of the patch is suitable for UWB response. A large egg-shaped slot, with parametrized dimensions, was made in the ground, as shown in Fig. 6.

**Figure 6.** Design with parametrized ground plane slot [14]

6 Advancements in Microstrip and Printed Antennas

**Figure 4.** Reflection coefficient of the UWB antenna in Fig. 3

shown in Fig. 5.

GHz, (c) 3.5 GHz, and (d) 5.1 GHz

**3.2. Effect of ground plane**

Four techniques are applied for good impedance matching over the UWB range: 1) the specially selected patch shape, 2) the tapered connection between the patch and the feed line, 3) the optimized partial ground plane, and 4) the slots whose design is based on the knowledge of fractal shapes. As a result, this antenna has an impedance bandwidth over the 2–11 GHz range, as shown in Fig. 4, and thus can operate in the bands used for UMTS, WLAN, WiMAX, and UWB applications. Consistent omnidirectional radiation patterns, and good gain and efficiency values characterize this UWB antenna. The radiation patterns are

**Figure 5.** Patterns of the antenna in Fig. 3 in the X−Z plane (dotted line) and Y−Z plane (solid line) for (a) 2.1 GHz, (b) 2.4

The effect of the ground plane on the performance of UWB antennas is studied in [14–16]. The design in [14] is a coplanar-waveguide-fed antenna based on an egg-shaped conductor, and is taken as an example. The shape of the patch is suitable for UWB response. A large egg-shaped slot, with parametrized dimensions, was made in the ground, as shown in Fig. 6. The effect of changing the parameter *Rg* on the reflection coefficient is shown in Fig. 7. The results show that a slot of a specific size (*Rg* = 14.5 mm) results in a UWB response, so does a partial rectangular ground plane (corresponds to *Rg* = ∞). The configurations of these two optimal designs are shown in Fig. 8.

**Figure 7.** Reflection coefficient of the antenna in Fig. 6 for different *Rg* values

The measured and computed reflection coefficients of the two optimal designs are given in Fig. 9. Since the studied antenna is the type of a printed monopole, both optimal cases (with ground slot or partial rectangular ground) have omnidirectional radiation patterns, as shown in the measured patterns of Fig. 10, taken at 4 GHz.

#### **4. Antennas with reconfigurable band rejection**

UWB technology is usually associated with the CR underlay mode [17]. It can, however, be implemented in the overlay mode. The difference between the two modes is the amount of transmitted power. In the underlay mode, UWB has a considerably restricted power, which is spread over a wide frequency band. In the overlay mode, however, the transmitted power can be much higher. It actually can be increased to a level that is comparable to the power

**Figure 10.** Measured radiation patterns of Optimal Design I (left column) and Optimal Design II (right column) in the XZ-plane

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Antennas that allow the use of UWB in overlay CR are discussed in this Section. Several band-notching techniques are used in such antennas, the most famous of which being the use of split-ring resonators (SRRs) and the complementary split-ring resnators (CSRRs), which are discussed in 4.1. UWB antennas with fixed band notches are reported in [20–23]. Some UWB antennas with reconfigurable band notches, which are suitable for CR, are discussed

Split-ring resonators (SRRs), originally proposed by Pendry *et al.* [24], have attracted great interest among electromagneticians and microwave engineers due to their applications to the synthesis of artificial materials (metamaterials) with negative effective permeability. From duality arguments, it has been shown that negative permittivity media can also be generated by means of resonant elements, namely, complementary split-ring resonators (CSRRs) [25]. These particles are simply the negative image of SRRs, and roughly behave as their dual

The basic topologies of the SRR and the CSRR, and their equivalent-circuit models, are shown in Fig. 11. The equivalent-circuit models are reported in [26]. The SRR consists of two concentric metallic split rings printed on a microwave dielectric circuit board. The complementary of a planar metallic structure is obtained by replacing the metal parts of the original structure with apertures, and the apertures with metal plates. According to their lumped element models, SRRs and CSRRs do resonate. At the resonance frequency, SRRs have negative permeability, and CSRRs give negative permittivity, properties that lead to

Single-ring SRRs and CSRRs are discussed in [27], where the relationship between the SRR and CSRR dimensions and their resonance frequencies are studied by simulations. The band rejection they cause about their resonance frequency can be controlled by mounting electronic switches across them and activating/deactivating these switches. This is illustrated in the example of Fig. 12, where two configurations of a reconfigurable bandstop filter are shown. In both configurations, a rectangular single-ring CSRR is incorporated in a 50-Ω microstrip

line, which is printed on a 1.52mm-thick Rogers RO3203 substrate with *ε<sup>r</sup>* = 3.02.

(solid line) and the YZ-plane (dotted line) [14]

**4.1. SRRs and CSRRs**

below.

counterparts.

band rejection.

**Figure 8.** Two UWB antennas with optimized ground planes [14]

**Figure 9.** Measured and computed reflection coefficients of the designs in Fig. 8

of licensed systems, which allows for communication over medium to long distances. But this mode is only applicable if two conditions are met: 1) if the UWB transmitter ensures that the targeted spectrum is completely free of signals of other systems, or shapes its pulse to have nulls in the bands used by these systems, and 2) if the regulations are revised to allow this mode of operation [18]. Pulse adaptation for overlay UWB CR has been discussed in [19]. UWB can also operate in both underlay and overlay modes simultaneously. This can happen by shaping the transmitted signal so as to make part of the spectrum occupied in an underlay mode and some other parts occupied in an overlay mode. In the overlay UWB scenario, the antenna at the front-end of the CR device should be capable of operating over the whole UWB range, for sensing and determining the bands that are being used by primary users, but should also be able to induce band notches in its frequency response to prevent interference to these users. Even if the UWB power is not increased, having these band notches prevent raising the noise floor of primary users.

**Figure 10.** Measured radiation patterns of Optimal Design I (left column) and Optimal Design II (right column) in the XZ-plane (solid line) and the YZ-plane (dotted line) [14]

Antennas that allow the use of UWB in overlay CR are discussed in this Section. Several band-notching techniques are used in such antennas, the most famous of which being the use of split-ring resonators (SRRs) and the complementary split-ring resnators (CSRRs), which are discussed in 4.1. UWB antennas with fixed band notches are reported in [20–23]. Some UWB antennas with reconfigurable band notches, which are suitable for CR, are discussed below.

#### **4.1. SRRs and CSRRs**

8 Advancements in Microstrip and Printed Antennas

**Figure 8.** Two UWB antennas with optimized ground planes [14]

**Figure 9.** Measured and computed reflection coefficients of the designs in Fig. 8

band notches prevent raising the noise floor of primary users.

(a) Optimal Design I (b) Optimal Design II

of licensed systems, which allows for communication over medium to long distances. But this mode is only applicable if two conditions are met: 1) if the UWB transmitter ensures that the targeted spectrum is completely free of signals of other systems, or shapes its pulse to have nulls in the bands used by these systems, and 2) if the regulations are revised to allow this mode of operation [18]. Pulse adaptation for overlay UWB CR has been discussed in [19]. UWB can also operate in both underlay and overlay modes simultaneously. This can happen by shaping the transmitted signal so as to make part of the spectrum occupied in an underlay mode and some other parts occupied in an overlay mode. In the overlay UWB scenario, the antenna at the front-end of the CR device should be capable of operating over the whole UWB range, for sensing and determining the bands that are being used by primary users, but should also be able to induce band notches in its frequency response to prevent interference to these users. Even if the UWB power is not increased, having these Split-ring resonators (SRRs), originally proposed by Pendry *et al.* [24], have attracted great interest among electromagneticians and microwave engineers due to their applications to the synthesis of artificial materials (metamaterials) with negative effective permeability. From duality arguments, it has been shown that negative permittivity media can also be generated by means of resonant elements, namely, complementary split-ring resonators (CSRRs) [25]. These particles are simply the negative image of SRRs, and roughly behave as their dual counterparts.

The basic topologies of the SRR and the CSRR, and their equivalent-circuit models, are shown in Fig. 11. The equivalent-circuit models are reported in [26]. The SRR consists of two concentric metallic split rings printed on a microwave dielectric circuit board. The complementary of a planar metallic structure is obtained by replacing the metal parts of the original structure with apertures, and the apertures with metal plates. According to their lumped element models, SRRs and CSRRs do resonate. At the resonance frequency, SRRs have negative permeability, and CSRRs give negative permittivity, properties that lead to band rejection.

Single-ring SRRs and CSRRs are discussed in [27], where the relationship between the SRR and CSRR dimensions and their resonance frequencies are studied by simulations. The band rejection they cause about their resonance frequency can be controlled by mounting electronic switches across them and activating/deactivating these switches. This is illustrated in the example of Fig. 12, where two configurations of a reconfigurable bandstop filter are shown. In both configurations, a rectangular single-ring CSRR is incorporated in a 50-Ω microstrip line, which is printed on a 1.52mm-thick Rogers RO3203 substrate with *ε<sup>r</sup>* = 3.02.

In the first configuration, an electronic switch (PIN diode or RF MEMS) is mounted on one side of the ring slot, as shown. Setting the switch ON leads to a resonating CSRR, which creates a stop band around the resonance frequency. When the switch is OFF, we end up with a complete unsplit ring, which does not have the characteristics of a CSRR, and as a result the stop band is removed. In the second configuration, a hard connection (or a capacitor with high capacitance) is present on one side of the ring slot, and a switch is mounted on the opposite side. When the switch is OFF, we have a CSRR with one gap, resonating at a certain frequency, and when the switch is ON, the CSRR will have two gaps, thus resonating at the higher frequency. The dimensions of the ring slot are chosen such that the stop band (of the single-gap case) is centered at 3.5 GHz. These dimensions are the same for both configurations.

The computed reflection and transmission coefficients for the first configuration are shown in Fig. 13. For the switch-ON case, a stop band, centered at 3.5 GHz, is created (Fig. 13(a)). When the switch is OFF, the stop band disappears, and the all-pass behavior is retrieved (Fig. 13(b)). Fig. 14 plots the computed reflection and transmission coefficients for the second configuration. A stop band, centered at 3.5 GHz, results when the switch is OFF, as shown in Fig. 14(b). A narrower stop band is created at a higher frequency, 6.6 GHz, when the switch is ON (Fig. 14(a)). It is to note that theses result hold if the CSRR is instead incorporated in the ground plane below the microstrip line. Similar properties hold for SRRs, as reported in [27]. The switching components can be replaced with varactors, to obtain notch tunability.

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The design of a UWB antenna with a single switchable band rejection is reported in [28]. Two inverted T-shaped slits are embedded on the ground plane to allow band rejection characteristic from 5 to 6 GHz, and a PIN diode is connected to each slit to enable the

In [29], a wideband antenna with reconfigurable rejection within the operation band is presented. The antenna is a CPW-fed bow-tie, where a slot etched along the bow-tie upper edge provides the rejection of a certain band. Six PIN diodes mounted across the slots are used as switching elements, but only four switching cases are of use: one results in a notch-free wideband response, and the remaining three result each in a rejection in separate

A UWB design with a single reconfigurable band notch is proposed in [30]. The configuration of this design and a photo of its fabricated prototype are shown in Fig. 15. Originally, the antenna is a UWB monopole, printed on a 30 × 30 × 1.6 mm<sup>3</sup> Rogers RO3006 substrate with a dielectric constant *ε<sup>r</sup>* = 6.15. It has a microstrip line feed and a partial ground plane. The patch is rectangular and is 14 mm × 15.5 mm in size, the ground is 30 mm × 10 mm, and the feed line is 2.4 mm × 10.5 mm. For better matching, the corners of the patch are rounded, by intersecting it with a circle of radius 8.75 mm, and a slit is etched in the ground below the

**4.2. Antennas with a single reconfigurable rejection band**

switching capability for this band rejection function.

2 3 4 5 6 7 8 9 10

S11 S21

**Figure 13.** Reflection coefficient and transmission of CSRR-based filter in Fig. 12(a). (a) Switch is ON. (b) Switch is OFF.


S Parameters [dB]

2 3 4 5 6 7 8 9 10

S11 S21

Frequency [GHz]

(b) Switch OFF

Frequency [GHz]

(a) Switch ON

bands.


S Parameters [dB]

**Figure 11.** Topologies of the: (a) SRR and (b) CSRR, and their equivalent-circuit models. Grey zones represent the metallization. [26]

**Figure 12.** Reconfigurable bandstop filter based on a CSRR. (a) Configuration 1: electronic switch mounted over ring slot. (b) Configuration 2: hard connection on one side and an electronic switch over the CSRR slot on the other side.

The computed reflection and transmission coefficients for the first configuration are shown in Fig. 13. For the switch-ON case, a stop band, centered at 3.5 GHz, is created (Fig. 13(a)). When the switch is OFF, the stop band disappears, and the all-pass behavior is retrieved (Fig. 13(b)). Fig. 14 plots the computed reflection and transmission coefficients for the second configuration. A stop band, centered at 3.5 GHz, results when the switch is OFF, as shown in Fig. 14(b). A narrower stop band is created at a higher frequency, 6.6 GHz, when the switch is ON (Fig. 14(a)). It is to note that theses result hold if the CSRR is instead incorporated in the ground plane below the microstrip line. Similar properties hold for SRRs, as reported in [27]. The switching components can be replaced with varactors, to obtain notch tunability.

#### **4.2. Antennas with a single reconfigurable rejection band**

10 Advancements in Microstrip and Printed Antennas

configurations.

[26]

In the first configuration, an electronic switch (PIN diode or RF MEMS) is mounted on one side of the ring slot, as shown. Setting the switch ON leads to a resonating CSRR, which creates a stop band around the resonance frequency. When the switch is OFF, we end up with a complete unsplit ring, which does not have the characteristics of a CSRR, and as a result the stop band is removed. In the second configuration, a hard connection (or a capacitor with high capacitance) is present on one side of the ring slot, and a switch is mounted on the opposite side. When the switch is OFF, we have a CSRR with one gap, resonating at a certain frequency, and when the switch is ON, the CSRR will have two gaps, thus resonating at the higher frequency. The dimensions of the ring slot are chosen such that the stop band (of the single-gap case) is centered at 3.5 GHz. These dimensions are the same for both

**Figure 11.** Topologies of the: (a) SRR and (b) CSRR, and their equivalent-circuit models. Grey zones represent the metallization.

(a) Configuration 1 (b) Configuration 2

**Figure 12.** Reconfigurable bandstop filter based on a CSRR. (a) Configuration 1: electronic switch mounted over ring slot. (b)

Configuration 2: hard connection on one side and an electronic switch over the CSRR slot on the other side.

The design of a UWB antenna with a single switchable band rejection is reported in [28]. Two inverted T-shaped slits are embedded on the ground plane to allow band rejection characteristic from 5 to 6 GHz, and a PIN diode is connected to each slit to enable the switching capability for this band rejection function.

In [29], a wideband antenna with reconfigurable rejection within the operation band is presented. The antenna is a CPW-fed bow-tie, where a slot etched along the bow-tie upper edge provides the rejection of a certain band. Six PIN diodes mounted across the slots are used as switching elements, but only four switching cases are of use: one results in a notch-free wideband response, and the remaining three result each in a rejection in separate bands.

A UWB design with a single reconfigurable band notch is proposed in [30]. The configuration of this design and a photo of its fabricated prototype are shown in Fig. 15. Originally, the antenna is a UWB monopole, printed on a 30 × 30 × 1.6 mm<sup>3</sup> Rogers RO3006 substrate with a dielectric constant *ε<sup>r</sup>* = 6.15. It has a microstrip line feed and a partial ground plane. The patch is rectangular and is 14 mm × 15.5 mm in size, the ground is 30 mm × 10 mm, and the feed line is 2.4 mm × 10.5 mm. For better matching, the corners of the patch are rounded, by intersecting it with a circle of radius 8.75 mm, and a slit is etched in the ground below the

**Figure 13.** Reflection coefficient and transmission of CSRR-based filter in Fig. 12(a). (a) Switch is ON. (b) Switch is OFF.

**Figure 14.** Reflection coefficient and transmission of CSRR-based filter in Fig. 12(b). (a) Switch is OFF. (b) Switch is ON.

(a) Computed (b) Measured

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The antenna reported in [31] is capable of inducing three band notches, which are independently controllable, using only three RF switches. Illustrated in Fig. 17, the antenna is a monopole printed on a 1.6-mm-thick Taconic TLY substrate with *ε<sup>r</sup>* = 2.2, and features a partial rectangular ground plane. The patch is rectangular in shape, but the corners of the rectangle around the feed line are rounded to create a matching section. The dimensions of the different parts are optimized for an impedance bandwidth covering the 2–11 GHz range.

(a) (b)

**Figure 17.** (a) Configuration of a UWB antenna with three independently reconfigurable band notches, and (b) photo of its

To create the band notches, two rectangular and one elliptical CSRRs are etched on the patch. Their shapes are selected to suit the part of the patch they are fitted in. Their sizes

prototype [31]

**Figure 16.** Computed and measured reflection coefficient for the different switching cases of the antenna in Fig. 15

**4.3. Antennas with multiple reconfigurable rejection bands**

**Figure 15.** Configuration and photo of an antenna with one reconfigurable rejection band [30]

feed. The slit is 3 mm × 1 mm. As a result, this antenna has an impedance bandwidth that covers the whole UWB frequency range. Four nested CSRRs are incorporated in the patch. Three electronic switches, 1mm×0.5mm in size, are mounted across the slots. The sequential activation (deactivation) of the switches leads to the functioning of a larger (smaller) CSRR, and thus results in a notch at a lower (higher) frequency. The following switching cases are considered: Case 1 when all three switches are ON, Case 2 when only S3 is deactivated, Case 3 when only S1 is ON, and finally Case 4 when all switches are OFF. The resulting reflection coefficient plots, corresponding to the different switching states, are shown in Fig. 16. The plots show one notch, which can occur in one of 3 bands, or can completely disappear. In the latter case, the antenna retrieves its UWB response, which enables it to sense the whole UWB range.

**Figure 16.** Computed and measured reflection coefficient for the different switching cases of the antenna in Fig. 15

#### **4.3. Antennas with multiple reconfigurable rejection bands**

12 Advancements in Microstrip and Printed Antennas

2 3 4 5 6 7 8 9 10

S11 S21

**Figure 14.** Reflection coefficient and transmission of CSRR-based filter in Fig. 12(b). (a) Switch is OFF. (b) Switch is ON.

(a) (b)

feed. The slit is 3 mm × 1 mm. As a result, this antenna has an impedance bandwidth that covers the whole UWB frequency range. Four nested CSRRs are incorporated in the patch. Three electronic switches, 1mm×0.5mm in size, are mounted across the slots. The sequential activation (deactivation) of the switches leads to the functioning of a larger (smaller) CSRR, and thus results in a notch at a lower (higher) frequency. The following switching cases are considered: Case 1 when all three switches are ON, Case 2 when only S3 is deactivated, Case 3 when only S1 is ON, and finally Case 4 when all switches are OFF. The resulting reflection coefficient plots, corresponding to the different switching states, are shown in Fig. 16. The plots show one notch, which can occur in one of 3 bands, or can completely disappear. In the latter case, the antenna retrieves its UWB response, which enables it to sense the whole

**Figure 15.** Configuration and photo of an antenna with one reconfigurable rejection band [30]


S Parameters [dB]

2 3 4 5 6 7 8 9 10

S11 S21

Frequency [GHz]

(b) Switch ON

Frequency [GHz]

(a) Switch OFF


UWB range.

S Parameters [dB]

The antenna reported in [31] is capable of inducing three band notches, which are independently controllable, using only three RF switches. Illustrated in Fig. 17, the antenna is a monopole printed on a 1.6-mm-thick Taconic TLY substrate with *ε<sup>r</sup>* = 2.2, and features a partial rectangular ground plane. The patch is rectangular in shape, but the corners of the rectangle around the feed line are rounded to create a matching section. The dimensions of the different parts are optimized for an impedance bandwidth covering the 2–11 GHz range.

**Figure 17.** (a) Configuration of a UWB antenna with three independently reconfigurable band notches, and (b) photo of its prototype [31]

To create the band notches, two rectangular and one elliptical CSRRs are etched on the patch. Their shapes are selected to suit the part of the patch they are fitted in. Their sizes


**Table 1.** The 8 switching cases for the design in Fig. 17 and the corresponding notched bands.

are optimized so that the larger rectangular CSRR causes a notch in the 2.4 GHz band, the smaller one in the 3.5 GHz band, and the elliptical one in the 5.2 GHz band. To enable band notch reconfigurability, three electronic switches (S1, S2, and S3) are mounted across the CSRRs.

(a) (b)

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**Figure 19.** (a) Configuration of a filter antenna with two reconfigurable band notches, and (b) photo of its prototype [32]

reflections at the antenna's port.

simpler to design.

cases.

**4.4. Filter antennas with reconfigurable band notches**

The antenna has omnidirectional radiation patterns. It also has good gain values in its band(s) of operation. In a notched band, the gain drops to negative values due to strong

A UWB antenna with reconfigurable band notches can be designed by incorporating a bandstop filter in the feed line of a UWB antenna. With this structure, the switching elements will be mounted on the feed line, away from the radiating patch, which makes the bias circuit

(a) (b)

**Figure 20.** (a) Simulated and (b) measured reflection coefficient plots for the antenna in Fig. 19 for the four adopted switching

A filter antenna with two reconfigurable rejection bands is presented in [32]. Its structure is shown in Fig. 19. The UWB antenna is based on a rounded patch and a partial rectangular

The state of a switch controls the notch causing by the corresponding CSRR. When S1 is OFF, the elliptical CSRR induces a notch in the 5.2 GHz band. When S2 is OFF, the large rectangular CSRR causes a notch in the 2.4 GHz band. For the smaller rectangular CSRR, a notch appears at 3.5 GHz when S3 is OFF. When a switch is ON, the corresponding CSRR behaves as one with two gaps, and its resonance moves up in frequency and becomes too weak to affect the UWB response of the antenna. The different switching cases lead to different band notch combinations. These include the scenarios of one, two, three concurrent notches, or no notch at all. In the latter case, the antenna has a UWB response, which is required for channel sensing.

There are eight possible switching scenarios for this antenna, which are listed in Table 1. Fig. 18 shows the computed and measured reflection coefficient plots for some of the switching cases. For Case 1, a UWB response is obtained. The results for Case 2 reveal a single notch in the 2.4 GHz band, those for Case 4 show a single notch in the 5.2 GHz band, the results for Case 5 show two notches in the 2.4 and 3.5 GHz bands, and those for Case 8 show three notches in the 2.4, 3.5, and 5.2 GHz bands. A notch in a certain band helps to prevent interference to a primary user or the service operated in that band.

**Figure 18.** (a) Simulated and (b) measured reflection coefficient plots for the antenna in Fig. 17 for some of the adopted switching cases

**Figure 19.** (a) Configuration of a filter antenna with two reconfigurable band notches, and (b) photo of its prototype [32]

The antenna has omnidirectional radiation patterns. It also has good gain values in its band(s) of operation. In a notched band, the gain drops to negative values due to strong reflections at the antenna's port.

#### **4.4. Filter antennas with reconfigurable band notches**

14 Advancements in Microstrip and Printed Antennas

the CSRRs.

switching cases

required for channel sensing.

**Case Notch bands (GHz) S1 S2 S3** 1 None (UWB operation) ON ON ON 2.4 ON OFF ON 3.5 ON ON OFF 5.2 OFF ON ON 2.4, 3.5 ON OFF OFF 2.4, 5.2 OFF OFF ON 3.5, 5.2 OFF ON OFF 8 2.4, 3.5, 5.2 OFF OFF OFF

are optimized so that the larger rectangular CSRR causes a notch in the 2.4 GHz band, the smaller one in the 3.5 GHz band, and the elliptical one in the 5.2 GHz band. To enable band notch reconfigurability, three electronic switches (S1, S2, and S3) are mounted across

The state of a switch controls the notch causing by the corresponding CSRR. When S1 is OFF, the elliptical CSRR induces a notch in the 5.2 GHz band. When S2 is OFF, the large rectangular CSRR causes a notch in the 2.4 GHz band. For the smaller rectangular CSRR, a notch appears at 3.5 GHz when S3 is OFF. When a switch is ON, the corresponding CSRR behaves as one with two gaps, and its resonance moves up in frequency and becomes too weak to affect the UWB response of the antenna. The different switching cases lead to different band notch combinations. These include the scenarios of one, two, three concurrent notches, or no notch at all. In the latter case, the antenna has a UWB response, which is

There are eight possible switching scenarios for this antenna, which are listed in Table 1. Fig. 18 shows the computed and measured reflection coefficient plots for some of the switching cases. For Case 1, a UWB response is obtained. The results for Case 2 reveal a single notch in the 2.4 GHz band, those for Case 4 show a single notch in the 5.2 GHz band, the results for Case 5 show two notches in the 2.4 and 3.5 GHz bands, and those for Case 8 show three notches in the 2.4, 3.5, and 5.2 GHz bands. A notch in a certain band helps to prevent

(a) (b)

**Figure 18.** (a) Simulated and (b) measured reflection coefficient plots for the antenna in Fig. 17 for some of the adopted

**Table 1.** The 8 switching cases for the design in Fig. 17 and the corresponding notched bands.

interference to a primary user or the service operated in that band.

A UWB antenna with reconfigurable band notches can be designed by incorporating a bandstop filter in the feed line of a UWB antenna. With this structure, the switching elements will be mounted on the feed line, away from the radiating patch, which makes the bias circuit simpler to design.

**Figure 20.** (a) Simulated and (b) measured reflection coefficient plots for the antenna in Fig. 19 for the four adopted switching cases.

A filter antenna with two reconfigurable rejection bands is presented in [32]. Its structure is shown in Fig. 19. The UWB antenna is based on a rounded patch and a partial rectangular

**Figure 22.** Realized peak gain of the antenna in Fig. 19. Case 1: gain is positive. Case 4: gain is negative in the 3.5 and 5.5

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Antennas designed for overlay CR should have the capability to sense the channel and communicate over a small portion of it. These antennas can be implemented as dual-port, where one port is UWB, and the other is narrowband and frequency reconfigurable. The design of UWB antennas was discussed in Section 3. They can also be designed as single-port, where the same port is used for both sensing and communicating, and thus should switch between wideband and narrowband operations. The advantages of each design will be

A dual-port antenna for overlay was proposed by Ebrahimi *et al.* [33, 34]. The structure consists of two printed antennas namely a wide- and a narrow-band antenna. Because the two antennas are in close proximity, high coupling exists between them, and the patterns of the NB antenna are affected by the presence of the UWB one. The authors successfully

A simpler design that offers good isolation between the two antenna ports is presented in [35]. The configuration of this design, which comprises two microstrip-line-fed monopoles sharing a common partial ground, is shown in Fig. 23. The sensing UWB antenna is based on an egg-shaped patch, obtained by combining a circle and an ellipse at their centers. A small tapered microstrip section is used to match the 50-Ω feed to the input impedance of the patch. The UWB response of the sensing antenna is guaranteed by the design of the patch, the partial ground plane, and the feed matching section. The reflection coefficient of

The communicating antenna is a microstrip line connected to a 50-Ω feed line via a matching section. Two electronic switches are incorporated along this line. By controlling the switches, the length of the antenna is changed, which leads to various resonance frequencies inside the UWB range. Three switching cases are considered: Case 1 where both switches are deactivated, Case 2 where Switch 1 is ON and Switch 2 is OFF, and Case 3 where both

designed modified versions of the antenna system to solve the coupling issue.

the sensing antenna, and its normalized 5-GHz patterns, are shown in Fig. 24.

GHz bands.

**5. Antennas for overlay CR**

revealed in the rest of this Section.

**5.1. Dual-port antennas for overlay CR**

**Figure 21.** Radiation patterns of the antenna in Fig. 19 in the H-plane (dotted line) and E-plane (solid line)

ground plane. The Rogers RO3203 material is used for the 1.52mm-thick substrate. A reconfigurable filter with two stop bands is incorporated along its microstrip feed line. The filter is based on one rectangular single-ring CSRR etched on the line, and two identical rectangular single-ring SRRs placed in close proximity to it. The resonance of the CSRR is controlled via a switch, and that of the two SRRs via two switches that are operated in parallel. As a result, there are four switching scenarios. The simulated and measured *S*<sup>11</sup> plots are shown in Fig. 20. Case 1, where no band notches exist, allows the antenna to sense the UWB range to determine the narrowband primary services that are transmitting inside the range. In the other three cases, the notches block the UWB pulse components in the 3.5 GHz band, the 5.5 GHz band, or both. It should be noted that notches due to the SRRs and the CSRR around the feed are stronger than those due to CSRRs or any notching structures implemented in the patch. This is because energy is concentrated in a smaller area in the feed, and coupling with the SRRs/CSRR is higher.

The normalized gain patterns of the filter antenna, for Case1, are shown in Fig. 21. The antenna has good omnidirectional patterns, and this is expected since its is a printed monopole with a small ground plane not covering the radiating patch. Since the patch is untouched, the patterns are independent of the switching cases. The realized peak gain of the antenna is plotted in Fig. 22 for Case 1 and Case 4. In Case 4, the gain drops sharply, to below -10 dB, at 3.5 GHz and at 5.5 GHz. In these two bands, very high reflections occur at the antenna's input. The gain drop in the notch band is large, as the coupling the CSRR and the SRRs cause is high. Due to the location of the switches, connecting the DC bias lines, especially to the SRRs, which are DC-separated from anything else, is an easy task. A wire can be used to drive the switch on the CSRR. A note is that extra band notches can be obtained by placing more SRRs around the feed line.

**Figure 22.** Realized peak gain of the antenna in Fig. 19. Case 1: gain is positive. Case 4: gain is negative in the 3.5 and 5.5 GHz bands.

#### **5. Antennas for overlay CR**

16 Advancements in Microstrip and Printed Antennas

**Figure 21.** Radiation patterns of the antenna in Fig. 19 in the H-plane (dotted line) and E-plane (solid line)

feed, and coupling with the SRRs/CSRR is higher.

obtained by placing more SRRs around the feed line.

ground plane. The Rogers RO3203 material is used for the 1.52mm-thick substrate. A reconfigurable filter with two stop bands is incorporated along its microstrip feed line. The filter is based on one rectangular single-ring CSRR etched on the line, and two identical rectangular single-ring SRRs placed in close proximity to it. The resonance of the CSRR is controlled via a switch, and that of the two SRRs via two switches that are operated in parallel. As a result, there are four switching scenarios. The simulated and measured *S*<sup>11</sup> plots are shown in Fig. 20. Case 1, where no band notches exist, allows the antenna to sense the UWB range to determine the narrowband primary services that are transmitting inside the range. In the other three cases, the notches block the UWB pulse components in the 3.5 GHz band, the 5.5 GHz band, or both. It should be noted that notches due to the SRRs and the CSRR around the feed are stronger than those due to CSRRs or any notching structures implemented in the patch. This is because energy is concentrated in a smaller area in the

The normalized gain patterns of the filter antenna, for Case1, are shown in Fig. 21. The antenna has good omnidirectional patterns, and this is expected since its is a printed monopole with a small ground plane not covering the radiating patch. Since the patch is untouched, the patterns are independent of the switching cases. The realized peak gain of the antenna is plotted in Fig. 22 for Case 1 and Case 4. In Case 4, the gain drops sharply, to below -10 dB, at 3.5 GHz and at 5.5 GHz. In these two bands, very high reflections occur at the antenna's input. The gain drop in the notch band is large, as the coupling the CSRR and the SRRs cause is high. Due to the location of the switches, connecting the DC bias lines, especially to the SRRs, which are DC-separated from anything else, is an easy task. A wire can be used to drive the switch on the CSRR. A note is that extra band notches can be Antennas designed for overlay CR should have the capability to sense the channel and communicate over a small portion of it. These antennas can be implemented as dual-port, where one port is UWB, and the other is narrowband and frequency reconfigurable. The design of UWB antennas was discussed in Section 3. They can also be designed as single-port, where the same port is used for both sensing and communicating, and thus should switch between wideband and narrowband operations. The advantages of each design will be revealed in the rest of this Section.

#### **5.1. Dual-port antennas for overlay CR**

A dual-port antenna for overlay was proposed by Ebrahimi *et al.* [33, 34]. The structure consists of two printed antennas namely a wide- and a narrow-band antenna. Because the two antennas are in close proximity, high coupling exists between them, and the patterns of the NB antenna are affected by the presence of the UWB one. The authors successfully designed modified versions of the antenna system to solve the coupling issue.

A simpler design that offers good isolation between the two antenna ports is presented in [35]. The configuration of this design, which comprises two microstrip-line-fed monopoles sharing a common partial ground, is shown in Fig. 23. The sensing UWB antenna is based on an egg-shaped patch, obtained by combining a circle and an ellipse at their centers. A small tapered microstrip section is used to match the 50-Ω feed to the input impedance of the patch. The UWB response of the sensing antenna is guaranteed by the design of the patch, the partial ground plane, and the feed matching section. The reflection coefficient of the sensing antenna, and its normalized 5-GHz patterns, are shown in Fig. 24.

The communicating antenna is a microstrip line connected to a 50-Ω feed line via a matching section. Two electronic switches are incorporated along this line. By controlling the switches, the length of the antenna is changed, which leads to various resonance frequencies inside the UWB range. Three switching cases are considered: Case 1 where both switches are deactivated, Case 2 where Switch 1 is ON and Switch 2 is OFF, and Case 3 where both

**Figure 23.** Dual-port UWB-NB antenna for overlay CR (a) configuration and (b) photo of a fabricated prototype [35]

**Figure 25.** Measured reflection coefficient of the communicating antenna [35]

and in very wide band mode when both patches are excited.

circuits have little interference to the antenna performance.

due to the presence of the UWB patch.

to design.

**5.2. Single-port antennas for overlay CR**

The communicating antenna has also omnidirectional patterns, but some degradation occurs

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Dual-port antennas enable simultaneous sensing and communicating over the channel, but have limitations in terms of their relatively large size, the coupling between the two ports, and the degraded patterns. These limitations are solved by the use of single-port antennas, but these are only suitable when the channel does not change very fast, and thus sensing and communication are possible, sequentially. Single-port CR antennas are also more challenging

A reconfigurable wideband/dual-band double C-slot microstrip patch antenna is proposed in [36]. The frequency tuning is performed by switching ON and OFF two patches. The antenna operates in one of two different dual-band modes when either patch is activated,

In [37], a single-port Vivaldi antenna with added switched band functionality to operate in a wideband or narrowband mode is presented. Frequency reconfigurability in this design is attained by inserting four pairs of ring slots into the structure, and switching them using PIN diodes. A wide bandwidth mode covering the 1.0–3.2 GHz range, and three narrowband modes within this range, can be selected. A single pair of ring slots, and fifteen PIN diode switches across, are used on the single-port Vivaldi design in [38]. For this antenna, a wideband operation is obtained over the 1–3 GHz band, inside which there are six narrowband states of use. In these two Vivaldi antenna designs, the switching elements, PIN diodes in this case, are mounted on the radiating parts of the antennas. This makes the design of the DC bias circuits a complex task, as the designers have to make sure these

**Figure 24.** (a) Reflection coefficient of the sensing UWB antenna, and (b) its normalized gain patterns at 5 GHz: H-plane (solid line) and the E-plane (dashed line) [35]

switches are activated. The resulting measured reflection coefficient plots are given in Fig. 25, which shows clear frequency reconfigurability and coverage of most of the UWB range.

The transmission *S*<sup>21</sup> at the resonance frequencies, for the three switching cases, are given in Table 2. Good isolation between the UWB and NB port is achieved, given the simplicity of the design.


**Table 2.** Transmission *S*<sup>21</sup> for the design in [35]

**Figure 25.** Measured reflection coefficient of the communicating antenna [35]

The communicating antenna has also omnidirectional patterns, but some degradation occurs due to the presence of the UWB patch.

#### **5.2. Single-port antennas for overlay CR**

18 Advancements in Microstrip and Printed Antennas

line) and the E-plane (dashed line) [35]

**Table 2.** Transmission *S*<sup>21</sup> for the design in [35]

the design.

(a) (b)

(a) (b)

**Case 1 Case 2 Case 3**

*f* (GHz) 5.55 9.15 4.85 8.15 4.44 7.41 10.33 *S*<sup>21</sup> (dB) -31.4 -14.8 -16.8 -15.5 -21.3 -22.6 -15.7

**Figure 24.** (a) Reflection coefficient of the sensing UWB antenna, and (b) its normalized gain patterns at 5 GHz: H-plane (solid

switches are activated. The resulting measured reflection coefficient plots are given in Fig. 25, which shows clear frequency reconfigurability and coverage of most of the UWB range. The transmission *S*<sup>21</sup> at the resonance frequencies, for the three switching cases, are given in Table 2. Good isolation between the UWB and NB port is achieved, given the simplicity of

**Figure 23.** Dual-port UWB-NB antenna for overlay CR (a) configuration and (b) photo of a fabricated prototype [35]

Dual-port antennas enable simultaneous sensing and communicating over the channel, but have limitations in terms of their relatively large size, the coupling between the two ports, and the degraded patterns. These limitations are solved by the use of single-port antennas, but these are only suitable when the channel does not change very fast, and thus sensing and communication are possible, sequentially. Single-port CR antennas are also more challenging to design.

A reconfigurable wideband/dual-band double C-slot microstrip patch antenna is proposed in [36]. The frequency tuning is performed by switching ON and OFF two patches. The antenna operates in one of two different dual-band modes when either patch is activated, and in very wide band mode when both patches are excited.

In [37], a single-port Vivaldi antenna with added switched band functionality to operate in a wideband or narrowband mode is presented. Frequency reconfigurability in this design is attained by inserting four pairs of ring slots into the structure, and switching them using PIN diodes. A wide bandwidth mode covering the 1.0–3.2 GHz range, and three narrowband modes within this range, can be selected. A single pair of ring slots, and fifteen PIN diode switches across, are used on the single-port Vivaldi design in [38]. For this antenna, a wideband operation is obtained over the 1–3 GHz band, inside which there are six narrowband states of use. In these two Vivaldi antenna designs, the switching elements, PIN diodes in this case, are mounted on the radiating parts of the antennas. This makes the design of the DC bias circuits a complex task, as the designers have to make sure these circuits have little interference to the antenna performance.

The single-port overlay CR antenna in [39] has the switching elements mounted along its microstrip feed line, away from the radiating patch. This property has the advantage that the DC bias circuit causes limited interference to the antenna characteristics. The antenna is initially UWB, which makes it sensing-capable. A reconfigurable bandpass filter is then embedded along its feed line. When activated, the filter can transform the UWB frequency response into a reconfigurable narrowband one, which is suitable for the communication operation of the CR system. The configuration of the antenna, and a closer view of its embedded filter part, are shown in Fig. 26. It features a partial rectangular ground plane, a rectangular patch, and a curved matching section between the microstrip feed line and the patch. The filter is based on a symmetrical defected microstrip structure (DMS) implemented in the feed line of the UWB antenna. It has a T-shaped slot, which by itself, has bandstop characteristics. However, when placed between a pair of gaps, which act as capacitors, a bandpass structure results [40].

**Case Switches in OFF state** 0 None (all ON) S0, S1, S6 S0, S1, S5 S0, S2, S5 S0, S3, S5 S0, S3, S4

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(a) (b)

Reconfigurable NB antennas can operate over a limited number of bands inside a designated frequency range. Tunable antennas, on the other hands, can be reconfigured to resonate at theoretically an infinite number of frequencies within a certain band. Tunable antennas can be used for communicating in a CR system, by tuning to a white space. They can also be employed for channel sensing, by progressively scanning small portions of the band. This depends of course on the rate at which the channel is changing, and on the tuning speed. A tunable filter antenna is shown in Fig. 28. The initially UWB design features a tunable bandpass filter embedded along its microstrip feed line. It has a rounded patch and a partial rectangular ground plane. The Rogers RO3203 material, with *ε<sup>r</sup>* = 3.02, is used for the 1.52mm-thick substrate. The filter is based on a T-shaped slot incorporated in the microstrip

For the purpose of achieving frequency tunability, a varactor is included in the design, as indicated. Changing the capacitance of the varactor changes the notch band caused by the T-slot, and as a result the narrow pass band of the overall filter. The DC lines of the varactors are connected with ease. Due to the presence of the two gaps, DC is separated from both the antenna port and the patch. Two surface-mount inductors are used over the DC lines as RF chokes. The reflection coefficient plots, obtained in Ansoft HFSS, are given in Fig. 29. They show narrowband tunability over the 4.5–7 GHz frequency range, for capacitance values

**Table 3.** The six adopted switching cases for the single-port design in [39]

**Figure 27.** (a) Simulated, and (b) measured reflection coefficient [39]

line between a pair of gaps.

between 0.3 and 7 pF.

**5.3. A Single-port tunable filter antenna for overlay CR**

**Figure 26.** A reconfigurable UWB/NB filter antenna. (a) Configuration, and (b) closer view of the embedded filter [39]

For the purpose of achieving frequency reconfigurability, three pairs of gaps are symmetrically placed around the T-slot, and seven electronic switches are placed across the slots as shown. Six switching cases are considered, as indicated in Table 3. Case 0 corresponds to all the switches being ON. In this case, the effect of the filter is canceled, bringing back the UWB response of the antenna. The frequency characteristics of the filter depend on the dimensions of the slots, and on the switching state.

The computed and measured reflection coefficient plots for the six switching cases are given in Fig. 27. The operation of the antenna makes it suitable for employment in cognitive radio applications, where Case 0 could be used for sensing the channel (to determine the white spaces), and the other cases for communicating in the corresponding white space. Further resonances can be obtained by including more gaps around the T-slot and appropriately choosing their locations and widths.


**Table 3.** The six adopted switching cases for the single-port design in [39]

20 Advancements in Microstrip and Printed Antennas

bandpass structure results [40].

The single-port overlay CR antenna in [39] has the switching elements mounted along its microstrip feed line, away from the radiating patch. This property has the advantage that the DC bias circuit causes limited interference to the antenna characteristics. The antenna is initially UWB, which makes it sensing-capable. A reconfigurable bandpass filter is then embedded along its feed line. When activated, the filter can transform the UWB frequency response into a reconfigurable narrowband one, which is suitable for the communication operation of the CR system. The configuration of the antenna, and a closer view of its embedded filter part, are shown in Fig. 26. It features a partial rectangular ground plane, a rectangular patch, and a curved matching section between the microstrip feed line and the patch. The filter is based on a symmetrical defected microstrip structure (DMS) implemented in the feed line of the UWB antenna. It has a T-shaped slot, which by itself, has bandstop characteristics. However, when placed between a pair of gaps, which act as capacitors, a

(a) (b)

**Figure 26.** A reconfigurable UWB/NB filter antenna. (a) Configuration, and (b) closer view of the embedded filter [39]

depend on the dimensions of the slots, and on the switching state.

choosing their locations and widths.

For the purpose of achieving frequency reconfigurability, three pairs of gaps are symmetrically placed around the T-slot, and seven electronic switches are placed across the slots as shown. Six switching cases are considered, as indicated in Table 3. Case 0 corresponds to all the switches being ON. In this case, the effect of the filter is canceled, bringing back the UWB response of the antenna. The frequency characteristics of the filter

The computed and measured reflection coefficient plots for the six switching cases are given in Fig. 27. The operation of the antenna makes it suitable for employment in cognitive radio applications, where Case 0 could be used for sensing the channel (to determine the white spaces), and the other cases for communicating in the corresponding white space. Further resonances can be obtained by including more gaps around the T-slot and appropriately

**Figure 27.** (a) Simulated, and (b) measured reflection coefficient [39]

#### **5.3. A Single-port tunable filter antenna for overlay CR**

Reconfigurable NB antennas can operate over a limited number of bands inside a designated frequency range. Tunable antennas, on the other hands, can be reconfigured to resonate at theoretically an infinite number of frequencies within a certain band. Tunable antennas can be used for communicating in a CR system, by tuning to a white space. They can also be employed for channel sensing, by progressively scanning small portions of the band. This depends of course on the rate at which the channel is changing, and on the tuning speed.

A tunable filter antenna is shown in Fig. 28. The initially UWB design features a tunable bandpass filter embedded along its microstrip feed line. It has a rounded patch and a partial rectangular ground plane. The Rogers RO3203 material, with *ε<sup>r</sup>* = 3.02, is used for the 1.52mm-thick substrate. The filter is based on a T-shaped slot incorporated in the microstrip line between a pair of gaps.

For the purpose of achieving frequency tunability, a varactor is included in the design, as indicated. Changing the capacitance of the varactor changes the notch band caused by the T-slot, and as a result the narrow pass band of the overall filter. The DC lines of the varactors are connected with ease. Due to the presence of the two gaps, DC is separated from both the antenna port and the patch. Two surface-mount inductors are used over the DC lines as RF chokes. The reflection coefficient plots, obtained in Ansoft HFSS, are given in Fig. 29. They show narrowband tunability over the 4.5–7 GHz frequency range, for capacitance values between 0.3 and 7 pF.

control the switches, but they will be installed away from the radiating patch. This makes

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This chapter has discussed the design of antennas for Cognitive Radio applications. CR is a revolutionary spectrum allocation technology that allows unlicensed users to access spectrum bands licensed to primary users, at the condition of avoiding interference to them. Spectrum underlay and spectrum overlay are two approaches to sharing spectrum between primary

UWB antennas are required for sensing in overlay CR, and for communicating in underlay CR. Modified UWB antennas with reconfigurable band notches allow to employ UWB technology in overlay CR and to achieve high-data-rate and long distances communications. Overlay CR requires reconfigurable wideband/narrowband antennas, to perform the two tasks of sensing a wide band and communicating over a narrow white space. UWB antennas, antennas with reconfigurable band rejections, and single-port/dual-port wide-narrowband

[1] Federal Communications Commission (2002) Spectrum Policy Task Force Report,

[2] Chen K.-C, Prasad R. (2009) Cognitive Radio Networks. ISBN 978-0-470-69689-7. West

[3] Zhao Q, Sadler B.M (2007) A Survey of Dynamic Spectrum Access: Signal Processing, Networking, and Regulatory Policy. IEEE signal processing magazine. 24:3:79–89.

[4] Low Z.N, Cheong J.H, Law, C.L (2005) Low-cost PCB Antenna for UWB Applications.

[5] Mehdipour A, Mohammadpour-Aghdam K, Faraji-Dana R, Kashani-Khatib M.-R (2008) A Novel Coplanar Waveguide-fed Slot Antenna for Ultra- wideband Applications. IEEE

[6] Oraizi H, Hedayati S (2011) Miniaturized UWB Monopole Microstrip Antenna Design by the Combination of Giusepe Peano and Sierpinski Carpet Fractals. IEEE antennas

and tunable antennas suitable for these approaches have been reported.

their design relatively simple.

**6. Summary**

and secondary users.

**Author details**

**References**

Mohammed Al-Husseini1, Karim Y. Kabalan1, Ali El-Hajj<sup>1</sup> and Christos G. Christodoulou2 1 American University of Beirut, Lebanon

Sussex, United Kingdom: John Wiley & Sons.

and wireless propagation letters. 10:67–70.

IEEE antennas and wireless propagation letters. 4:237–239.

transactions on antennas and propagation. 56:12:3857–3862.

2 University of New Mexico, USA

Technical Report.

**Figure 28.** (a) Configuration of a tunable filter antenna, and (b) photo of a fabricated prototype. This antenna has a tunable bandpass filter embedded along its feed line.

**Figure 29.** Reflection coefficient of the tunable filter antenna. Narrowband tunability is achieved

A UWB operation of the antenna can be made possible by installing 3 switching elements (e.g. PIN diodes) across the T-shaped slot and the two gaps. When these switches are ON, the effect of the narrowband bandpass filter is canceled, and the original UWB response of the antenna is retrieved. Tunablility would still be possible by putting the switches to the OFF state and adjusting the varactor capacitance. Extra DC biasing lines are required to control the switches, but they will be installed away from the radiating patch. This makes their design relatively simple.

### **6. Summary**

22 Advancements in Microstrip and Printed Antennas

bandpass filter embedded along its feed line.





Reflection Coefficient [dB]


0

(a) (b)

C = 0.30 pF C = 0.35 pF C = 0.40 pF C = 0.50 pF C = 0.60 pF C = 0.70 pF C = 0.80 pF C = 1.00 pF C = 1.50 pF C = 2.20 pF C = 4.00 pF C = 7.00 pF

**Figure 28.** (a) Configuration of a tunable filter antenna, and (b) photo of a fabricated prototype. This antenna has a tunable

4 5 6 7 8

Frequency [GHz]

A UWB operation of the antenna can be made possible by installing 3 switching elements (e.g. PIN diodes) across the T-shaped slot and the two gaps. When these switches are ON, the effect of the narrowband bandpass filter is canceled, and the original UWB response of the antenna is retrieved. Tunablility would still be possible by putting the switches to the OFF state and adjusting the varactor capacitance. Extra DC biasing lines are required to

**Figure 29.** Reflection coefficient of the tunable filter antenna. Narrowband tunability is achieved

This chapter has discussed the design of antennas for Cognitive Radio applications. CR is a revolutionary spectrum allocation technology that allows unlicensed users to access spectrum bands licensed to primary users, at the condition of avoiding interference to them. Spectrum underlay and spectrum overlay are two approaches to sharing spectrum between primary and secondary users.

UWB antennas are required for sensing in overlay CR, and for communicating in underlay CR. Modified UWB antennas with reconfigurable band notches allow to employ UWB technology in overlay CR and to achieve high-data-rate and long distances communications. Overlay CR requires reconfigurable wideband/narrowband antennas, to perform the two tasks of sensing a wide band and communicating over a narrow white space. UWB antennas, antennas with reconfigurable band rejections, and single-port/dual-port wide-narrowband and tunable antennas suitable for these approaches have been reported.

### **Author details**

Mohammed Al-Husseini1, Karim Y. Kabalan1, Ali El-Hajj1 and Christos G. Christodoulou2

1 American University of Beirut, Lebanon

2 University of New Mexico, USA

#### **References**


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[30] Al-Husseini M, Costantine J, Christodoulou C.G, Barbin S.E, El-Hajj A, Kabalan, K.Y (2010) A Reconfigurable Frequency-notched UWB Antenna with Split-ring Resonators. Proceedings of the 2010 Asia-Pacific Microwave Conference (APMC 2010). Yokohama,

[31] Al-Husseini M, Ramadan A, Tawk Y, Christodoulou C.G, El-Hajj A, Kabalan K.Y (2011) Design Based on Complementary Split-ring Resonators of an Antenna with Controllable Band Notches for UWB Cognitive Radio Applications. Proceedings of 2011 IEEE AP-S International Symposium on Antennas and Propagation (IEEE AP-S 2011). Spokane,

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[20] Kelly J.R, Hall P.S, Gardner P, (2011) Band-notched UWB Antenna Incorporating a Microstrip Open-loop Resonator. IEEE transactions on antennas and propagation. 59:8:3045–3048.

24 Advancements in Microstrip and Printed Antennas

propagation letters. 10:682–685.

Nov 2009. pp. II-151–II-153.

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[7] Azari A (2011) A New Super Wideband Fractal Microstrip Antenna. IEEE transactions

[8] Ghaderi M.-R, Mohajeri F (2011) A Compact Hexagonal Wide-slot Antenna with Microstrip-fed Monopole for UWB Application. IEEE antennas and wireless

[9] Liu W, Yin Y, Xu W, Zuo S (2011) Compact Open-slot Antenna with Bandwidth

[10] Al-Husseini M, Ramadan A, El-Hajj A, Kabalan K.Y (2008) A 1.9-13.5 GHz Low-cost Microstrip Antenna. Proceedings of the 2008 International Wireless Communications and Mobile Computing Conference (IWCMC 2008). Crete Island, Greece. 6–8 Aug 2008.

[11] Al-Husseini M, Ramadan A, El-Hajj A, Kabalan K.Y (2009) Design of a Compact and Low-cost Fractal-based UWB PCB Antenna. Proceedings of the 26th National Radio

[12] Al-Husseini M, Tawk Y, El-Hajj A, Kabalan K.Y (2009) A Low-cost Microstrip Antenna for 3G/WLAN/WiMAX and UWB Applications. Proceedings of the 2009 International Conference on Advances in Computational Tools for Engineering Applications (ACTEA

[13] Al-Husseini M, Ramadan A, Tawk Y, El-Hajj A, Kabalan, K.Y (2009) Design and Ground Plane Consideration of a CPW-fed UWB Antenna. Proceedings of the 2009 International Conference on Electrical and Electronics Engineering (ELECO 2009). Bursa, Turkey. 5–8

[14] Al-Husseini M, Ramadan A, Tawk Y, El-Hajj A, Kabalan, K.Y (2011) Design and Ground Plane Optimization of a CPW-fed UWB Antenna. Turkish journal of electrical

[15] Curto S, John M, Ammann M (2007) Groundplane Dependent Performance of Printed Antenna for MB-OFDM-UWB. Proceedings of the IEEE 65th Vehicular Technology

[16] OOi P.C, Selvan K.T (2010) The Effect of Ground Plane on the Performance of a Square Loop CPW-fed Printed Antenna. Progress in electromagnetics research letters.

[17] Arslan H, Sahin M (2007) UWB-based Cognitive Radio Networks. In: Hossain E, Bhargava V, editors. Cognitive Wireless Communication Networks. US: Springer.

[18] Zhang H, Zhou X, Chen T (2009) Ultra-wideband Cognitive Radio for Dynamic Spectrum Accessing Networks. In: Xiao Y, Hu F, editors. Cognitive Radio Networks.

[19] Safatly L, Al-Husseini M, El-Hajj A, Kabalan K.Y (2012) Advanced Techniques and Antenna Design for Pulse Shaping in UWB Cognitive Radio. International journal of

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**Chapter 15**

**Design, Fabrication, and Testing of Flexible Antennas**

Recent years have witnessed a great deal of interest from both academia and industry in the field of flexible electronics. In fact, this research topic tops the pyramid of research priorities

According to market analysis, the revenue of flexible electronics is estimated to be 30 billion

Their light weight, low-cost manufacturing, ease of fabrication, and the availability of inex‐ pensive flexible substrates (i.e.: papers, textiles, and plastics) make flexible electronics an ap‐ pealing candidate for the next generation of consumer electronics [2]. Moreover, recent developments in miniaturized and flexible energy storage and self-powered wireless com‐

Consistently, flexible electronic systems require the integration of flexible antennas operat‐ ing in specific frequency bands to provide wireless connectivity which is highly demanded

Needless to say, the efficiency of these systems primarily depends on the characteristics of the integrated antenna. The nature of flexible wireless technologies requires the integration of flexible, light weight, compact, and low profile antennas. At the same time, these anten‐ nas should be mechanically robust, efficient with a reasonably wide bandwidth and desira‐

This chapter deals with the design, numerical simulation, fabrication process and methods, flexibility tests, and measurements of flexible antennas. As a benchmark, a flexible, compact, and low profile (50.8 μm) printed monopole antenna intended for the ISM band applications at 2.45 GHz is presented and discussed in details. The antenna is based on a Kapton Polyi‐

> © 2013 Khaleel et al.; licensee InTech. This is an open access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

© 2013 Khaleel et al.; licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Haider R. Khaleel, Hussain M. Al-Rizzo and

Additional information is available at the end of the chapter

requested by many national research agencies.

USD in 2017 and over 300 billion USD in 2028 [1].

by today's information oriented society.

ble radiation characteristics.

ponents paved the road for the commercialization of such systems [3].

Ayman I. Abbosh

**1. Introduction**

http://dx.doi.org/10.5772/50841


### **Design, Fabrication, and Testing of Flexible Antennas**

Haider R. Khaleel, Hussain M. Al-Rizzo and Ayman I. Abbosh

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/50841

#### **1. Introduction**

26 Advancements in Microstrip and Printed Antennas

362 Advancement in Microstrip Antennas with Recent Applications

Canada. 11–17 Jul 2010.

16–17 Nov 2009. pp.141–144.

59:11:4008–4015.

Sep 2011. pp. 902–904.

pp. 1214–1217.

Progress in electromagnetics research B. 37:327–342.

(EuCAP2009). Berlin, Germany. 23–27 Mar 2009. pp. 809–812

[32] Al-Husseini M, Safatly L, El-Hajj A, Kabalan K.Y, Christodoulou C.G (2012) Reconfigurable Filter Antennas for Pulse Adaptation in UWB Cognitive Radio Systems.

[33] Ebrahimi E, Hall P.S (2009) A Dual Port Wide-narrowband Antenna for Cognitive Radio. Proceedings of the third European Conference on Antennas and Propagation

[34] Ebrahimi E, Kelly J.R, Hall P.S (2011) Integrated Wide-narrowband Antenna for Multi-standard Radio. IEEE transactions on antennas and propagation. 59:7:2628–2635.

[35] Al-Husseini M, Tawk Y, Christodoulou C.G, El-Hajj A, Kabalan K.Y (2010) A Reconfigurable Cognitive Radio Antenna Design. Proceedings of the 2010 IEEE AP-S International Symposium on Antennas and Propagation (IEEE AP-S 2010). Toronto, ON,

[36] Abu Tarboush H.F, Khan S, NilavalanR, Al-Raweshidy H.S, Budimir D (2009) Reconfigurable Wideband Patch Antenna for Cognitive Radio. Proceedings of the 2009 Loughborough antennas and propagation conference (LAPC 2009), Loughborough, UK.

[37] Hamid M.R, Gardner P, Hall P.S, Ghanem F (2011) Switched-band Vivaldi Antenna.

[38] Hamid M.R, Gardner P, Hall P.S, Ghanem F (2011) Vivaldi Antenna with Integrated Switchable Band Pass Resonator. IEEE transactions on antennas and propagation.

[39] Al-Husseini M, Ramadan A, Zamudio M.E, Christodoulou C.G, El-Hajj A, Kabalan K.Y (2011) A UWB Antenna Combined with a Reconfigurable Bandpass Filter for Cognitive Radio Applications. Proceedings of the 2011 IEEE-APS Topical Conference on Antennas and Propagation in Wireless Communications (IEEE APWC 2011). Torino, Italy. 12–16

[40] Kazerooni M, Cheldavi A, Kamarei M (2009) A Novel Bandpass Defected Microstrip Structure (DMS) Filter for Planar Circuits. Proceedings of the 2009 Progress in Electromagnetics Research Symposium (PIERS 2009). Moscow, Russia. 18–21 Aug 2009.

IEEE transactions on antennas and propagation. 59:5:1472–1480.

Recent years have witnessed a great deal of interest from both academia and industry in the field of flexible electronics. In fact, this research topic tops the pyramid of research priorities requested by many national research agencies.

According to market analysis, the revenue of flexible electronics is estimated to be 30 billion USD in 2017 and over 300 billion USD in 2028 [1].

Their light weight, low-cost manufacturing, ease of fabrication, and the availability of inex‐ pensive flexible substrates (i.e.: papers, textiles, and plastics) make flexible electronics an ap‐ pealing candidate for the next generation of consumer electronics [2]. Moreover, recent developments in miniaturized and flexible energy storage and self-powered wireless com‐ ponents paved the road for the commercialization of such systems [3].

Consistently, flexible electronic systems require the integration of flexible antennas operat‐ ing in specific frequency bands to provide wireless connectivity which is highly demanded by today's information oriented society.

Needless to say, the efficiency of these systems primarily depends on the characteristics of the integrated antenna. The nature of flexible wireless technologies requires the integration of flexible, light weight, compact, and low profile antennas. At the same time, these anten‐ nas should be mechanically robust, efficient with a reasonably wide bandwidth and desira‐ ble radiation characteristics.

This chapter deals with the design, numerical simulation, fabrication process and methods, flexibility tests, and measurements of flexible antennas. As a benchmark, a flexible, compact, and low profile (50.8 μm) printed monopole antenna intended for the ISM band applications at 2.45 GHz is presented and discussed in details. The antenna is based on a Kapton Polyi‐

mide substrate and fabricated using the ink-jet technology. Finally, the performance of the antenna is compared with different antenna types reported in the literature in terms of elec‐ tromagnetic performance and physical properties.

range ( tan *δ* =0.002 ). Furthermore, Kapton Polyimide offers a very low profile (50.8 μm) yet very robust with a tensile strength of 165 MPa at 73°F, a dielectric strength of 3500-7000 volts/mil, and a temperature rating of -65 to 150°C [10]. Other Polymer based and synthe‐

Design, Fabrication, and Testing of Flexible Antennas

http://dx.doi.org/10.5772/50841

365

It is worth mentioning that there are several techniques used to characterize the electromag‐ netic properties of thin and flexible films/substrates such as: the near field microscopy, co‐ planar waveguide approach, differential open resonator method, and goniometric timedomain spectroscopy method [15-18]. However, the most popular method based on measurements of deposited transmission lines incorporating the material to be characterized which determine the dielectric constants of thin films and the conductivities of the metallic

Needless to say, conventional microstrip antennas are not a practical solution for flexible electronics due to their inherently narrow bandwidth which is a function of the substrate's thickness. In [20], a flexible aperture coupled antenna is reported. This technique is known to enhance the impedance bandwidth significantly, however, it leads to an increase in the overall profile; moreover, it involves multi layers, which complicates the fabrication process.

Planar Inverted-F antennas (PIFA) are widely used in mobile phones due the fact that wider impedance bandwidth is obtained despite the presence of a ground plane. Also, antennas incorporating a ground plane promote reduced Specific Absorption Rate (SAR); further‐

In [21], a 50mm × 19mm textile based broadband PIFA fabricated using conductive textiles is proposed for Wireless Body Area Network (WBAN) applications. Although the antenna exhibits a good impedance bandwidth and radiation characteristics, its overall thickness is 6mm which is considered high for the technology under consideration; moreover, it in‐

On the other hand, planar monopole and dipole antennas have received much interest over other antenna types due to their relatively large impedance bandwidth, low profile, ease of fabrication, and omni directional radiation pattern which is highly preferred in many wire‐

Given the technology envisioned in this chapter, Co-Planar Waveguide (CPW) is preferred over other feeding techniques since no via holes or shorting pins are involved, in addition to several useful characteristics such as: low radiation losses, larger bandwidth, improved im‐ pedance matching, and more importantly, both radiating element and ground plane are printed on the same side of the substrate, which promotes low fabrication cost and complex‐

more, their matching is less affected by the proximity of the human body.

volves a multi-layer complex, and inaccurate fabrication process.

ity in addition to the capability of roll to roll production.

sized flexible substrates have been also used in several designs [11-14].

lines over a broad frequency range [19].

**3. Choice of Antenna Type**

less schemes.

**Figure 1.** Flexible printed monopole antenna based on Kapton Polyimide substrate.

#### **2. Choice of Antenna Substrate**

To comply with flexible technologies, integrated components need to be highly flexible and mechanically robust; they also have to exhibit high tolerance levels in terms of bending re‐ peatability and thermal endurance. A plethora of design approaches of flexible and confor‐ mal antennas were reported in the literature including Electro-textile [4], paper-based [5], fluidic [6], and synthesized flexible substrates [7]. In [4], a 150 mm × 180 mm flexible Electro textile antenna based on a 4 mm felt fabric is proposed. The antenna operates in the ISM 2.45 GHz band. Although it is suitable for wearable and conformal applications, fabric substrates are prone to discontinuities, fluids absorption, and crumpling.

In [5], a flexible single band antenna printed on a 46mm × 30mm paper-based substrate was proposed for integration into flexible displays for WLAN applications. However, paper based substrates are found to be not robust enough and introduce discontinuities when used in applications that require high levels of bending and rolling. Moreover, they have a relatively high loss factor (loss tangent (tan*δ*) is around 0.07 at 2.45 GHz) which compromis‐ es the antenna's efficiency [8].

Kapton Polyimide film was chosen as the antenna substrate in [9] due to its good balance of physical, chemical, and electrical properties with a low loss factor over a wide frequency range ( tan *δ* =0.002 ). Furthermore, Kapton Polyimide offers a very low profile (50.8 μm) yet very robust with a tensile strength of 165 MPa at 73°F, a dielectric strength of 3500-7000 volts/mil, and a temperature rating of -65 to 150°C [10]. Other Polymer based and synthe‐ sized flexible substrates have been also used in several designs [11-14].

It is worth mentioning that there are several techniques used to characterize the electromag‐ netic properties of thin and flexible films/substrates such as: the near field microscopy, co‐ planar waveguide approach, differential open resonator method, and goniometric timedomain spectroscopy method [15-18]. However, the most popular method based on measurements of deposited transmission lines incorporating the material to be characterized which determine the dielectric constants of thin films and the conductivities of the metallic lines over a broad frequency range [19].

### **3. Choice of Antenna Type**

mide substrate and fabricated using the ink-jet technology. Finally, the performance of the antenna is compared with different antenna types reported in the literature in terms of elec‐

To comply with flexible technologies, integrated components need to be highly flexible and mechanically robust; they also have to exhibit high tolerance levels in terms of bending re‐ peatability and thermal endurance. A plethora of design approaches of flexible and confor‐ mal antennas were reported in the literature including Electro-textile [4], paper-based [5], fluidic [6], and synthesized flexible substrates [7]. In [4], a 150 mm × 180 mm flexible Electro textile antenna based on a 4 mm felt fabric is proposed. The antenna operates in the ISM 2.45 GHz band. Although it is suitable for wearable and conformal applications, fabric substrates

In [5], a flexible single band antenna printed on a 46mm × 30mm paper-based substrate was proposed for integration into flexible displays for WLAN applications. However, paper based substrates are found to be not robust enough and introduce discontinuities when used in applications that require high levels of bending and rolling. Moreover, they have a relatively high loss factor (loss tangent (tan*δ*) is around 0.07 at 2.45 GHz) which compromis‐

Kapton Polyimide film was chosen as the antenna substrate in [9] due to its good balance of physical, chemical, and electrical properties with a low loss factor over a wide frequency

tromagnetic performance and physical properties.

364 Advancement in Microstrip Antennas with Recent Applications

**Figure 1.** Flexible printed monopole antenna based on Kapton Polyimide substrate.

are prone to discontinuities, fluids absorption, and crumpling.

**2. Choice of Antenna Substrate**

es the antenna's efficiency [8].

Needless to say, conventional microstrip antennas are not a practical solution for flexible electronics due to their inherently narrow bandwidth which is a function of the substrate's thickness. In [20], a flexible aperture coupled antenna is reported. This technique is known to enhance the impedance bandwidth significantly, however, it leads to an increase in the overall profile; moreover, it involves multi layers, which complicates the fabrication process.

Planar Inverted-F antennas (PIFA) are widely used in mobile phones due the fact that wider impedance bandwidth is obtained despite the presence of a ground plane. Also, antennas incorporating a ground plane promote reduced Specific Absorption Rate (SAR); further‐ more, their matching is less affected by the proximity of the human body.

In [21], a 50mm × 19mm textile based broadband PIFA fabricated using conductive textiles is proposed for Wireless Body Area Network (WBAN) applications. Although the antenna exhibits a good impedance bandwidth and radiation characteristics, its overall thickness is 6mm which is considered high for the technology under consideration; moreover, it in‐ volves a multi-layer complex, and inaccurate fabrication process.

On the other hand, planar monopole and dipole antennas have received much interest over other antenna types due to their relatively large impedance bandwidth, low profile, ease of fabrication, and omni directional radiation pattern which is highly preferred in many wire‐ less schemes.

Given the technology envisioned in this chapter, Co-Planar Waveguide (CPW) is preferred over other feeding techniques since no via holes or shorting pins are involved, in addition to several useful characteristics such as: low radiation losses, larger bandwidth, improved im‐ pedance matching, and more importantly, both radiating element and ground plane are printed on the same side of the substrate, which promotes low fabrication cost and complex‐ ity in addition to the capability of roll to roll production.

#### **4. Choice of Fabrication Method**

This section reviews the currently available methods for fabrication of flexible and wearable antennas. Method overview, advantages, and drawbacks of each technique are discussed.

with ink viscosity and surface energy of the substrate [22]. Figure 2 depicts a scheme of the

Design, Fabrication, and Testing of Flexible Antennas

http://dx.doi.org/10.5772/50841

367

Chemical etching often accompanied by photolithography is the process of fabricating met‐ allic patterns using a photoresist and etchants to mill out a selected area corrosively. This technique has emerged in the 1960s as a branch-out of the Printed Circuit Board (PCB) in‐ dustry. Chemical etching gained a wide popularity since it can produce highly complex pat‐

Photoresist materials are organic polymers whose chemical characteristics change when ex‐ posed to ultraviolet light. When the exposed area becomes more soluble in the developer, the photoresist is positive. While if it becomes less soluble, the compound is considered a

A major drawback of negative resists is that the exposed regions swell as the counterpart is dissolved by the developer, which compromises the resolution of the process. Swelling oc‐ curs due to the penetration of the developer solution into the photoresist material which in turn leads to a distortion in the patterned region [27]. Hence, current practice in the photoli‐ thography based antenna and RF circuits Industry relies mainly on positive resists since

Although patterns with high complexity and fine details can be produced using this techni‐ que, its lengthy process, low throughput, involvement of dangerous chemicals (neutraliza‐ tion is required), clean room requirement, in addition to byproduct and waste leftovers are major drawbacks of this technology. The reader is referred to [28] for further information on

screen printing process.

**4.2. Chemical Etching**

negative resist.

terns with high resolution accurately [27].

they present higher resolution than negative resists.

**Figure 3.** Process flow of the chemical etching (photolithography) process.

this method. The chemical etching process is illustrated in Figure 3.

#### **4.1. Screen Printing**

Screen printing is one of the simplest and most cost effective techniques used by electronics manufacturers. This technique is based on a woven screen that has different thicknesses and thread densities. To produce a printed pattern, a squeegee blade is driven down forcing the screen into contact with the affixed substrate. This in turn forces the ink to be ejected through the exposed areas of the screen on the substrate, and thus, the desired pattern is formed [22]. Polyester and stainless steel are the most common materials used in this technology.

Three different screen printing methods are currently used: flat bed, cylinder, and rotary. Flat bed is the simplest and most common screen printing method. Cylinder screen printing is quite similar to the flat bed except the pattern is deposited as the substrate rotates while attached to the screen roll. In rotary screen, ink and squeegee assembly are rotated inside a rolled screen where impression cylinder produces pressure to substrate [23]. Rotary screen enables much higher throughput capacity than flat bed screen; hence, it is often integrated into a roll to roll production line.

**Figure 2.** Illustration of the screen printing process.

Screen printing is an additive process as opposed to the subtractive process of chemical etching which makes it a more cost-effective and environmentally friendly. Rather than masking a screen, the patterned mask is applied onto the substrate directly where the con‐ ductive ink is administered and thermally cured. Several RFIDs and flexible transparent an‐ tennas have been prototyped successfully using this technique [24-26]. However, there are some problems associated with this technique including the limited control over the thick‐ ness, number of passes, and resolution of the printed patterns. Layer consistency is also a challenge, as thermal curing of solvent based inks could leave behind artifacts that change with ink viscosity and surface energy of the substrate [22]. Figure 2 depicts a scheme of the screen printing process.

#### **4.2. Chemical Etching**

**4. Choice of Fabrication Method**

366 Advancement in Microstrip Antennas with Recent Applications

**4.1. Screen Printing**

into a roll to roll production line.

**Figure 2.** Illustration of the screen printing process.

This section reviews the currently available methods for fabrication of flexible and wearable antennas. Method overview, advantages, and drawbacks of each technique are discussed.

Screen printing is one of the simplest and most cost effective techniques used by electronics manufacturers. This technique is based on a woven screen that has different thicknesses and thread densities. To produce a printed pattern, a squeegee blade is driven down forcing the screen into contact with the affixed substrate. This in turn forces the ink to be ejected through the exposed areas of the screen on the substrate, and thus, the desired pattern is formed [22].

Three different screen printing methods are currently used: flat bed, cylinder, and rotary. Flat bed is the simplest and most common screen printing method. Cylinder screen printing is quite similar to the flat bed except the pattern is deposited as the substrate rotates while attached to the screen roll. In rotary screen, ink and squeegee assembly are rotated inside a rolled screen where impression cylinder produces pressure to substrate [23]. Rotary screen enables much higher throughput capacity than flat bed screen; hence, it is often integrated

Screen printing is an additive process as opposed to the subtractive process of chemical etching which makes it a more cost-effective and environmentally friendly. Rather than masking a screen, the patterned mask is applied onto the substrate directly where the con‐ ductive ink is administered and thermally cured. Several RFIDs and flexible transparent an‐ tennas have been prototyped successfully using this technique [24-26]. However, there are some problems associated with this technique including the limited control over the thick‐ ness, number of passes, and resolution of the printed patterns. Layer consistency is also a challenge, as thermal curing of solvent based inks could leave behind artifacts that change

Polyester and stainless steel are the most common materials used in this technology.

Chemical etching often accompanied by photolithography is the process of fabricating met‐ allic patterns using a photoresist and etchants to mill out a selected area corrosively. This technique has emerged in the 1960s as a branch-out of the Printed Circuit Board (PCB) in‐ dustry. Chemical etching gained a wide popularity since it can produce highly complex pat‐ terns with high resolution accurately [27].

Photoresist materials are organic polymers whose chemical characteristics change when ex‐ posed to ultraviolet light. When the exposed area becomes more soluble in the developer, the photoresist is positive. While if it becomes less soluble, the compound is considered a negative resist.

A major drawback of negative resists is that the exposed regions swell as the counterpart is dissolved by the developer, which compromises the resolution of the process. Swelling oc‐ curs due to the penetration of the developer solution into the photoresist material which in turn leads to a distortion in the patterned region [27]. Hence, current practice in the photoli‐ thography based antenna and RF circuits Industry relies mainly on positive resists since they present higher resolution than negative resists.

**Figure 3.** Process flow of the chemical etching (photolithography) process.

Although patterns with high complexity and fine details can be produced using this techni‐ que, its lengthy process, low throughput, involvement of dangerous chemicals (neutraliza‐ tion is required), clean room requirement, in addition to byproduct and waste leftovers are major drawbacks of this technology. The reader is referred to [28] for further information on this method. The chemical etching process is illustrated in Figure 3.

#### **4.3. Flexography**

Flexography is a type of relief printing. An image is produced by a printmaking process where a protuberating surface of the printing plate matrix is inked while the recessed areas are free of ink. Image printing is a simple process since it only involves inking the protrud‐ ing surface of the matrix and bringing it in contact with the substrate [29]. Due to its rela‐ tively fine resolution, low cost, and high throughput, flexography gained a great interest by RFID antenna manufacturers. Moreover, this technique requires a lower viscosity ink than screen printing inks, and yields imaged (printed) dry films of a thickness of less than 2.5 μm. Hence, flexography inks need to posses higher bulk conductivity than those used in screen printing to compensate for the increase in sheet resistance since the efficiency of printed an‐ tennas depends mainly on the electrical conductivity of the traced pattern. Substrate param‐ eters like surface porosity, hydrophobicity, and surface energy have a direct influence on the ink film thickness of the printed trace [23]. The consistency in ink film thickness and line width has also a profound impact on the sheet resistance. The process scheme is demon‐ strated in Figure 4.

This new technology utilizes conductive inks based on different nano-structural materials such as silver nano-particle based ink, which is widely used due to its high conductivity. This type of printing technique can be categorized into two types: drop-on-demand and continuous inkjet. Drop on demand print heads apply pressurized pulses to ink with either a piezo or thermo element in which drives a drop from a nozzle when needed. Most printed electronics manufacturers utilize the piezo pulse type [31]. Printing quality depends mainly on the ink characteristics such as viscosity, surface tension, and particle size. The surface topology of the substrate, the platen temperature and the print head parameters are also im‐

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369

Printing processes and setups are completely controlled from the user's computer, and do not require a clean-room environment which reduces the levels of environmental contami‐

Unlike photolithography, which is a subtractive method since it involves removing unwant‐ ed pattern from the substrate's metallic/conductive side; inkjet printing deposits a controlled amount of ink droplets from the nozzle to the specified position. Hence, no waste or by‐ product is produced resulting in an economical, clean, and fast solution. Figure 5 depicts the

To benchmark the performance of flexible antennas, a single band printed monopole is pre‐ sented in this section which has the merits of light weight, ultra low profile (50.8 μm), large bandwidth, robustness, compactness, and high efficiency. The antenna design which is fab‐

portant factors.

nation [31].

ink jet printing process.

**Figure 5.** Illustration of the ink jet (droplet on-demand) printing process.

**5. Benchmarking Prototype**

**Figure 4.** Illustration of the flexography printing process based on flexible relief plate.

#### **4.4. Ink Jet Printing**

Inkjet printing of RF circuits and antennas using highly conductive inks have become ex‐ tremely popular in recent years. New inkjet material printers operate by depositing ink droplets of a size down to few pico liters, and hence, these high resolution printers can pro‐ duce compact designs with tiny details accurately [30].

This new technology utilizes conductive inks based on different nano-structural materials such as silver nano-particle based ink, which is widely used due to its high conductivity. This type of printing technique can be categorized into two types: drop-on-demand and continuous inkjet. Drop on demand print heads apply pressurized pulses to ink with either a piezo or thermo element in which drives a drop from a nozzle when needed. Most printed electronics manufacturers utilize the piezo pulse type [31]. Printing quality depends mainly on the ink characteristics such as viscosity, surface tension, and particle size. The surface topology of the substrate, the platen temperature and the print head parameters are also im‐ portant factors.

Printing processes and setups are completely controlled from the user's computer, and do not require a clean-room environment which reduces the levels of environmental contami‐ nation [31].

Unlike photolithography, which is a subtractive method since it involves removing unwant‐ ed pattern from the substrate's metallic/conductive side; inkjet printing deposits a controlled amount of ink droplets from the nozzle to the specified position. Hence, no waste or by‐ product is produced resulting in an economical, clean, and fast solution. Figure 5 depicts the ink jet printing process.

**Figure 5.** Illustration of the ink jet (droplet on-demand) printing process.

#### **5. Benchmarking Prototype**

**4.3. Flexography**

368 Advancement in Microstrip Antennas with Recent Applications

strated in Figure 4.

**4.4. Ink Jet Printing**

Flexography is a type of relief printing. An image is produced by a printmaking process where a protuberating surface of the printing plate matrix is inked while the recessed areas are free of ink. Image printing is a simple process since it only involves inking the protrud‐ ing surface of the matrix and bringing it in contact with the substrate [29]. Due to its rela‐ tively fine resolution, low cost, and high throughput, flexography gained a great interest by RFID antenna manufacturers. Moreover, this technique requires a lower viscosity ink than screen printing inks, and yields imaged (printed) dry films of a thickness of less than 2.5 μm. Hence, flexography inks need to posses higher bulk conductivity than those used in screen printing to compensate for the increase in sheet resistance since the efficiency of printed an‐ tennas depends mainly on the electrical conductivity of the traced pattern. Substrate param‐ eters like surface porosity, hydrophobicity, and surface energy have a direct influence on the ink film thickness of the printed trace [23]. The consistency in ink film thickness and line width has also a profound impact on the sheet resistance. The process scheme is demon‐

**Figure 4.** Illustration of the flexography printing process based on flexible relief plate.

duce compact designs with tiny details accurately [30].

Inkjet printing of RF circuits and antennas using highly conductive inks have become ex‐ tremely popular in recent years. New inkjet material printers operate by depositing ink droplets of a size down to few pico liters, and hence, these high resolution printers can pro‐

To benchmark the performance of flexible antennas, a single band printed monopole is pre‐ sented in this section which has the merits of light weight, ultra low profile (50.8 μm), large bandwidth, robustness, compactness, and high efficiency. The antenna design which is fab‐ ricated using the inkjet printing technology covers the ISM 2.45 GHz and fed by a CPW feed. Moreover, the performance of the antenna is evaluated under bending effects in terms of impedance matching and shift in resonant frequency. Finally, the characteristics of the an‐ tenna under study are compared to several flexible antenna types reported in the literature.

between the arms which in turn degrades the impedance matching. The split ring monopole is fed by CPW feed, which adds the merit of fabrication simplicity since both the radiating

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**Figure 6.** Dielectric constant Vs. frequency for the HN type Kapton polyimide (125 µm). The characteristics are similar to the 50 µm used in the reported prototype. Curve A is for measurement at 25°C (77°F) and 45% RH with the electric field in the plane of the sheet, while Curve B is the same measurement after conditioning the film at 100°C [10].

**Figure 7.** Geometry and dimensions of the reported Split Ring printed monopole antenna (the grey colored area rep‐

resents the ground plane and the radiating element).

element and ground plane are printed on the same side of the substrate.

#### **5.1. ISM Band Printed Monopole Antenna**

The ISM 2.45 GHz band is internationally recognized as one the most commonly used stand‐ ards in wireless communication systems [32]. For example, all of Wireless local-area net‐ works (WLAN), IEEE 802.11/WiFi, Bluetooth and Personal Area Network (PAN) IEEE 802.15.4, ZigBee utilize the ISM 2.45 GHz band. Additionally, several potential applications based on these technologies are possibly applied in the future.

Obviously, the integration of a wireless connectivity based on the abovementioned technolo‐ gies within flexible devices triggers the need for ultra light/thin/flexible antennas. At the same time, these antennas should be robust, cost effective, and highly efficient with desira‐ ble radiation characteristics.

In response to such needs, several design approaches of flexible and conformal antennas based on flexible substrates were reported in the literature [33-39]. In [32], a flexible antenna printed on a 46mm x 30mm paper-based substrate was proposed for integration into flexible displays for WLAN applications. However, paper based substrates are found to be not ro‐ bust enough and introduce discontinuities when used in applications that require high lev‐ els of bending and rolling as mentioned earlier. Moreover, they have a high loss factor (loss tangent (tan*δ*) is around 0.07 at 2.45 GHz) which compromises the antenna's efficiency. In [34], a stretchable antenna based on an elastic substrate is presented. The design offers a good solution in terms of flexibility and stretchability; however, it involves a complex man‐ ufacturing process where the conductors are realized by injecting a room temperature liquid metal alloy into molded micro-structured channels on an elastic dielectric material followed by channels encapsulation. In [35], a conformal exponentially tapered slot antenna based on a 200 μm Liquid Crystal Polymer (LCP) substrate is reported. The design exhibits excellent radiation characteristics; however, the dimensions (130mm × 43mm) are too large for inte‐ gration within modern compact and flexible electronics. In this section, a flexible compact split ring printed monopole antenna intended for flexible/wearable/conformal applications is presented. The antenna is printed on a 50.8 μm Kapton substrate and fed by a CPW. Both radiating element and ground plane are printed on the same side of the substrate which pro‐ motes low fabrication cost and complexity in addition to roll to roll production.

#### *5.1.1. Antenna Design*

As shown in Figure 2, the antenna consists of a square split ring shaped radiating element fed by a CPW. The winding lengthens the current path which in turn reduces the structure size without significant efficiency degradation or disturbance to the radiation pattern. The separation distance between the arms is optimized as 5 mm to achieve the least return loss. It is worth mentioning that a smaller separation leads to an increased capacitive coupling between the arms which in turn degrades the impedance matching. The split ring monopole is fed by CPW feed, which adds the merit of fabrication simplicity since both the radiating element and ground plane are printed on the same side of the substrate.

ricated using the inkjet printing technology covers the ISM 2.45 GHz and fed by a CPW feed. Moreover, the performance of the antenna is evaluated under bending effects in terms of impedance matching and shift in resonant frequency. Finally, the characteristics of the an‐ tenna under study are compared to several flexible antenna types reported in the literature.

The ISM 2.45 GHz band is internationally recognized as one the most commonly used stand‐ ards in wireless communication systems [32]. For example, all of Wireless local-area net‐ works (WLAN), IEEE 802.11/WiFi, Bluetooth and Personal Area Network (PAN) IEEE 802.15.4, ZigBee utilize the ISM 2.45 GHz band. Additionally, several potential applications

Obviously, the integration of a wireless connectivity based on the abovementioned technolo‐ gies within flexible devices triggers the need for ultra light/thin/flexible antennas. At the same time, these antennas should be robust, cost effective, and highly efficient with desira‐

In response to such needs, several design approaches of flexible and conformal antennas based on flexible substrates were reported in the literature [33-39]. In [32], a flexible antenna printed on a 46mm x 30mm paper-based substrate was proposed for integration into flexible displays for WLAN applications. However, paper based substrates are found to be not ro‐ bust enough and introduce discontinuities when used in applications that require high lev‐ els of bending and rolling as mentioned earlier. Moreover, they have a high loss factor (loss tangent (tan*δ*) is around 0.07 at 2.45 GHz) which compromises the antenna's efficiency. In [34], a stretchable antenna based on an elastic substrate is presented. The design offers a good solution in terms of flexibility and stretchability; however, it involves a complex man‐ ufacturing process where the conductors are realized by injecting a room temperature liquid metal alloy into molded micro-structured channels on an elastic dielectric material followed by channels encapsulation. In [35], a conformal exponentially tapered slot antenna based on a 200 μm Liquid Crystal Polymer (LCP) substrate is reported. The design exhibits excellent radiation characteristics; however, the dimensions (130mm × 43mm) are too large for inte‐ gration within modern compact and flexible electronics. In this section, a flexible compact split ring printed monopole antenna intended for flexible/wearable/conformal applications is presented. The antenna is printed on a 50.8 μm Kapton substrate and fed by a CPW. Both radiating element and ground plane are printed on the same side of the substrate which pro‐

motes low fabrication cost and complexity in addition to roll to roll production.

As shown in Figure 2, the antenna consists of a square split ring shaped radiating element fed by a CPW. The winding lengthens the current path which in turn reduces the structure size without significant efficiency degradation or disturbance to the radiation pattern. The separation distance between the arms is optimized as 5 mm to achieve the least return loss. It is worth mentioning that a smaller separation leads to an increased capacitive coupling

**5.1. ISM Band Printed Monopole Antenna**

370 Advancement in Microstrip Antennas with Recent Applications

ble radiation characteristics.

*5.1.1. Antenna Design*

based on these technologies are possibly applied in the future.

**Figure 6.** Dielectric constant Vs. frequency for the HN type Kapton polyimide (125 µm). The characteristics are similar to the 50 µm used in the reported prototype. Curve A is for measurement at 25°C (77°F) and 45% RH with the electric field in the plane of the sheet, while Curve B is the same measurement after conditioning the film at 100°C [10].

**Figure 7.** Geometry and dimensions of the reported Split Ring printed monopole antenna (the grey colored area rep‐ resents the ground plane and the radiating element).

The antenna structure is printed on a 38 mm × 25 mm Kapton Polyimide substrate with a dielectric constant of 3.4 and a loss tangent of 0.002 (dielectric constant versus frequency is provided in Figure 6). The geometry and dimensions of the antenna are depicted in Figure 7 and Table 1.

tix Drop Manager Software in a Gerber file format which contains all the geometrical dimensions of the antenna design. Moreover, all printing processes and setup conditions can be control‐ led using the Dimatix Drop Manager software such as the number of layers to be printed, heating temperature of the platen desk, number of nozzles used in operation and height of

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A conductive ink based on sliver nano particles is deposited over the substrate utilizing 16 nozzles with 25 μm drop spacing. After the printing process is completed, thermal anneal‐ ing is required to evaporate excess solvent and to remove ink impurities. Furthermore, the thermal annealing process provides an increased bond of the deposited material. The report‐ ed antenna is cured at 100° for 4 hours by a LPKF Protoflow industrial oven. It is worth mentioning that 2 layers of ink were deposited on the substrate to achieve a robust and continuous radiating element and more importantly to increase the electrical conductivity. It should be noted that due to the excellent thermal rating of kapton polyimide (-65 to 150°C), no shrinking was experienced during the annealing process. For measurement purposes, the

antenna is fed by a 50 Ω SubMiniature Version A (SMA) coaxial RF connector.

**Figure 9.** Final printed Polyimide based antenna prototype after thermal annealing.

The S-parameters were measured using an Agilent PNA-X series N5242A Vector Network Analyzer (VNA) with (10 MHz-26.5 GHz) frequency range. As can be seen in Figure 10, a good agreement is achieved between the simulated and measured reflection coefficient S11 for the

*5.2.1. Reflection Coefficient S11*

the cartridge head with respect to the substrate.


**Table 1.** ISM Band Printed Monopole Antenna Dimensions in Millimeter.

#### *5.1.2. Simulations, Fabrication and Measurements*

Design and analysis of the reported printed monopole antennas have been carried out using the full wave simulation software CST Microwave Studio which is based on the Finite Inte‐ gration Technique (FIT) [40].

To ensure a high simulation accuracy, the number of mesh cells was mainly determined through sufficient meshing of the antenna element where the smallest geometric detail (i.e. CPW gap, microstrip line, etc,..) is covered by at least three mesh cells both horizontally and vertically. The total number of the mesh cells generated for the antenna under study is 498,750 cells.

**Figure 8.** Installing the polyimide substrate on the platen of the material printer.

Before starting the printing process which is performed using the Dimatix DMP 2831 Fuji‐ film material printer [41], the final simulated design is exported to the printer using Dima‐ tix Drop Manager Software in a Gerber file format which contains all the geometrical dimensions of the antenna design. Moreover, all printing processes and setup conditions can be control‐ led using the Dimatix Drop Manager software such as the number of layers to be printed, heating temperature of the platen desk, number of nozzles used in operation and height of the cartridge head with respect to the substrate.

A conductive ink based on sliver nano particles is deposited over the substrate utilizing 16 nozzles with 25 μm drop spacing. After the printing process is completed, thermal anneal‐ ing is required to evaporate excess solvent and to remove ink impurities. Furthermore, the thermal annealing process provides an increased bond of the deposited material. The report‐ ed antenna is cured at 100° for 4 hours by a LPKF Protoflow industrial oven. It is worth mentioning that 2 layers of ink were deposited on the substrate to achieve a robust and continuous radiating element and more importantly to increase the electrical conductivity. It should be noted that due to the excellent thermal rating of kapton polyimide (-65 to 150°C), no shrinking was experienced during the annealing process. For measurement purposes, the antenna is fed by a 50 Ω SubMiniature Version A (SMA) coaxial RF connector.

**Figure 9.** Final printed Polyimide based antenna prototype after thermal annealing.

#### *5.2.1. Reflection Coefficient S11*

The antenna structure is printed on a 38 mm × 25 mm Kapton Polyimide substrate with a dielectric constant of 3.4 and a loss tangent of 0.002 (dielectric constant versus frequency is provided in Figure 6). The geometry and dimensions of the antenna are depicted in Figure 7

Design and analysis of the reported printed monopole antennas have been carried out using the full wave simulation software CST Microwave Studio which is based on the Finite Inte‐

To ensure a high simulation accuracy, the number of mesh cells was mainly determined through sufficient meshing of the antenna element where the smallest geometric detail (i.e. CPW gap, microstrip line, etc,..) is covered by at least three mesh cells both horizontally and vertically. The total number of the mesh cells generated for the antenna under study is 498,750 cells.

Before starting the printing process which is performed using the Dimatix DMP 2831 Fuji‐ film material printer [41], the final simulated design is exported to the printer using Dima‐

*L1* 38 *W1* 25 *L2* 26 *W2* 18 *L3* 19 *W3* 10.5 *L4* 9.5 *W4* 6.5 *L5* 3 *W5* 2 *G1* 2 *G2* 0.5

**Table 1.** ISM Band Printed Monopole Antenna Dimensions in Millimeter.

**Figure 8.** Installing the polyimide substrate on the platen of the material printer.

*5.1.2. Simulations, Fabrication and Measurements*

372 Advancement in Microstrip Antennas with Recent Applications

gration Technique (FIT) [40].

and Table 1.

The S-parameters were measured using an Agilent PNA-X series N5242A Vector Network Analyzer (VNA) with (10 MHz-26.5 GHz) frequency range. As can be seen in Figure 10, a good agreement is achieved between the simulated and measured reflection coefficient S11 for the split ring antenna. The simulated return loss for the antenna is 27 dB at 2.45 GHz, with a -10 dB bandwidth of 430 MHz. The measured return loss is -28.5 dB at 2.39 GHz with a -10 dB bandwidth of 540 MHz. The increase in the measured bandwidth is attributed to the de‐ creased electrical conductivity caused by the solvent and impurities found in the silver nanoparticle ink, which in turn increases the quality factor and leads to bandwidth enlargement.

E-plane (*YZ* cut) and H-plane (*XZ* cut) far-field radiation patterns are shown in Figure 12. It can be seen that the radiation power is omni-directional at the resonant frequency. The an‐ tenna achieved a measured gain of 1.65 dBi which fairly agrees with the simulated value.

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**Figure 12.** Measured and simulated radiation patterns for the split ring printed monopole at 2.45 GHz (a) E-plane (*YZ*)

and (b) H-plane (*XZ*).

**Figure 10.** Measured and simulated reflection coefficient S11 for the split ring antenna.

#### **5.2.2 Far-field Radiation Patterns**

The far-field radiation patterns of the principal planes (E and H) were measured in a fully equipped anechoic chamber. The Antenna Under Test (AUT) was placed on an ETS Lindg‐ ren 2090 positioner and aligned to a horn antenna with adjustable polarization.

**Figure 11.** Radiation pattern measurement setup inside an anechoic chamber.

E-plane (*YZ* cut) and H-plane (*XZ* cut) far-field radiation patterns are shown in Figure 12. It can be seen that the radiation power is omni-directional at the resonant frequency. The an‐ tenna achieved a measured gain of 1.65 dBi which fairly agrees with the simulated value.

split ring antenna. The simulated return loss for the antenna is 27 dB at 2.45 GHz, with a -10 dB bandwidth of 430 MHz. The measured return loss is -28.5 dB at 2.39 GHz with a -10 dB bandwidth of 540 MHz. The increase in the measured bandwidth is attributed to the de‐ creased electrical conductivity caused by the solvent and impurities found in the silver nanoparticle ink, which in turn increases the quality factor and leads to bandwidth enlargement.

**Figure 10.** Measured and simulated reflection coefficient S11 for the split ring antenna.

**Figure 11.** Radiation pattern measurement setup inside an anechoic chamber.

The far-field radiation patterns of the principal planes (E and H) were measured in a fully equipped anechoic chamber. The Antenna Under Test (AUT) was placed on an ETS Lindg‐

ren 2090 positioner and aligned to a horn antenna with adjustable polarization.

**5.2.2 Far-field Radiation Patterns**

374 Advancement in Microstrip Antennas with Recent Applications

**Figure 12.** Measured and simulated radiation patterns for the split ring printed monopole at 2.45 GHz (a) E-plane (*YZ*) and (b) H-plane (*XZ*).

#### *5.2.3. Flexibility Tests*

Since the antenna is expected to be bent and rolled when worn or integrated within flexible devices, three tests need to be conducted for operative validation:

ics a moderate extent of bending; while a shift of 80 MHz is observed in the extreme case where the antenna is curved on a 8mm radius foam cylinder, while the antenna is less affect‐ ed when bent in the vertical plane. However, the impedance bandwidth of the AUT is rela‐ tively large, which could overcome the shift caused by the bending effect. Figure 13 shows the flexibility test setup for the dual band antenna rolled on a foam cylinder with an 8 mm radius. Figure 14 depicts the reflection coefficient of the bent cases in both horizontal and

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**Figure 14.** Measured S11 for the reported antenna when bent on a foam cylinder with different radii (*r*=10 mm and

*r*=8 mm) to mimic different bending extents. (a) horizontal plane, (b) vertical plane.

vertical planes compared to the flat case.


As stated before, Polyimide Kapton substrate was chosen for this technology mainly due to its physical robustness and high flexibility. Furthermore, the fabricated prototype demon‐ strated an excellent performance as it was tested repeatedly against bending, twisting, and rolling effects.

AUT is conformed on foam cylinders with different radii (the first is *r*=10mm and the sec‐ ond is *r*=8mm) to emulate different extents of bending while it is connected to the net‐ work analyzer.

**Figure 13.** Flexibility test setup (AUT is conformed over a cylindrical foam with different radii to reflect different ex‐ tents of bending).

As can be seen in Figure 14, around 35 MHz shift to a higher resonant frequency is experi‐ enced when the antenna is horizontally conformed on a 10 mm radius cylinder which mim‐ ics a moderate extent of bending; while a shift of 80 MHz is observed in the extreme case where the antenna is curved on a 8mm radius foam cylinder, while the antenna is less affect‐ ed when bent in the vertical plane. However, the impedance bandwidth of the AUT is rela‐ tively large, which could overcome the shift caused by the bending effect. Figure 13 shows the flexibility test setup for the dual band antenna rolled on a foam cylinder with an 8 mm radius. Figure 14 depicts the reflection coefficient of the bent cases in both horizontal and vertical planes compared to the flat case.

*5.2.3. Flexibility Tests*

rolling effects.

work analyzer.

tents of bending).

Since the antenna is expected to be bent and rolled when worn or integrated within flexible

**•** Durability and robustness tests are required, which is performed by repeated testing of the fabricated antennas under bending, rolling and twisting to monitor the deposited conduc‐ tive ink for any deformations, discontinuities, and to ensure there are no cracks wrinkles or permanent folds are introduced, which might compromise the antennas performance. **•** Resonant frequency and return loss need to be evaluated under bending conditions since they are prone to shift/decrease due to impedance mismatch and a change in the effective

**•** Radiation patterns and gain of the antenna are required to be tested for distortion and/or

As stated before, Polyimide Kapton substrate was chosen for this technology mainly due to its physical robustness and high flexibility. Furthermore, the fabricated prototype demon‐ strated an excellent performance as it was tested repeatedly against bending, twisting, and

AUT is conformed on foam cylinders with different radii (the first is *r*=10mm and the sec‐ ond is *r*=8mm) to emulate different extents of bending while it is connected to the net‐

**Figure 13.** Flexibility test setup (AUT is conformed over a cylindrical foam with different radii to reflect different ex‐

As can be seen in Figure 14, around 35 MHz shift to a higher resonant frequency is experi‐ enced when the antenna is horizontally conformed on a 10 mm radius cylinder which mim‐

devices, three tests need to be conducted for operative validation:

electrical length of the radiating elements.

376 Advancement in Microstrip Antennas with Recent Applications

degradation when conformed on a curved surface.

**Figure 14.** Measured S11 for the reported antenna when bent on a foam cylinder with different radii (*r*=10 mm and *r*=8 mm) to mimic different bending extents. (a) horizontal plane, (b) vertical plane.

#### **5.2.4 Comparative Study**

The split ring antenna design is compared to different types of flexible antennas reported in [4]-[7]. Given the applications envisioned in this study, the comparative study is focused on compactness (size and thickness), electrical properties and robustness. Robustness encom‐ passes the major mechanical properties related to flexible/conformal electronic devices such as tensile strength, flexural strength, deformability, and thermal stability. Fabrication com‐ plexity criterion is also considered in this comparative study. Table 2 depicts these charac‐ teristics of the antenna under study.

2.45 GHz ISM band is presented which has the merits of light weight, ultra low profile, wide bandwidth, robustness, compactness, and high efficiency. The reported design is based on a Kapton Polyimide substrate which is known for its flexibility, robustness and low dielectric losses. The prototype was fabricated using the inkjet printing technology. Furthermore, the antenna is tested under bending effects since it is expected to be flexed or conformed on curved surfaces. Flexibility, robustness, compactness, fabrication simplicity along with good radiation characteristics suggest that the reported methodology, antenna type and substrate

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379

is a reasonable candidate for integration within flexible electronics.

\*Address all correspondence to: hrkhaleel@ualr.edu

posium (VTS), 2010 28th, 19-22 April , 84.

, Hussain M. Al-Rizzo and Ayman I. Abbosh

Department of Systems Engineering, University of Arkansas at Little Rock,, USA

[1] Hu, J. (2010). Overview of flexible electronics from ITRI's viewpoint. VLSI Test Sym‐

[2] Nathan, A., & Chalamala, B. R. (2005, July). Special Issue on Flexible Electronics Technology, Part 1: Systems and Applications. *Proceedings of the IEEE*, 93(7),

[3] Yongan, Huang, Chen, Jiankui, Yin, Zhouping, & Xiong, Youlun. (2011). Roll-to-Roll Processing of Flexible Heterogeneous Electronics With Low Interfacial Residual Stress. Components, Packaging and Manufacturing Technology IEEE Transactions

[4] Salonen, P., Kim , Jaehoon, & Rahmat-Samii, Y. (2004). Dual-band E-shaped patch wearable textile antenna. *IEEE Antennas and Propagation Society Symposium*, 1,

[5] Anagnostou, D. E., Gheethan, A. A., Amert, A. K., & Whites, K. W. (2010, Nov). A Direct-Write Printed Antenna on Paper-Based Organic Substrate for Flexible Dis‐

[6] Masahiro, Kubo., Xiaofeng, Li., Choongik, Kim., Michinao, Hashimoto., Wiley, Benja‐ min J., Ham, Donhee., & George, M. (2010). White sides Stretchable Microfl uidic Ra‐

plays and WLAN Applications. *Display Technology, Journal of*, 6(11), 558-564.

dio frequency Antennas,. *Wiley Inter science, Adv. Mater.*, 22, 2749-2752.

**Author details**

Haider R. Khaleel\*

**References**

1235-1238.

466-469.

on Sept., 1(9), 1368-1377.


**Table 2.** Comparative Study of Different Types of Flexible Antennas.

As shown in Table 2, the antenna reported in this chapter offers a relatively smaller size, highly robust and flexible design. Furthermore, the antenna is printable and provides low cost and roll to roll production.

#### **6. Conclusion**

In this chapter, the design, fabrication, and measurement of flexible antennas are discussed in details. Types of substrates and available fabrication methodologies for flexible antennas are reviewed. As a benchmark, a single band printed monopole antenna operating in the 2.45 GHz ISM band is presented which has the merits of light weight, ultra low profile, wide bandwidth, robustness, compactness, and high efficiency. The reported design is based on a Kapton Polyimide substrate which is known for its flexibility, robustness and low dielectric losses. The prototype was fabricated using the inkjet printing technology. Furthermore, the antenna is tested under bending effects since it is expected to be flexed or conformed on curved surfaces. Flexibility, robustness, compactness, fabrication simplicity along with good radiation characteristics suggest that the reported methodology, antenna type and substrate is a reasonable candidate for integration within flexible electronics.

### **Author details**

**5.2.4 Comparative Study**

teristics of the antenna under study.

378 Advancement in Microstrip Antennas with Recent Applications

**Character-istics Polyimide**

**Band/fₒ** Single/ 2.45

**Tensile strength** High (165

**Flexural strength** High (50000

**Dielectric loss**

**Fabrication complexity**

cost and roll to roll production.

**6. Conclusion**

**based antenna**

Low loss tan δ=0.002

> Simple/ printable

**Table 2.** Comparative Study of Different Types of Flexible Antennas.

The split ring antenna design is compared to different types of flexible antennas reported in [4]-[7]. Given the applications envisioned in this study, the comparative study is focused on compactness (size and thickness), electrical properties and robustness. Robustness encom‐ passes the major mechanical properties related to flexible/conformal electronic devices such as tensile strength, flexural strength, deformability, and thermal stability. Fabrication com‐ plexity criterion is also considered in this comparative study. Table 2 depicts these charac‐

**Size in mm** 38 x 27 180 x 150 46 x 35 65 x 10 39 x 25 **Thickness mm** 0.05 4 0.25 1 0.13

**Substrate** Polyimide ɛr=3.4 Felt fabric ɛr=1.5 Paper ɛr= 3.4 PDMS ɛr= 2.67 PEN film ɛr=3.2

**Deform-ability** Low High High High Low **Thermal stability** High Low Low Low High

As shown in Table 2, the antenna reported in this chapter offers a relatively smaller size, highly robust and flexible design. Furthermore, the antenna is printable and provides low

In this chapter, the design, fabrication, and measurement of flexible antennas are discussed in details. Types of substrates and available fabrication methodologies for flexible antennas are reviewed. As a benchmark, a single band printed monopole antenna operating in the

**Paper based antenna [5]**

GHz Dual/2.2, 3 GHz Single/2.4 GHz Variable Single/7.6 GHz

Medium loss tan δ=0.065

MPA) Low (2.7 MPA) Low (30 MPA) Low (3.9 MPA) High (74 MPA)

p.s.i) Low (8900 p.s.i) Low (7200 p.s.i) Low (650 p.s.i) High (13640

printable Simple/Printable Complex/Non-

**Fluidic antenna [6]**

High loss tan δ=0.37 **Flexible Bow-tie antenna [7]**

> Low loss tan δ=0.015

> > p.s.i)

printable Simple/Printable

**Textile antenna [4]**

> Low loss tan δ=0.02

Complex/Non-

Haider R. Khaleel\* , Hussain M. Al-Rizzo and Ayman I. Abbosh

\*Address all correspondence to: hrkhaleel@ualr.edu

Department of Systems Engineering, University of Arkansas at Little Rock,, USA

#### **References**


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[19] Janezic, M. D., Williams, D. F., Blaschke, V., Karamcheti, A., & Chi, Shih. Chang. (2003, Jan). Permittivity characterization of low-k thin films from transmission-line measurements. *Microwave Theory and Techniques, IEEE Transactions on*, 51(1), 132-136.

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[21] Soh, P. J., Vandenbosch, G. A. E., Soo, Liam Ooi, & Rais, N. H. M. (2012, Jan). Design of a Broadband All-Textile Slotted PIFA. *Antennas and Propagation, IEEE Transactions*

[22] Gamota, D. R., Brazis, P., Kalyanasundaram, K., & Zhang, J. (2004). Printed organic and molecular electronics. USA 2004, Kluwer Academic Publishers.

[23] Halonen, E., Kaija, K., Mantysalo, M., Kemppainen, A., Osterbacka, R., & Bjorklund, N. (2009, 15-18 June). Evaluation of printed electronics manufacturing line with sen‐ sor platform application. Paper presented at Microelectronics and Packaging Confer‐

[24] Kirsch, N. J., Vacirca, N. A., Plowman, E. E., Kurzweg, T. P., Fontecchio, A. K., & Dandekar, K. R. (2009, 27-28 April). Optically transparent conductive polymer RFID meandering dipole antenna. Paper presented at RFID, 2009 IEEE International Con‐

[25] Kirsch, N. J., Vacirca, N. A., Kurzweg, T. P., Fontecchio, A. K., & Dandekar, K. R. (2010, 11-13 Oct.) Performance of transparent conductive polymer antennas in a MIMO ad-hoc network. Paper presented at Wireless and Mobile Computing, Net‐ working and Communications (WiMob), 2010 IEEE 6th International Conference on.

[26] Leung, S. Y. Y., & Lam, D. C. C. (2007, July). Performance of Printed Polymer-Based RFID Antenna on Curvilinear Surface. *Electronics Packaging Manufacturing, IEEE*

[27] Kirschman, R. (1999). *Fabrication of Passive Components for High Temperature Instrumen‐*

[28] Mitzner, Kraig. (2009). Complete pcb design using orcad capture and pcb editor.

[29] Niir Board. (2011). *Handbook On Printing Technology (offset, Gravure, Flexo, Screen) 2nd*. [30] Lakafosis, V., Rida, A., Vyas, R., Li, Yang., Nikolaou, S., & Tentzeris, M. M. (2010, Sept). Progress Towards the First Wireless Sensor Networks Consisting of Inkjet-Printed, Paper-Based RFID-Enabled Sensor Tags. *Proceedings of the IEEE*, 98(9),

[31] Orecchini, G., Alimenti, F., Palazzari, V., Rida, A., Tentzeris, M. M., & Roselli, L. (2011, June 6). Design and fabrication of ultra-low cost radio frequency identification

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### *Edited by Ahmed Kishk*

The book discusses basic and advanced concepts of microstrip antennas, including design procedure and recent applications. Book topics include discussion of arrays, spectral domain, high Tc superconducting microstrip antennas, optimization, multiband, dual and circular polarization, microstrip to waveguide transitions, and improving bandwidth and resonance frequency. Antenna synthesis, materials, microstrip circuits, spectral domain, waveform evaluation, aperture coupled antenna geometry and miniaturization are further book topics. Planar UWB antennas are widely covered and new dual polarized UWB antennas are newly introduced. Design of UWB antennas with single or multi notch bands are also considered. Recent applications such as, cognitive radio, reconfigurable antennas, wearable antennas, and flexible antennas are presented. The book audience will be comprised of electrical and computer engineers and other scientists well versed in microstrip antenna technology.

Advancement in Microstrip Antennas with Recent Applications

Advancement in Microstrip

Antennas with Recent

Applications

*Edited by Ahmed Kishk*

Photo by lucato / iStock