**5.1. For ultra wideband application**

A tendency and well known ultra wideband application is the Digital Video Broadcasting-Handheld (DVB-H). Indeed, the allocated frequency range is divided in two sub-bands, i.e. [470 MHz – 790 MHz] and [790 MHz – 862 MHz]. Considering the entire DVB-H frequency range, it presents 58% of bandwidth and a few kind of antenna can cover it instantaneously while being miniature and having good performances.

As presented in previous parts, for a single-mode excitation, the DRAs bandwidth doesn't exceed 15%. Recently, different shapes and stacked resonators have been proposed to enhance the bandwidth. These techniques are generally difficult to implement without increasing the size of the antenna. A novel DRA design method is proposed in this part with detailed parametric studies [37]. After seeing the proposed DRA and its miniaturization technique, the study will focus on the bandwidth enhancement. In order to design and optimize the proposed antenna, both eigenmode solver and the Finite Integration Temporal method of CST Microwave Studio are used. The measurement and simulation results will be shown, followed by a discussion.

Previous parts have detailed the miniaturization technique by using a circular sector DRAs. It has been shown that resonance frequencies depend on whether parts of some faces are coated with a metal or not. According to the equation 7 and in order to easily integrate the antenna inside a terminal, *d* has to be lower than 25 mm. In this case, the lowest mathematical resonance frequency which can be obtained is 949MHz. The best compromise between the size and the desired frequency band is to use the HE½1 mode (Table 4). In order to entirely cover the DVB-H band, the non-coated face of the selected shape is transformed to a cubic part. The final designed DR shape is shown Figure 19.

**Figure 19.** Proposed DR shape (a) and metallization of two different sides (b)

As mentioned previously, coating some faces with a metal allows resonance frequencies decreasing. A further parametric study is performed in order to determine how to coat some faces with a metal and how to choose the resonator position on the ground plane. In the Figure 19, *a* and *b* are defined as the metallization lengths of two different metallic sides.

Figure 20 represents the variation of the first and the second resonance frequencies mode according to *a* and *b* lengths.

**Figure 20.** Variation of the resonance frequency of the 1st (a) and 2nd (b) modes according to a and b lengths

The resonance frequencies decrease significantly when the two metallic sides have the same potential and each one covers half of the face. So, the optimum metallization lengths are *a*=50mm and *b*=50mm and the two lowest resonance frequencies are 568MHz and 1GHz. Through the above results, the chosen structure looks like the Figure 19 with *a*=*b*=50mm. Now, it is necessary first of all to glance over the E-field distribution inside the DR. Figure 21 shows the E-field inside the DR for the first and the second mode respectively at 568 MHz and 1 GHz. The first mode is derivative from the HE½1 mode.

**Figure 21.** E fields of the two first modes

44 Dielectric Material

partial electric boundary on sides of DRAs.

while being miniature and having good performances.

to a cubic part. The final designed DR shape is shown Figure 19.

**Figure 19.** Proposed DR shape (a) and metallization of two different sides (b)

**5.1. For ultra wideband application** 

shown, followed by a discussion.

This part is divided in two sub-sections. The first one will focus on the bandwidth enhancement of a DRA for ultra wideband applications, and the second part will aim multiband applications. The common thread is the miniaturization of DRAs while obtaining good performances. For that, new hybrid modes will be studied with the application of

A tendency and well known ultra wideband application is the Digital Video Broadcasting-Handheld (DVB-H). Indeed, the allocated frequency range is divided in two sub-bands, i.e. [470 MHz – 790 MHz] and [790 MHz – 862 MHz]. Considering the entire DVB-H frequency range, it presents 58% of bandwidth and a few kind of antenna can cover it instantaneously

As presented in previous parts, for a single-mode excitation, the DRAs bandwidth doesn't exceed 15%. Recently, different shapes and stacked resonators have been proposed to enhance the bandwidth. These techniques are generally difficult to implement without increasing the size of the antenna. A novel DRA design method is proposed in this part with detailed parametric studies [37]. After seeing the proposed DRA and its miniaturization technique, the study will focus on the bandwidth enhancement. In order to design and optimize the proposed antenna, both eigenmode solver and the Finite Integration Temporal method of CST Microwave Studio are used. The measurement and simulation results will be

Previous parts have detailed the miniaturization technique by using a circular sector DRAs. It has been shown that resonance frequencies depend on whether parts of some faces are coated with a metal or not. According to the equation 7 and in order to easily integrate the antenna inside a terminal, *d* has to be lower than 25 mm. In this case, the lowest mathematical resonance frequency which can be obtained is 949MHz. The best compromise between the size and the desired frequency band is to use the HE½1 mode (Table 4). In order to entirely cover the DVB-H band, the non-coated face of the selected shape is transformed

(a) (b)

In the following, a ground plane is inserted on the lower metallic face as shown Figure 22. The ground plane size is 230mm x 130mm, chosen to correspond to a standard DVB-H handheld receiver. The DR of λ0/7xλ0/13xλ0/28 dimensions at 470 MHz is mounted on such ground plane.

Considering the E fields distribution, a probe is chosen to feed the DRA. It is placed on the lateral metallic side in order to simultaneously excite the two first modes (Figure 22).

Dielectric Materials for Compact Dielectric Resonator Antenna Applications 47

**0.4 0.5 0.6 0.7 0.8 0.9 <sup>1</sup> 1.1 -35**

**Frequency (GHz)**

**Measured S11 Simulated S11**

The input impedance and the coefficient reflection are shown Figure 25.

**Figure 25.** Measured and simulated input impedances (a) and S11 parameters (b)

the proposed antenna are illustrated Figure 26.

**0.4 0.5 0.6 0.7 0.8 0.9 1 1.1**

**Frequency (GHz)**

and measured radiation patterns at 620MHz and 870MHz are shown Table 5.

efficiencies. The measured efficiency remains higher than 75% above 600MHz.

Radiation patterns of the antenna were characterized in an anechoic chamber. Simulated

(a) (b)

**-30 -25 -20 -15 -10 -5 0**

**-8**

**S11 (dB)**

**Simulated Re(Zin) Simulated Im(Zin) Measured Re(Zin) Measured Im(Zin)**

It can be noticed that the *xz*-plane and *yz*-plane radiation patterns indicate a correct omnidirectional radiation patterns. The antenna operates at two different modes inside the operating band, so the antenna doesn't offer the same radiation pattern at the two resonance frequencies, particularly in the *xy*-plane. The simulated and measured total efficiencies of

This efficiency disagreement is probably due to the coaxial feeding cable, the discrepancy between the simulated and measured reflection coefficient and the dielectric loss tangent.

In spite of this 10% difference, there is a good agreement between measured and simulated

MHz – 1.05 GHz].

**Figure 24.** Antenna prototype

**-50**

**0**

**50**

**Input impedance Zin ()**

**100**

There is a good agreement between simulations and measurements. It should be underlined that these simulations series take into account that the probe is inserted in a 2 mm diameter hole instead of a 1.5 mm hole due to mechanical constraint, therefore an air gap appears around the probe. This air gap results in lowering the DRA effective dielectric constant, which is turn lower the Q-factor accompanied by a shift in the resonance frequency. Furthermore, after material characterizations, the ceramic's dielectric constant turns out equal to 9.5 at 500 MHz. So, the -8dB impedance matching bandwidth is included in [540

To reach an ultra wideband, the probe position, diameter and length are tuned. The optimal probe diameter and length values are 1.5mm and 49mm.

Following this optimization, input impedance and return loss are shown in the Figure 23.

**Figure 22.** Antenna design fed by a coaxial probe

**Figure 23.** Input impedance (a) and S11 parameter (b) of the considered DRA

They show that two modes exist at 0.6 MHz and 0.93 MHz, excited by the coaxial probe. As expected, a difference appears between resonance frequency values which are obtained by the eigenmode solver and electromagnetic simulation. This discrepancy is due to the presence of the probe and the ground plane. Furthermore, the boundary conditions applied during the modal analysis are perfect electric or magnetic conditions on the DR walls contrarily to the electromagnetic simulation.

The simulated coefficient reflection shows 70% of bandwidth for a -8 dB impedance bandwidth definition over the frequency range [466 MHz – 935 MHz]. The -8dB impedance bandwidth definition is sufficient to achieve a good efficiency. Thus, the antenna satisfies the DVB-H system specifications mentioned previously. To demonstrate the antenna performances, the antenna has been realized with a ceramic material. It is fed by a probe and is half-mounted on a ground plane. The manufactured antenna is shown Figure 24.

The input impedance and the coefficient reflection are shown Figure 25.

There is a good agreement between simulations and measurements. It should be underlined that these simulations series take into account that the probe is inserted in a 2 mm diameter hole instead of a 1.5 mm hole due to mechanical constraint, therefore an air gap appears around the probe. This air gap results in lowering the DRA effective dielectric constant, which is turn lower the Q-factor accompanied by a shift in the resonance frequency. Furthermore, after material characterizations, the ceramic's dielectric constant turns out equal to 9.5 at 500 MHz. So, the -8dB impedance matching bandwidth is included in [540 MHz – 1.05 GHz].

**Figure 24.** Antenna prototype

46 Dielectric Material

Considering the E fields distribution, a probe is chosen to feed the DRA. It is placed on the

To reach an ultra wideband, the probe position, diameter and length are tuned. The optimal

Following this optimization, input impedance and return loss are shown in the Figure 23.

**λ0=/28 230 mm**

**Re(Ze) Im(Ze)**

**Feeding probe ε=10**

**130 mm**

**0.4 0.5 0.6 0.7 0.8 0.9 <sup>1</sup> 1.1 -20**

**0.466 0.935**

**Frequency (GHz)**

lateral metallic side in order to simultaneously excite the two first modes (Figure 22).

probe diameter and length values are 1.5mm and 49mm.

**λ0=/13**

**λ0=/7**

**Figure 23.** Input impedance (a) and S11 parameter (b) of the considered DRA

They show that two modes exist at 0.6 MHz and 0.93 MHz, excited by the coaxial probe. As expected, a difference appears between resonance frequency values which are obtained by the eigenmode solver and electromagnetic simulation. This discrepancy is due to the presence of the probe and the ground plane. Furthermore, the boundary conditions applied during the modal analysis are perfect electric or magnetic conditions on the DR walls

(a) (b)

**-15 -10 -5 0**

**-8**

**S11 (dB)**

The simulated coefficient reflection shows 70% of bandwidth for a -8 dB impedance bandwidth definition over the frequency range [466 MHz – 935 MHz]. The -8dB impedance bandwidth definition is sufficient to achieve a good efficiency. Thus, the antenna satisfies the DVB-H system specifications mentioned previously. To demonstrate the antenna performances, the antenna has been realized with a ceramic material. It is fed by a probe and is half-mounted on a ground plane. The manufactured antenna is shown Figure 24.

**Figure 22.** Antenna design fed by a coaxial probe

**-50**

**0**

**Input impedance Zin ()**

**50**

**100**

**0.4 0.5 0.6 0.7 0.8 0.9 1 1.1**

**Frequency (GHz)**

contrarily to the electromagnetic simulation.

**Figure 25.** Measured and simulated input impedances (a) and S11 parameters (b)

Radiation patterns of the antenna were characterized in an anechoic chamber. Simulated and measured radiation patterns at 620MHz and 870MHz are shown Table 5.

It can be noticed that the *xz*-plane and *yz*-plane radiation patterns indicate a correct omnidirectional radiation patterns. The antenna operates at two different modes inside the operating band, so the antenna doesn't offer the same radiation pattern at the two resonance frequencies, particularly in the *xy*-plane. The simulated and measured total efficiencies of the proposed antenna are illustrated Figure 26.

This efficiency disagreement is probably due to the coaxial feeding cable, the discrepancy between the simulated and measured reflection coefficient and the dielectric loss tangent.

In spite of this 10% difference, there is a good agreement between measured and simulated efficiencies. The measured efficiency remains higher than 75% above 600MHz.

This part has proposed a novel dielectric resonator antenna. Parametric studies have been realized to decrease resonance frequencies and to increase impedance matching bandwidth. The results have shown that by shaping the dielectric structure and coating some faces with a metal, resonance frequencies have been reduced for a fixed structure size. Good performances have been obtained and the proposed antenna can be used for DVB-H and/or GSM900 applications with a wide bandwidth and a good efficiency.

Dielectric Materials for Compact Dielectric Resonator Antenna Applications 49

Recently, many studies have been devoted to multi-band antennas [38-41], some of them dealing with DRAs [42-44]. A dielectric resonator indeed supports more than one resonant mode at two close frequencies, which allows them to meet the requirements of different applications with a unique device. Some studies furthermore use both the dielectric resonator and the feeding mechanism as radiator elements [43-45]. This explains why the DRAs present a major advantage for multi-standard devices, when compared to other kinds

The objective of this part is to show the integration of a small antenna in a multi-band mobile handheld device, working on the nine channels of the second sub-band of the DVB-H, i.e. [790 MHz – 862 MHz], the WiFi band at 2.4 GHz and the WiMax band at 3.5 GHz. Additionally, in order to improve the quality and the reliability of the wireless links, i.e. obtain pattern diversity, two antennas will be integrated in the same device. Setting maximum limits, we have decided to integrate two orthogonally aligned antennas in the allocated space of 30 mm x 41 mm (λ0/13 x λ0/9 at 800 MHz), on a 230 mm x 130 mm ground plane. Each DRA will therefore have to be very compact to be able to fit in such a limited area, and will operate around 850 MHz, 2.4 GHz and 3.5 GHz, thus covering the nine

Firstly, only one radiating element is described. It will be done thanks to a modal analysis [46] of the dielectric resonator. The DRA design and the choice of the dielectric permittivity will be discussed. As for the previous sub-section, both the Finite Integration Temporal method and the eigenmode solver of CST Microwave Studio were used to carry out this

Getting three resonant frequencies for a single dielectric resonator is aiming. Firstly, the resonator has to be integrated in a handheld receiver, which means that it will be placed on a FR4 substrate (εr=4.9). Secondly, to be integrated in a handheld device, the allocated space

41 mm

(a) (b) (c)

**Figure 27.** Top view of the defined PCB card (a), allocated size for the antenna system (b) and bottom

The dimensions and dielectric permittivity of each resonator need to be chosen according to

As a result, the dimensions of one resonator were chosen to be 25 mm x 10 mm x 4 mm, with

a very high dielectric constant r of 37. The resulting geometry is depicted Figure 28.

FR4 substrate

for the antenna system must not exceed 41mm x 30 mm x 4mm as shown Figure 27.

30 mm

these constraints, to ensure the integration of the antenna in the final device.

channels of the DVB-H band, the WiMax and the WiFi band.

130 mm

230 mm

view of the defined PCB card (c)

of antennas.

work.

**Table 5.** Simulated (blue line) and measured (red line) radiation patterns at 620 MHz and 870 MHz

**Figure 26.** Measured and simulated total efficiencies of the considered DRA

### **5.2. For multiband application**

In the last decade, the huge demand for mobile and portable communication systems has led to an increased need for more compact antenna designs. This aspect is even more critical when several wireless technologies have to be integrated on the same mobile wireless communicator. All the new services and the increased user density are driving the antenna design toward multi-band operation.

Recently, many studies have been devoted to multi-band antennas [38-41], some of them dealing with DRAs [42-44]. A dielectric resonator indeed supports more than one resonant mode at two close frequencies, which allows them to meet the requirements of different applications with a unique device. Some studies furthermore use both the dielectric resonator and the feeding mechanism as radiator elements [43-45]. This explains why the DRAs present a major advantage for multi-standard devices, when compared to other kinds of antennas.

48 Dielectric Material

f=620 MHz

f=870 MHz

This part has proposed a novel dielectric resonator antenna. Parametric studies have been realized to decrease resonance frequencies and to increase impedance matching bandwidth. The results have shown that by shaping the dielectric structure and coating some faces with a metal, resonance frequencies have been reduced for a fixed structure size. Good performances have been obtained and the proposed antenna can be used for DVB-H and/or

**Table 5.** Simulated (blue line) and measured (red line) radiation patterns at 620 MHz and 870 MHz

**0.5 0.6 0.7 0.8 0.9 1 1.1**

**Measured Simulated**

**Frequency (GHz)**

In the last decade, the huge demand for mobile and portable communication systems has led to an increased need for more compact antenna designs. This aspect is even more critical when several wireless technologies have to be integrated on the same mobile wireless communicator. All the new services and the increased user density are driving the antenna

**0.4 0.5 0.6 0.7 0.8 0.9 1**

**Figure 26.** Measured and simulated total efficiencies of the considered DRA

**Total efficiency (x100) (%)**

**5.2. For multiband application** 

design toward multi-band operation.

φ=0° φ=90° θ=90°

GSM900 applications with a wide bandwidth and a good efficiency.

The objective of this part is to show the integration of a small antenna in a multi-band mobile handheld device, working on the nine channels of the second sub-band of the DVB-H, i.e. [790 MHz – 862 MHz], the WiFi band at 2.4 GHz and the WiMax band at 3.5 GHz. Additionally, in order to improve the quality and the reliability of the wireless links, i.e. obtain pattern diversity, two antennas will be integrated in the same device. Setting maximum limits, we have decided to integrate two orthogonally aligned antennas in the allocated space of 30 mm x 41 mm (λ0/13 x λ0/9 at 800 MHz), on a 230 mm x 130 mm ground plane. Each DRA will therefore have to be very compact to be able to fit in such a limited area, and will operate around 850 MHz, 2.4 GHz and 3.5 GHz, thus covering the nine channels of the DVB-H band, the WiMax and the WiFi band.

Firstly, only one radiating element is described. It will be done thanks to a modal analysis [46] of the dielectric resonator. The DRA design and the choice of the dielectric permittivity will be discussed. As for the previous sub-section, both the Finite Integration Temporal method and the eigenmode solver of CST Microwave Studio were used to carry out this work.

Getting three resonant frequencies for a single dielectric resonator is aiming. Firstly, the resonator has to be integrated in a handheld receiver, which means that it will be placed on a FR4 substrate (εr=4.9). Secondly, to be integrated in a handheld device, the allocated space for the antenna system must not exceed 41mm x 30 mm x 4mm as shown Figure 27.

**Figure 27.** Top view of the defined PCB card (a), allocated size for the antenna system (b) and bottom view of the defined PCB card (c)

The dimensions and dielectric permittivity of each resonator need to be chosen according to these constraints, to ensure the integration of the antenna in the final device.

As a result, the dimensions of one resonator were chosen to be 25 mm x 10 mm x 4 mm, with a very high dielectric constant r of 37. The resulting geometry is depicted Figure 28.

Dielectric Materials for Compact Dielectric Resonator Antenna Applications 51

The length is set to have the first resonance at 800 MHz

**Width of the printed line**

**Figure 30.** : Resonance frequency of the first dielectric resonator mode according to the width and the

Same studies have been performed for different shapes of the monopole. All these studies allowed the shape, length and width of the monopole to be set, which led to the final design of the dielectric resonator. Table 6 shows the values of the resonance frequencies for the first

three modes inside the resonator, which were obtained through the modal analysis.

now be referred as the "monopole", will be set to obtain the first resonance frequency

 This being done, the second and third band will be covered by the resonances of the dielectric resonator disturbed by the presence of the monopole. As previously explained, the length, width and shape of this monopole entail a modification of the DRA resonance frequencies. The shape and width of the monopole will therefore be optimized to obtain the desired resonance frequencies, while its length remains set by

 An important point concerns the radiation Q-factor. The high dielectric permittivity involves a high Q-factor [7]. With such an r, the radiation Q-factor is also important, making it difficult to obtain a wide impedance bandwidth for a given mode. So, to have a suitable impedance bandwidth, the antenna will have to be matched between two peaks of the real part of input impedance. By this way, the resonances won't have to be close to the operating bands. The goal being to match the DRA over the WiFi and WiMax bands, the monopole (its shape and width) has to be optimized to obtain

 A modal analysis has been performed to show the variations of the resonance frequencies of the first three modes, according to the shape and width of the monopole. Figure 30 shows the resonance frequency of the first mode according to the monopole geometry. With the first resonance frequency set to match the antenna at 2.4 GHz and the length of the monopole set to have the first resonance at 800 MHz, the graph of the modal analysis allows an easy determination of the monopole width, which is 1 mm.

resonance frequencies around 2 GHz, 2.8 GHz and 4 GHz.

First desired resonance frequency to match the antenna at 2.4 GHz

length of the monopole previously defined

around 800 MHz.

the first resonance.

**Figure 28.** Dimensions and properties of one resonator (a) and E-field distribution of the first natural mode for the resonator in (b) in the z=2mm plane

The first natural mode of this resonator is the TE11δ (Figure 28) which resonates at 3.99 GHz. This resonance frequency remains too high for the intended applications. It has however been shown [37], that this resonance frequency depends on the metallization of the DRA's faces. It is therefore necessary to envision the feeding mechanism of the resonator, before performing the modal analysis. Indeed, if the antenna is not fed by proximity coupling, the excitation cannot be ignored during the modal analysis. Furthermore, it can be used to adjust the resonance frequency of the resonator, especially in the case of an electrically small DRA. In this study, the feeding will indeed play a preeminent role. The chosen excitation is a line printed on the FR4 substrate and positioned under the dielectric resonator as shown Figure 29. The E-field distribution of the first mode (Figure 29) is completely different from the one obtained without this line. Indeed, as stated before, the E-field and H-field distributions inside the DRA depend on the boundary conditions on its faces. The feeding line introduces partially perfect electric conducting conditions, which disturb the field distribution inside the resonator when compared with the TE11δ mode. As a result, the resonance frequency of each mode will vary in accordance with the length and width of the feeding line (defined in the Figure 29).

**Figure 29.** Design of the DRA fed by the printed line and E-field distribution of the first natural mode for the resonator fed by the line

Considering the previous study, the resonator design method is as follows.

 First of all, in spite of the high dielectric permittivity, there is no mode around 800 MHz. In order to allow the antenna to operate on the nine channels of the DVB-H band, the printed line will be designed to resonate around 800 MHz. It will therefore behave like a printed monopole loaded by a dielectric. The length of this line, which will from now be referred as the "monopole", will be set to obtain the first resonance frequency around 800 MHz.

50 Dielectric Material

dielectric permittivity ε=37

25 mm

mode for the resonator in (b) in the z=2mm plane

feeding line (defined in the Figure 29).

for the resonator fed by the line

10 mm

4 mm

FR4 substrate

length

**Figure 29.** Design of the DRA fed by the printed line and E-field distribution of the first natural mode

 First of all, in spite of the high dielectric permittivity, there is no mode around 800 MHz. In order to allow the antenna to operate on the nine channels of the DVB-H band, the printed line will be designed to resonate around 800 MHz. It will therefore behave like a printed monopole loaded by a dielectric. The length of this line, which will from

FR4

Considering the previous study, the resonator design method is as follows.

Printed line

width

**Figure 28.** Dimensions and properties of one resonator (a) and E-field distribution of the first natural

The first natural mode of this resonator is the TE11δ (Figure 28) which resonates at 3.99 GHz. This resonance frequency remains too high for the intended applications. It has however been shown [37], that this resonance frequency depends on the metallization of the DRA's faces. It is therefore necessary to envision the feeding mechanism of the resonator, before performing the modal analysis. Indeed, if the antenna is not fed by proximity coupling, the excitation cannot be ignored during the modal analysis. Furthermore, it can be used to adjust the resonance frequency of the resonator, especially in the case of an electrically small DRA. In this study, the feeding will indeed play a preeminent role. The chosen excitation is a line printed on the FR4 substrate and positioned under the dielectric resonator as shown Figure 29. The E-field distribution of the first mode (Figure 29) is completely different from the one obtained without this line. Indeed, as stated before, the E-field and H-field distributions inside the DRA depend on the boundary conditions on its faces. The feeding line introduces partially perfect electric conducting conditions, which disturb the field distribution inside the resonator when compared with the TE11δ mode. As a result, the resonance frequency of each mode will vary in accordance with the length and width of the

(a) (b)


**Figure 30.** : Resonance frequency of the first dielectric resonator mode according to the width and the length of the monopole previously defined

Same studies have been performed for different shapes of the monopole. All these studies allowed the shape, length and width of the monopole to be set, which led to the final design of the dielectric resonator. Table 6 shows the values of the resonance frequencies for the first three modes inside the resonator, which were obtained through the modal analysis.


Dielectric Materials for Compact Dielectric Resonator Antenna Applications 53

disposed in a 30 mm x 41 mm area, both fed by a printed line acting as a monopole. Each line is fed by a 50Ω coaxial cable. They are studied on a 230 mm x 130 mm ground plane, as defined by the specifications. It will be shown in a following section that the antenna matching is not affected by the ground plane dimensions. In order to ascertain the

The resonators are manufactured with a ceramic material with a dielectric permittivity of 37 and a loss tangent tanδ=0.005 on the 0.5-10 GHz band. During the simulation and measurements, each resonator has been excited by a printed line fed by a coaxial cable.

During the manufacturing process of the antenna, a special care has to be given to minimize the air gap between the excitation and the resonator. In the case of this prototype, the

(a) (b)

The comparison between the simulated and measured results is now studied. The first ones have been obtained using the transient solver of CST Microwave Studio, while the measurements have been performed inside an anechoic chamber. Both the simulated S11

(a) (b)

**-30 -25 -20 -15 -10 -5 0**

**S22 (dB)**

**0.5 <sup>1</sup> 1.5 <sup>2</sup> 2.5 <sup>3</sup> 3.5 <sup>4</sup> -35**

**Frequency (GHz)**

**Measured Simulated**

**Measured Simulated**

performances of this antenna, a prototype has been fabricated as shown Figure 32.

resonators have been pressed onto the PCB to avoid this air gap.

**41 mm 30 mm**

**Figure 32.** Final simulated design (a) and the corresponding prototype (b)

and S22 parameters are compared with the measured ones Figure 33.

**Figure 33.** Measured and simulated S11 (a) and S22 (b) parameters of the DRA

**0.5 <sup>1</sup> 1.5 <sup>2</sup> 2.5 <sup>3</sup> 3.5 <sup>4</sup> -35**

**Frequency (GHz)**

Measured and simulated performances of the antenna

**Dimensions : 25x10x4 mm** 

Input1

**=37 tan =5.10-3**

**Dimensions of the ground plane : 230x130 mm**

Input 2

**-30 -25 -20 -15 -10 -5 0**

**S11 (dB)**

**Table 6.** Values of Resonance Frequency for the Three First Modes

Electromagnetic study of the DRA

In order to validate the previous modal study, the dielectric resonator, placed in the area dedicated to the antenna (Figure 27) and fed by a 50Ω discrete port, has been simulated with the FIT method using CST Microwave Studio. It must be noticed that a discrete port is modeled by a lumped element, consisting of a current source with a 50 Ω inner impedance that excites and absorbs power.

The Figure 31 shows the simulated input impedance of the dielectric resonator with its feed. The resonance frequencies are in agreement with the modal analysis. The radiation Q-factor is important, and the input impedance variations confirm that the antenna matching is easier between two resonances. The reflection coefficient is shown Figure 31. The antenna is matched on all of the desired bands, i.e. the nine channels of DVB-H going from 790 MHz to 862 MHz, the WiFi band at 2.4 GHz and the WiMax band at 3.5 GHz. It can be noted that the matching over the first band is obtained due to the resonance of the λ/4 monopole.

**Figure 31.** Input impedance (a) and S11 parameter of the fed DRA

The dielectric resonator must now be integrated in its context, i. e. on a 230 mm x 130 mm ground plane, chosen to correspond to a standard DVB-H handheld receiver. As explained before, another specification was to obtain a reconfigurable radiation pattern. Thus, two instances of the previously studied resonator are orthogonally integrated on the ground plane.

Final structure

Based on the previous parametric study, the final structure has been designed as shown Figure 32. In order to obtain pattern diversity, two dielectric resonators are orthogonally disposed in a 30 mm x 41 mm area, both fed by a printed line acting as a monopole. Each line is fed by a 50Ω coaxial cable. They are studied on a 230 mm x 130 mm ground plane, as defined by the specifications. It will be shown in a following section that the antenna matching is not affected by the ground plane dimensions. In order to ascertain the performances of this antenna, a prototype has been fabricated as shown Figure 32.

The resonators are manufactured with a ceramic material with a dielectric permittivity of 37 and a loss tangent tanδ=0.005 on the 0.5-10 GHz band. During the simulation and measurements, each resonator has been excited by a printed line fed by a coaxial cable.

During the manufacturing process of the antenna, a special care has to be given to minimize the air gap between the excitation and the resonator. In the case of this prototype, the resonators have been pressed onto the PCB to avoid this air gap.

**Figure 32.** Final simulated design (a) and the corresponding prototype (b)

Measured and simulated performances of the antenna

52 Dielectric Material

plane.

Final structure

**Input impedance Zin**

Mode Resonance frequency

In order to validate the previous modal study, the dielectric resonator, placed in the area dedicated to the antenna (Figure 27) and fed by a 50Ω discrete port, has been simulated with the FIT method using CST Microwave Studio. It must be noticed that a discrete port is modeled by a lumped element, consisting of a current source with a 50 Ω inner impedance

The Figure 31 shows the simulated input impedance of the dielectric resonator with its feed. The resonance frequencies are in agreement with the modal analysis. The radiation Q-factor is important, and the input impedance variations confirm that the antenna matching is easier between two resonances. The reflection coefficient is shown Figure 31. The antenna is matched on all of the desired bands, i.e. the nine channels of DVB-H going from 790 MHz to 862 MHz, the WiFi band at 2.4 GHz and the WiMax band at 3.5 GHz. It can be noted that the

The dielectric resonator must now be integrated in its context, i. e. on a 230 mm x 130 mm ground plane, chosen to correspond to a standard DVB-H handheld receiver. As explained before, another specification was to obtain a reconfigurable radiation pattern. Thus, two instances of the previously studied resonator are orthogonally integrated on the ground

**-15**

**-10**

**S11 (dB)**

**-5**

**0**

**0.5 <sup>1</sup> 1.5 <sup>2</sup> 2.5 <sup>3</sup> 3.5 <sup>4</sup> -20**

**Frequency (GHz)**

(a) (b)

Based on the previous parametric study, the final structure has been designed as shown Figure 32. In order to obtain pattern diversity, two dielectric resonators are orthogonally

matching over the first band is obtained due to the resonance of the λ/4 monopole.

**Re (Zin) Im (Zin)**

First mode 1.969 GHz Second mode 2.773 GHz Third mode 4.135 GHz

**Table 6.** Values of Resonance Frequency for the Three First Modes

**Figure 31.** Input impedance (a) and S11 parameter of the fed DRA

**1.5 <sup>2</sup> 2.5 <sup>3</sup> 3.5 <sup>4</sup> 4.5 -800**

**Frequency (GHz)**

**1.86 2.75 4.12**

**Antenna matching**

Electromagnetic study of the DRA

that excites and absorbs power.

The comparison between the simulated and measured results is now studied. The first ones have been obtained using the transient solver of CST Microwave Studio, while the measurements have been performed inside an anechoic chamber. Both the simulated S11 and S22 parameters are compared with the measured ones Figure 33.

**Figure 33.** Measured and simulated S11 (a) and S22 (b) parameters of the DRA

#### 54 Dielectric Material

The measurements and simulations are in very good agreement. Moreover, the antenna is matched over all the desired bands and for both inputs.

Dielectric Materials for Compact Dielectric Resonator Antenna Applications 55

To conclude, an affordable chapter has been presented allowing the reader to find an overview of main DRA shapes, properties and approaches while appreciating the influence and the impact of the dielectric material properties. Indeed, a broad spectrum of dielectric materials can be used depending on the intended application. In addition to the advantages common to all DRAs described at the beginning of this chapter, a dedicated part has focused on other advantages of compact DRAs, which are desirable for many emerging wireless and mobile communication systems. Finally, a specific part had presented relevant data for postgraduate researchers, antenna design engineers in general and particularly the ones engaged in the innovative design of mobile and wireless systems by focusing on the hybrid

modes creation to enhance the bandwidth or develop multiband antennas.

[1] R. D. Richtmyer, "Dielectric Resonator", J. Appl. Phys., vol. 10, pp. 391-398, Jun. 1939 [2] D. Kajfez and P. Guillon, Eds., Dielectric Resonators. Norwood, MA: Artech House,

[3] Cohn, S.B., "Microwave Bandpass Filters Containing High-Q Dielectric Resonators," Microwave Theory and Techniques, IEEE Transactions on, vol.16, no.4, pp. 218- 227,

[4] S. J. Fiedziuszko, "Microwave Dielectric Resonators'', Microwave Journal, vol. 29,

[5] S.A. Long, M.W. McAllister and L.C. Shen, "The Resonant Dielectric Cavity Antenna", IEEE Transactions on Antennas and Propagation, Vol. 31, n°3, March 1983, pp. 406-412 [6] A. Petosa, A. Ittipiboon, Y.M.M. Antar and D. Roscoe, "Recent Advances in Dielectric Resonator Antenna Technology", IEEE Antennas and Propagation Magazine, Vol. 40,

[7] K.M Luk and K.W Leung, "Dielectric Resonator Antennas", Electronic & Electrical

[8] R.K Mongia and A. Ittipiboon, "Theoretical And Experimental Investigations on Rectangular Dielectric Resonator Antenna", IEEE Transactions on Antennas and

[9] D. Drossos, Z. Wu and L.E. Davis, "Theoretical and experimental investigation of cylindrical Dielectric Resonator Antennas", Microwave and Optical Technology Letters,

[10] R. K. Mongia and P. Bhartia, "Dielectric Resonator Antennas – A review and General Design Relations for resonant Frequency and Bandwidth", International Journal of

Propagation, Vol. 45, n°9, September 1997, pp. 1348-1356

**6. Conclusion** 

**Author details** 

**7. References** 

1986

Apr 1968

L. Huitema and T. Monediere

*University of Limoges, Xlim Laboratory, France* 

September 1986, pp 189-200

n°3, 06/1998, pp. 35-48

Engineering Research Studies

Vol. 13, No. 3, pp. 119-123, October 1996

Radiation patterns: The radiation patterns have been measured inside an anechoic chamber. Table 7 shows the 3D simulated radiation patterns for both inputs at 830 MHz, 2.4 GHz and 3.5 GHz.

It can be seen that the radiation pattern at a given frequency will depend on the excited port. While promising, this result is not sufficient to conclude that the radiation pattern is reconfigured. More details are explained in [47] (this requires the characterization of the whole system in a reverberation chamber, in order to determine the correlation coefficient).

Thus, this study started with modal analyses, which allowed the shape and dimensions of the antenna's excitation to be defined with a dual objective. Indeed, this line had first to behave like a monopole and cover the nine channels of the DVB-H band (going from 790 MHz to 862 MHz). Secondly, it had to excite the dielectric resonator and set its resonance frequencies so as to match the antenna on the WiFi and WiMax bands.

After performing these preliminary studies, two instances of the conceived dielectric resonator have been orthogonally integrated on a 230 mm x 130 mm ground plane, which is consistent with a tablet. Finally, the antenna system, which only occupies a 30 mm x 41 mm area, is matched on the three desired bands, i.e. the nine channels of the DVB-H band, the WiFi band and the WiMax band with pattern diversity.

This part has presented the design method, the realization and the measurement of a two compact DRAs, one for ultra wideband application and the second for multiband applications.


**Table 7.** 3D Radiation patterns at 830 MHz, 2.4 GHz and 3.5 GHz for the two inputs
