**4.1. Single-ended bistatic radar system**

At first a single-ended radar demonstrator was developed in the project, with which the potential of impulse-radio UWB sensing is evaluated. The demonstrator is built combining commercially available components for the low frequency operation control and components tailored for UWB operation dealing with the signals in the 3.1-10.6 GHz band. The UWB components are developed and fabricated in the aforementioned SiGe HBT production technology. A block diagram of the single-ended bistatic radar system is depicted in Fig. 40. The sensor uses separate antennas for transmitter and receiver to avoid losses due to power divider structures on the feedline of a single transceiver antenna. Besides a heavy crosstalk into the low-noise amplifier (LNA) of the receiver is avoided by using two antennas.

An ultra-broadband Vivaldi antenna is chosen, which consists of an exponentially tapered slot on a microstrip substrate. The transition from microstrip to slot line is done by a Marchand balun, as discussed in [22]. On the feeding line of the transmit antenna an impulse generator IC is mounted, which emits an impulse with a shape very similar to the fifth derivative of a Gaussian bell shape with a standard deviation *σ* = 51 ps. This impulse shape fits into 472 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications <sup>35</sup> 473 UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications

**Figure 40.** Block diagram of the single-ended bistatic radar system.

34 Will-be-set-by-IN-TECH

surface is presented in Fig. 39. The times of flight between the transmitter at (5 cm,5 cm) and the individual elements of the receiver array at *z* = -20 cm are calculated using electromagnetic

−10 0 10 20 30

**Figure 39.** Calculated wavefronts in a medium with *ε*r=10 based on EM field simulations. The receiving

The above mentioned 2D localization procedure leads to a belt of refracted wavefronts with a focussing point where the receiver has been placed in the simulation. To decrease calculation time it is also possible to search for the narrowest point in the wavefront belt instead of calculating the intersections. This leads to an estimated transmitter position of

It is obvious that in the example of Fig. 39 a smaller number of wavefronts and thus less receiving antennas would be sufficient for a successful localization of the transmitter. But in practical applications this high number of antennas might still be required as a dense sensor array is rather needed for surface estimation than for solving the localization problem.

At first a single-ended radar demonstrator was developed in the project, with which the potential of impulse-radio UWB sensing is evaluated. The demonstrator is built combining commercially available components for the low frequency operation control and components tailored for UWB operation dealing with the signals in the 3.1-10.6 GHz band. The UWB components are developed and fabricated in the aforementioned SiGe HBT production technology. A block diagram of the single-ended bistatic radar system is depicted in Fig. 40. The sensor uses separate antennas for transmitter and receiver to avoid losses due to power divider structures on the feedline of a single transceiver antenna. Besides a heavy crosstalk

into the low-noise amplifier (LNA) of the receiver is avoided by using two antennas.

An ultra-broadband Vivaldi antenna is chosen, which consists of an exponentially tapered slot on a microstrip substrate. The transition from microstrip to slot line is done by a Marchand balun, as discussed in [22]. On the feeding line of the transmit antenna an impulse generator IC is mounted, which emits an impulse with a shape very similar to the fifth derivative of a Gaussian bell shape with a standard deviation *σ* = 51 ps. This impulse shape fits into

antennas are placed along the *x*-axis at *z*=-20, the transmitter is positioned at (5 cm,5 cm).

x / cm

field simulation software [6].

air

−20

(4.60 cm,4.94 cm), having an error of about 4 mm.

**4. Systems design and measurement results**

**4.1. Single-ended bistatic radar system**

−10

−5 / cm z

−15

5

0

15

10

medium

the 3.1-10.6 GHz UWB spectral mask allocated by the FCC in the United States and has a voltage amplitude of 600 mV peak-to-peak [8]. The impulse generator radiates an impulse at every rising slope of an input trigger signal. Here a sinusoidal signal is used to trigger the impulse generator, so at every rising edge of the sinusoidal signal an impulse is emitted which results in a continuous impulse train. The sinusoidal signal is supplied from one source of the direct-digital-synthesizer (DDS) AD9959. All four clock sources of the AD9959 (three are used) are synchronized among each other to allow a phase and frequency stable operation between the signals. The transmitter is adjusted to generate an impulse train with a repetition rate of *frep* = 200 MHz. To reduce impulse-to-impulse jitter, spurious emissions of the 200 MHz sinusoidal trigger signal are filtered by a narrowband helix filter. The generated impulse train is continuously radiated by the antennas, is reflected at the desired object and enters the receiver. The reflection at the object causes a phase inversion to the impulses, therefore it is received with inverted amplitude. Additionally the impulse train is fed to the receiver by a direct and non-inversed coupling between the antennas.

**Figure 41.** Picture of (a) Vivaldi transmit antenna with mounted impulse generator IC and (b) Vivaldi receive antenna with correlator IC and baseband circuit.

These signals are processed in the receiver by a monolithic correlator IC, which consist of a UWB LNA, a four-quadrant multiplier, a template impulse generator generating a fifth Gaussian derivative impulse corresponding to the transmit impulse, and a first integrating low-pass filter with a cut-off frequency of 800 MHz [9]. The template impulse generator is driven by a second clock source of the AD9959 at a repetition rate of

*frep* − Δ*f* = 200 MHz - 25 Hz. This sinusoidal clock signal is filtered as well with a helix filter to improve jitter performance. The output signal of the correlator IC is processed with a baseband circuit, where it is amplified and further integrated by a low-pass filter with a cut-off frequency of 25 kHz. After this it is fed to a data-logger, which samples the generated signals with a sampling period of 30 μs and transfers them to a PC for further processing. Additionally a synchronizing signal of Δ*f* = 25 Hz from the third DDS clock signal is sampled

method is illustrated in Fig. 42 by showing the signals appearing in the receiver. In the upper trace of Fig. 42(a) the received signal at the output of the LNA in the correlator IC is shown. An impulse from both, the non-inverted direct coupling and the inverted reflection at the object is shown. The impulses are repeated at the repetition rate *frep*. The template impulse in the correlator IC, shown in the lower part of Fig. 42(a), is operating at a repetition rate *frep* − Δ*f* . Therefore it is continuously changing its time alignment to the received signals and appears

> −80 −70 −60 −50 −40 −30 −20

Distance /mm

Fourier

**Figure 44.** Measurement of sinusoidally moving metal plate placed in front of the antennas at a distance

**Figure 45.** Time domain breathing measurements of (a) a male test person lying on the back and (b) a

At each position of the impulses the cross-correlation is computed by multiplying the signals and integrating them. This operation can be seen in Fig. 42(b) in the region where one received impulse and the template impulse are overlapping each other. The lower part of the figure illustrates the output of both filters. The solid line represents the output signal of the 800 MHz filter which integrates each single impulse. The dashed line shows the output signal of the baseband filter with a cut-off frequency of 25 kHz, which integrates a multitude of single

Transform /dB

0 1 2 3 4 5 6 7 8

UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications

475

Frequency/ Hz

0 1 2 3 4 5 6 7 8

Time/ s

(b)

(b)

sweeping through the impulse sequence.

0 1 2 3 4 5 6 7 8

Time/ s

0 1 2 3 4 5 6 7 8

Time /s

(a)

seven-week old child sleeping in bed.

of 19.6 cm and a deviation of approx. 1 mm in (a) time and (b) frequency domain.

(a)

194

Distance /mm 195

196

Distance /mm 197

198

**Figure 42.** (a) Illustration of the template impulse sweeping through the antenna receive signal. (b) Illustration of the evolving correlation signal from receive and template signal in the region where the impulses overlap.

and transferred to the PC as well. In the PC a post-processing of the continuous data stream is done using a custom software written in Lab Windows/CVI. Pictures of transmit and receive antenna with mounted components are shown in Fig. 41. A detailed description and supporting measurements of the hardware can be found in [7, 31].

**Figure 43.** Illustration of the correlation signal at the receiver antenna output port, together with the Δ*f*-signal for separation of the repetition.

To determine the distance between sensor and desired object and the variation of the distance between them, a time-of-flight measurement is applied using a sweeping impulse correlation as explained in [10]. The sweeping impulse correlation is very similar to an undersampling technique and avoids high-speed sampling of the gigahertz-range UWB impulses. This method is illustrated in Fig. 42 by showing the signals appearing in the receiver. In the upper trace of Fig. 42(a) the received signal at the output of the LNA in the correlator IC is shown. An impulse from both, the non-inverted direct coupling and the inverted reflection at the object is shown. The impulses are repeated at the repetition rate *frep*. The template impulse in the correlator IC, shown in the lower part of Fig. 42(a), is operating at a repetition rate *frep* − Δ*f* . Therefore it is continuously changing its time alignment to the received signals and appears sweeping through the impulse sequence.

36 Will-be-set-by-IN-TECH

*frep* − Δ*f* = 200 MHz - 25 Hz. This sinusoidal clock signal is filtered as well with a helix filter to improve jitter performance. The output signal of the correlator IC is processed with a baseband circuit, where it is amplified and further integrated by a low-pass filter with a cut-off frequency of 25 kHz. After this it is fed to a data-logger, which samples the generated signals with a sampling period of 30 μs and transfers them to a PC for further processing. Additionally a synchronizing signal of Δ*f* = 25 Hz from the third DDS clock signal is sampled

Antenna

(f rep)

(f rep−Δf)

**Figure 42.** (a) Illustration of the template impulse sweeping through the antenna receive signal. (b) Illustration of the evolving correlation signal from receive and template signal in the region where the

and transferred to the PC as well. In the PC a post-processing of the continuous data stream is done using a custom software written in Lab Windows/CVI. Pictures of transmit and receive antenna with mounted components are shown in Fig. 41. A detailed description and

> 0 10 20 30 40 50 60 70 Time /ms

Object

1/Δf

**Figure 43.** Illustration of the correlation signal at the receiver antenna output port, together with the

To determine the distance between sensor and desired object and the variation of the distance between them, a time-of-flight measurement is applied using a sweeping impulse correlation as explained in [10]. The sweeping impulse correlation is very similar to an undersampling technique and avoids high-speed sampling of the gigahertz-range UWB impulses. This

Coupling

Template

τ

Filter

τ<sup>=</sup> 0 ps 10ps 20 ps...

(b)

 0 1 2 3 4 5 6 7 8 9 10 11 12 Time /ns

(a)

1/frep

Object

1/(f rep−Δf)

supporting measurements of the hardware can be found in [7, 31].

Corr.

Δf−Signal

Δ*f*-signal for separation of the repetition.

Template

impulses overlap.

Coupling

**Figure 44.** Measurement of sinusoidally moving metal plate placed in front of the antennas at a distance of 19.6 cm and a deviation of approx. 1 mm in (a) time and (b) frequency domain.

**Figure 45.** Time domain breathing measurements of (a) a male test person lying on the back and (b) a seven-week old child sleeping in bed.

At each position of the impulses the cross-correlation is computed by multiplying the signals and integrating them. This operation can be seen in Fig. 42(b) in the region where one received impulse and the template impulse are overlapping each other. The lower part of the figure illustrates the output of both filters. The solid line represents the output signal of the 800 MHz filter which integrates each single impulse. The dashed line shows the output signal of the baseband filter with a cut-off frequency of 25 kHz, which integrates a multitude of single

#### 38 Will-be-set-by-IN-TECH 476 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications <sup>39</sup>

impulses. By this procedure, a cross-correlation curve of two fifth Gaussian derivatives is developing. The correlated curve of the above illustration example is depicted in Fig. 43. The correlation impulses from the direct coupling and the reflection at the object can be distinguished. As discussed, the reflection at the object is mirrored, because the impulse is inverted by the reflection. This signal is present at the output of the receive antenna and is sampled by the data-logger. The correlation signal is continuously repeated with a repetition rate of Δ*f* . For a separation of the correlation sweeps, the Δ*f*-signal is sampled as well by the data-logger. When the object under investigation is now moving, the part of the correlation signal coming from the object reflection is correspondingly changing its alignment to the Δ*f*-signal and the movement can be measured. The movement determination of the object is continuously done by software in the PC. First a separation of the correlation sweeps by the rising slope of the Δ*f*-signal is performed. Then both slopes of the correlation curves, the slope from the object and the slope from the direct coupling, are tracked and their positions continuously monitored. Tracking both slopes yielded best precision, compared to tracking the minimum of the correlation curve or only the slope of the correlation signal from the object [31].

(a) (b)

**Figure 46.** Photographs of transmit antenna with mounted differential impulse generator IC and the

generator and the correlation receiver front-end are mounted chip-on-board at the feeding points of the dipole antennas. Fig. 46 shows the pictures of transmit and receive antennas

Using the same DDS clock generator and post-processing software as described in Sec. 4.1, the ability of the realized differential bistatic radar system for tracking a metal plate which moves back and forth with a sinusoidal deviation is demonstrated. The measured result in the time domain can be seen in Fig. 47(a). A deviation amplitude of around 1.5 mm can be

−120

0 2 4 6 8 10 12

UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications

477

Frequency / Hz (b) Spectrum

−80

−40

Fourier Transform / dB

**Figure 47.** Measurement results of a moving metal plate in front of the bistatic radar with a distance of

clearly seen. Fig. 47(b) shows the calculated spectrum information of the measurement. The maximum point is clearly visible and indicates the movement frequency of the metal plate. A common application for this radar system is the detection of vital signs. Here, an adult male with pronounced tachypnea is seated 5 cm from the radar. Fig. 48(a) shows the recorded time domain data. The breathing pattern is clearly visible and its amplitude is around 5 mm. The time domain data is Fourier transformed to frequency domain as shown in Fig. 48(b). It

0

complete receiver with RF frontend IC and baseband circuit.

0 1 2 3 4

Time / s (a) Time domain

clearly indicates that the respiration rate is around 35/min.

with mounted differential ICs.

346

35 cm in time and frequency domains.

348

350

Distance / mm

352

354

To measure the precision of the sensor a metal plate is placed in front, which is mounted on a sledge driven by an eccentric disk, moving the metal plate forward and backward with an approximately sinusoidal deviation. Fig. 44(a) shows a time domain record of a movement measurement with the metal plate placed at a distance of 19.6 cm, a deviation amplitude of approximately 1 mm and a repetition rate of around 1.35 cycles/s. The movement is clearly resolved by the measurement. In Fig. 44(b) the calculated spectrum of the measurement can be seen. The frequency maximum is very clearly visible and verifies a precision of the demonstrator in the millimeter to sub-millimeter range.

In a further measurement the sensor is pointed to the abdomen of a male test person lying on the back at a distance of approximately 25 cm1. At the abdomen the largest breathing amplitude occurs. Fig. 45(a) shows a breathing measurement in case the person is breathing normally. The breathing amplitude exceeds 10 mm and the repetition rate is around 2.5 cycles/s. For a further measurement the demonstrator is placed towards a sleeping seven-week old child lying on the back at a distance of approximately 16.3 cm. Fig. 45(b) shows a rhythmic breathing period with a movement of around 1 mm in the direction of the sensor and a repetition rate of 1.5 cycles/s. These measurements show, that the sensor can be used to monitor the breathing of adults and infants lying on the back and that breathing patterns can clearly be detected using the single-ended bistatic impulse-radio UWB radar demonstrator.
