**2.2. Transmitter MMICs**

All integrated circuits reported in this section were realized in an inexpensive Si/SiGe HBT technology offered by Telefunken Semiconductors GmbH. Two kinds of transistors, with high fT (fT = 80 GHz, BVCEO = 2.4 V) and with high breakdown voltage (fT = 50 GHz, BVCEO = 4.3 V) are available simultaneously. The process incorporates 4 types of resistors, MIM capacitors, as well as 3 metalization layers. All the devices were fabricated on a low resistivity 20Ω cm

**Figure 5.** Reflection and radiation performance of the tissue optimized antenna.

6 Will-be-set-by-IN-TECH

structure, two substrate layers with slots in the top and bottom layer metalizations are used. The monopole is arranged in the center metalization layer and is fed by a triplate line. In this way, the buried feeding is insulated from the adjacent highly lossy human tissue. The antenna dimensions need to be optimized according to the surrounding medium of the antenna. The width of the antenna is 11 mm assuming skin tissue around the antenna with a typical permittivity of 28 at 6.85 GHz. The size reduction factor compared to an antenna designed for

air instead of skin tissue is 5.4 leading automatically to a miniaturized UWB antenna.

(a) Sketch (b) Photograph

**Figure 4.** Sketch of the tissue optimized antenna and photograph including microstrip-to-coaxial

results agree very well and show a return loss of more than 10 dB above 3.8 GHz.

In order to connect the antenna to a coaxial cable, a broadband transition from the triplate line to a coaxial line is applied [19]. The realized antenna including this transition is depicted in Fig. 4(b). The characterization of the tissue optimized antenna is performed in a tank filled with tissue-mimicking liquid approximating the permittivity and loss behavior of skin tissue. The chosen liquid is a 50% sugar solution in water [20]. Fig. 5(a) shows the return loss of the antenna being inside the sugar-water solution and compares its performance with a measurement, where the antenna is placed on both sides on human skin. Both measurement

The radiation pattern is also measured in the tissue-mimicking liquid using two identical antennas and applying the two-antenna method. The obtained radiation pattern for the H-plane is presented in Fig. 5(b). There, the losses of the tissue-mimicking medium are compensated. Since the losses are increasing significantly with increasing frequency, measurements only up to 9 GHz are possible limited by the dynamic range of the measurement setup. Within this frequency range, a desired uniform and broad characteristic is achieved. Hence, UWB performance for a sufficiently small antenna for implants is demonstrated. For catheter localization additional miniaturization is required. A possible approach as well as more details about all presented antennas in Sec. 2.1 can be found in [20].

All integrated circuits reported in this section were realized in an inexpensive Si/SiGe HBT technology offered by Telefunken Semiconductors GmbH. Two kinds of transistors, with high fT (fT = 80 GHz, BVCEO = 2.4 V) and with high breakdown voltage (fT = 50 GHz, BVCEO = 4.3 V) are available simultaneously. The process incorporates 4 types of resistors, MIM capacitors, as well as 3 metalization layers. All the devices were fabricated on a low resistivity 20Ω cm

transition.

**2.2. Transmitter MMICs**

substrate. The technology is fully adequate for impulse-radio-ultra-wideband (IR-UWB) applications.

Generating short time-domain impulses making efficient use of the spectral mask is the key challenge in IR-UWB systems. Approaches include the up-conversion of base band pulses to the allocated UWB frequency band using an oscillator and mixer [39] and direct generation based on damped relaxation oscillator [8]. Here impulse generators based on a quenched-oscillator concept with great circuit simplicity are presented. A cross-coupled LC oscillator is chosen as the core to introduce tunability of the waveform and the inherent convenience of achieving biphase modulation.

**Figure 6.** Complete circuit schematic of the UWB impulse generator. The dashed components (C1*b*, C2 and T7) show the extension for tunability of the impulse shape for different spectral masks and transistor level simulation: the collector potential of T3 and the collector current of T4.

Fig. 6(a) shows the impulse generator circuit. First, disregard the components with a dashed line, which are the extention for tunability of the impulse shape. T1 and T2 form a Schmitt trigger, creating a fast rising edge at the collector of T2 when a positive input signal triggers T1 to be on. This reduces the effect of the time-domain influence of the input clock signal

#### 8 Will-be-set-by-IN-TECH 446 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications <sup>9</sup>

on the output impulses. After an initial sharp increase, the collector potential of T2 is pulled down again to a lower constant value by the current mirror formed by T3 and T4. So the collector potential of T3 has a spike performance before it becomes stable, as shown in Fig. 6(b), which correspondingly generates a collector current spike at T4, creating the envelope of the output impulse. During the rest of the period, the collector current of T4 is too low to turn the oscillator on since T3 is chosen much larger than T4. The width of the current spike is determined predominately by the time constant *τ* of the charging circuit including the resistor R1 and the base-emitter capacitor C*be*<sup>3</sup> of T3. *τ* can be written as

$$
\pi = R\_1 \mathbb{C}\_{\mathbb{H}^3}.\tag{1}
$$

−200 −100 0 100 200

−200 −100 0 100 200

*τ*<sup>1</sup> can be written as

frequency can be expressed as

 0 2 4 6 8 10 12 Time / ns

 0 2 4 6 8 10 12 Time / ns

(a) Time domain

the FCC mask for indoor UWB applications.

f=100 MHz

f=1.3 GHz

indoor spectral mask. The impulse generator is shown in Fig. 7(a)

−100 −90 −80 −70 −60 −50 −40 −30 −20

PSD / dBm/MHz

**Figure 8.** Measured results of time-domain output impulse waveforms at 100 MHz and 1.3 GHz repetition rate and the spectrum of the 100 MHz impulse train, demonstrating compliance with the FCC

displayed using the subtraction feature of a real time oscilloscope. The measured results show a peak-peak amplitude of 200 mV, and a full width at half maximum (FWHM) of the envelope of 0.3 ns. This circuit has a very low power consumption: 6 mW at 100 MHz and 10 mW at 1.3 GHz. The spectrum of the measured 100 MHz impulse train can be seen in Fig. 8(b). The maximum power spectral density (PSD) is -41.3 dBm per spectral line, and it has a 10 dB bandwith of 4.9 GHz from 3.5-8.4 GHz. It shows that the output spectrum complies well with

The two parameters (*τ* and *ω*0) which determine the impulse parameters (envelope and osillation frequency) are easily modified. This is shown by the dashed components in Fig. 6(a). C2 and T7 are introduced to modify the envelope by changing the capacitance between the base of T3 and ground, switching *C*<sup>2</sup> in and out. When V1 = 0, T7 is off, *τ* is the same as before, resulting the emitted impulses to conform to the FCC mask. When V1 = 1 V, T7 is on, the charging circuit will include R1 and C*be*<sup>3</sup> in parallel with *C*2. In this case, the time constant

Since C2 is chosen much larger than C*be*3, the envelope width of the impulses is larger in this situation, suiting for ECC or Japanese masks depending on the center frequency adjustment. The tank circuit capacitance is now formed by C1*<sup>a</sup>* in series with a varactor C1*b*. The oscillation

Through changing the varactor capacitance C1*<sup>b</sup>* with V2, the center frequency *ω*<sup>0</sup> is adjustable. Depending on the applied voltages V1 and V2, the generated impulses conform to the FCC, ECC mask, or Japanese mask. The microphotogragh of this tunable impulse generator can be

By setting V1 = 0 and V2 = 2 V, the LC oscillator is triggered with a shorter current spike. So the generated waveform is similar as shown in Fig. 8(a), targetting the FCC mask. This impulse

*<sup>C</sup>*1*a*+*C*1*<sup>b</sup>* + *Cpara*)(*L*<sup>1</sup> + *L*2)

*<sup>ω</sup>*<sup>0</sup> <sup>=</sup> <sup>1</sup> 

seen in Fig. 7(b). It is quite compact with an area of 0.53 x 0.61 mm2.

( *<sup>C</sup>*1*aC*<sup>1</sup>*<sup>b</sup>*

0 2 4 6 8 10 12

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447

FCC indoor mask Measured

Frequency / GHz

(b) Spectrum

*τ*<sup>1</sup> = *R*1(*Cbe*<sup>3</sup> + *C*2). (3)

. (4)

Amplitude / mV

Amplitude / mV

The amplitude of the current spike can be easily adjusted by changing the size of T4. The repetition rate of the current spike train is equal to the input signal frequency and limited by the spike width.

The LC oscillator is activated by the current spikes once the collector currents of the cross-coupled pair (T5 and T6) are high enough to create a negative real part of the impedance. A slight asymmetry in the cross-coupled pair ensures that the oscillation always starts with the same phase and shortens the start-up time which in turn reduces power consumption because the necessary current spike width for a given output impulse envelope is shortened. R2 is placed to quench the oscillator off more quickly immediately after the current spikes have disappeared. Thus, short-time domain impulses with a repetition rate equal to the input signal frequency are generated. The center frequency *ω*<sup>0</sup> of the oscillation is mainly determined by L1, L2, C1*<sup>a</sup>* and the parasitic capacitance from the cross-coupled pair C*para*. *ω*<sup>0</sup> can be expressed as

$$
\omega\_0 = \frac{1}{\sqrt{(\mathbb{C}\_{1a} + \mathbb{C}\_{para})(L\_1 + L\_2)}}.\tag{2}
$$

It is designed to be around 6 GHz to fit the FCC spectral mask. The microphotograph of this realized impulse generator is shown in Fig. 7(a). It is a quite compact design with an area of 0.50 x 0.60 mm2 due to a simple circuit topology.

**Figure 7.** Microphotographs of (a) the realized UWB impulse generator shown in Fig. 6(a) excluding the components with a dashed outline and (b) the impulse generator tunable to FCC, ECC and Japanese spectral masks.

The impulses measured on-chip in time domain are shown in Fig. 8(a). The impulse generator is fed with a 100 MHz and 1.3 GHz sinusoidal signal separately. The differential signal is

8 Will-be-set-by-IN-TECH

on the output impulses. After an initial sharp increase, the collector potential of T2 is pulled down again to a lower constant value by the current mirror formed by T3 and T4. So the collector potential of T3 has a spike performance before it becomes stable, as shown in Fig. 6(b), which correspondingly generates a collector current spike at T4, creating the envelope of the output impulse. During the rest of the period, the collector current of T4 is too low to turn the oscillator on since T3 is chosen much larger than T4. The width of the current spike is determined predominately by the time constant *τ* of the charging circuit including the resistor

The amplitude of the current spike can be easily adjusted by changing the size of T4. The repetition rate of the current spike train is equal to the input signal frequency and limited by

The LC oscillator is activated by the current spikes once the collector currents of the cross-coupled pair (T5 and T6) are high enough to create a negative real part of the impedance. A slight asymmetry in the cross-coupled pair ensures that the oscillation always starts with the same phase and shortens the start-up time which in turn reduces power consumption because the necessary current spike width for a given output impulse envelope is shortened. R2 is placed to quench the oscillator off more quickly immediately after the current spikes have disappeared. Thus, short-time domain impulses with a repetition rate equal to the input signal frequency are generated. The center frequency *ω*<sup>0</sup> of the oscillation is mainly determined by L1, L2, C1*<sup>a</sup>* and the parasitic capacitance from the cross-coupled pair C*para*. *ω*<sup>0</sup>

(*C*1*<sup>a</sup>* + *Cpara*)(*L*<sup>1</sup> + *L*2)

It is designed to be around 6 GHz to fit the FCC spectral mask. The microphotograph of this realized impulse generator is shown in Fig. 7(a). It is a quite compact design with an area of

(a) (b)

**Figure 7.** Microphotographs of (a) the realized UWB impulse generator shown in Fig. 6(a) excluding the components with a dashed outline and (b) the impulse generator tunable to FCC, ECC and Japanese

The impulses measured on-chip in time domain are shown in Fig. 8(a). The impulse generator is fed with a 100 MHz and 1.3 GHz sinusoidal signal separately. The differential signal is

*<sup>ω</sup>*<sup>0</sup> <sup>=</sup> <sup>1</sup> 

*τ* = *R*1*Cbe*3. (1)

. (2)

R1 and the base-emitter capacitor C*be*<sup>3</sup> of T3. *τ* can be written as

the spike width.

can be expressed as

spectral masks.

0.50 x 0.60 mm2 due to a simple circuit topology.

**Figure 8.** Measured results of time-domain output impulse waveforms at 100 MHz and 1.3 GHz repetition rate and the spectrum of the 100 MHz impulse train, demonstrating compliance with the FCC indoor spectral mask. The impulse generator is shown in Fig. 7(a)

displayed using the subtraction feature of a real time oscilloscope. The measured results show a peak-peak amplitude of 200 mV, and a full width at half maximum (FWHM) of the envelope of 0.3 ns. This circuit has a very low power consumption: 6 mW at 100 MHz and 10 mW at 1.3 GHz. The spectrum of the measured 100 MHz impulse train can be seen in Fig. 8(b). The maximum power spectral density (PSD) is -41.3 dBm per spectral line, and it has a 10 dB bandwith of 4.9 GHz from 3.5-8.4 GHz. It shows that the output spectrum complies well with the FCC mask for indoor UWB applications.

The two parameters (*τ* and *ω*0) which determine the impulse parameters (envelope and osillation frequency) are easily modified. This is shown by the dashed components in Fig. 6(a). C2 and T7 are introduced to modify the envelope by changing the capacitance between the base of T3 and ground, switching *C*<sup>2</sup> in and out. When V1 = 0, T7 is off, *τ* is the same as before, resulting the emitted impulses to conform to the FCC mask. When V1 = 1 V, T7 is on, the charging circuit will include R1 and C*be*<sup>3</sup> in parallel with *C*2. In this case, the time constant *τ*<sup>1</sup> can be written as

$$
\pi\_1 = R\_1(\mathbb{C}\_{be3} + \mathbb{C}\_2). \tag{3}
$$

Since C2 is chosen much larger than C*be*3, the envelope width of the impulses is larger in this situation, suiting for ECC or Japanese masks depending on the center frequency adjustment. The tank circuit capacitance is now formed by C1*<sup>a</sup>* in series with a varactor C1*b*. The oscillation frequency can be expressed as

$$\omega\_0 = \frac{1}{\sqrt{(\frac{\mathbb{C}\_{1d}\mathbb{C}\_{1b}}{\mathbb{C}\_{1a} + \mathbb{C}\_{1b}} + \mathbb{C}\_{para})(L\_1 + L\_2)}}.\tag{4}$$

Through changing the varactor capacitance C1*<sup>b</sup>* with V2, the center frequency *ω*<sup>0</sup> is adjustable. Depending on the applied voltages V1 and V2, the generated impulses conform to the FCC, ECC mask, or Japanese mask. The microphotogragh of this tunable impulse generator can be seen in Fig. 7(b). It is quite compact with an area of 0.53 x 0.61 mm2.

By setting V1 = 0 and V2 = 2 V, the LC oscillator is triggered with a shorter current spike. So the generated waveform is similar as shown in Fig. 8(a), targetting the FCC mask. This impulse generator is suitable for the ECC mask when V1 = 1 V and V2 = 2.3 V. Under these conditions, the LC oscillator is triggered by a longer current spike with a FWHM of 2 ns. The measured output impulse train can be seen in Fig. 9(a). The impulses have a peak-peak amplitude of

The performance under these three modes is summarized in Tab. 1. This impulse generator can be used for on-off keying (OOK) and pulse-position modulation (PPM) in all these three

V1=0, V2=2.0 V

V1=1 V, V2=2.3 V

V1=1 V, V2=6 V **Table 1.** Performance summary of the tunable impulse generator.

impulse generator with biphase modulation function.

an area of 0.56 X 0.53 mm2.

Mode 10dB bandwidth V*PP* power cons. (setting) (GHz) (V) (mW) FCC 4.2 0.36 6

449

UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications

ECC 0.6 0.5 10

Japanese mask 0.6 0.5 10

Biphase modulation capability can be introduced by modifying the DC currents flowing in the individual branches of the differential LC oscillator. As shown in Fig. 11(a), additional branch currents are set through current mirrors T7, T9 and T8, T10. When the input data signal is

(a) Circuit schematic (b) Microphotograph

**Figure 11.** Adding biphase capability to the impulse generator in Fig. 6 and the microphotograph of the

low, transistor T9 is off, and the collector current I1 of T7 is zero because of the current mirror configuration to T9. Meanwhile, the applied voltage V3 will generate a collector current I2 in T8 through the current mirror configuration of T8 and T10, because T11 is off. When the data signal is high, T10 switches into saturation and T8 blocks, which causes I2 to be zero. At the same time, the high potential at the base of T9 introduces a collector current I1 in T7. Thus, oscillation will start in one of these two phase states once the tail transistor T4 is turned on, constituting the biphase modulation. Additionally, this asymmetry shortens the start-up time, which in turn reduces the power consumption. The fabricated IC is shown in Fig. 11(b). It has

The measured time domain waveforms with different voltage potentials applied to the data port can be seen in Fig. 12(a). The results show a peak-peak amplitude of 260 mV and an envelope width of 0.3 ns FWHM. The orientation of the impulses is clearly reversed, showing a perfect biphase modulation. Fig. 12(b) shows the spectrum of a 200 MHz impulse train with the data port connected to ground. It is centered around 6 GHz with a 10 dB bandwidth of

modes.

**Figure 9.** Measured results of impulse waveform in time domain and normalized PSD of the tunable impulse generator output signal setting for compliance with the ECC UWB mask.

0.5 V. The circuit has a total power consumption of 10 mW and a maximum output impulse repetition rate exceeding 300 MHz in this case. The normalized PSD of the impulse train has a center frequency around 7 GHz with a 10 dB bandwidth of 600 MHz, shown in Fig. 9(b). It fits well into the ECC mask. By changing the value of V2, the center frequency of the impulses will be shifted, this makes the circuit usable for the Japanese mask. The measured output signal

**Figure 10.** Measured results of time domain waveform and normalized PSD of the output signal targeting the Japanese UWB mask.

in the time domain with V*tune*<sup>2</sup> = 6 V is shown in Fig. 10(a). The measured impulse train has a similar envelope as the mode targeting the ECC mask because the triggered current spike has the same width. The peak-peak amplitude of the impulses whose envelope has a FWHM of 2 ns is 0.5 V. The complete power consumption in this mode is 10 mW. The spectrum is presented in Fig. 10(b). It shows that the center frequency is shifted to 8.7 GHz for a good fit to the Japanese mask.

The performance under these three modes is summarized in Tab. 1. This impulse generator can be used for on-off keying (OOK) and pulse-position modulation (PPM) in all these three modes.


**Table 1.** Performance summary of the tunable impulse generator.

10 Will-be-set-by-IN-TECH

generator is suitable for the ECC mask when V1 = 1 V and V2 = 2.3 V. Under these conditions, the LC oscillator is triggered by a longer current spike with a FWHM of 2 ns. The measured output impulse train can be seen in Fig. 9(a). The impulses have a peak-peak amplitude of

> −60 −50 −40 −30 −20 −10 0 10 20

> −60 −50 −40 −30 −20 −10 0 10 20

Normalized PSD / dB

**Figure 10.** Measured results of time domain waveform and normalized PSD of the output signal

in the time domain with V*tune*<sup>2</sup> = 6 V is shown in Fig. 10(a). The measured impulse train has a similar envelope as the mode targeting the ECC mask because the triggered current spike has the same width. The peak-peak amplitude of the impulses whose envelope has a FWHM of 2 ns is 0.5 V. The complete power consumption in this mode is 10 mW. The spectrum is presented in Fig. 10(b). It shows that the center frequency is shifted to 8.7 GHz for a good fit

ECC mask Measured

Normalised PSD / dB

**Figure 9.** Measured results of impulse waveform in time domain and normalized PSD of the tunable

0.5 V. The circuit has a total power consumption of 10 mW and a maximum output impulse repetition rate exceeding 300 MHz in this case. The normalized PSD of the impulse train has a center frequency around 7 GHz with a 10 dB bandwidth of 600 MHz, shown in Fig. 9(b). It fits well into the ECC mask. By changing the value of V2, the center frequency of the impulses will be shifted, this makes the circuit usable for the Japanese mask. The measured output signal

impulse generator output signal setting for compliance with the ECC UWB mask.

0 2 4 6 8 10 12

Frequency / GHz

0 2 4 6 8 10 12

Frequency / GHz

(b) Spectrum

(b) Spectrum

Japanese mask Measured

−0.3 −0.2 −0.1 0 0.1 0.2 0.3

−0.3 −0.2 −0.1 0 0.1 0.2 0.3

targeting the Japanese UWB mask.

to the Japanese mask.

Amplitude / V

Amplitude / V

0 2 4 6 8 10 12 14

Time / ns (a) Time domain

0 2 4 6 8 10 12 14

Time / ns (a) Time domain Biphase modulation capability can be introduced by modifying the DC currents flowing in the individual branches of the differential LC oscillator. As shown in Fig. 11(a), additional branch currents are set through current mirrors T7, T9 and T8, T10. When the input data signal is

**Figure 11.** Adding biphase capability to the impulse generator in Fig. 6 and the microphotograph of the impulse generator with biphase modulation function.

low, transistor T9 is off, and the collector current I1 of T7 is zero because of the current mirror configuration to T9. Meanwhile, the applied voltage V3 will generate a collector current I2 in T8 through the current mirror configuration of T8 and T10, because T11 is off. When the data signal is high, T10 switches into saturation and T8 blocks, which causes I2 to be zero. At the same time, the high potential at the base of T9 introduces a collector current I1 in T7. Thus, oscillation will start in one of these two phase states once the tail transistor T4 is turned on, constituting the biphase modulation. Additionally, this asymmetry shortens the start-up time, which in turn reduces the power consumption. The fabricated IC is shown in Fig. 11(b). It has an area of 0.56 X 0.53 mm2.

The measured time domain waveforms with different voltage potentials applied to the data port can be seen in Fig. 12(a). The results show a peak-peak amplitude of 260 mV and an envelope width of 0.3 ns FWHM. The orientation of the impulses is clearly reversed, showing a perfect biphase modulation. Fig. 12(b) shows the spectrum of a 200 MHz impulse train with the data port connected to ground. It is centered around 6 GHz with a 10 dB bandwidth of

12 Will-be-set-by-IN-TECH 450 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications UWB in Medicine – High Performance UWB Systems for Biomedical Diagnostics and Short Range Communications <sup>13</sup>

(a) Circuit schematic (b) Microphotograph

**Figure 13.** Complete circuit schematic and the chip microphotograph of the fully differential low-noise

the bandwidth and improves the input matching simultaneously. Careful selection of input transistor size and adjusting the bias point was done as a compromise between optimum current density for minimum noise figure, noise-matched input impedance and achievable bandwidth. The emitter size of T1 is chosen to be 0.5 *μ*m x 24.7 *μ*m and the emitter current is 5 mA. The wide band noise and input power match were accomplished by the selection of input transistor with suitable biasing and shunt-shunt resistive feedback. A negligible penalty, with a maximum value of 0.2 dB, is achieved within the entire band for not achieving noise match exactly. T5, T6 form a differential emitter follower buffer. The emitter degeneration

The microphotograph of this differential LNA is shown in Fig. 13(b). Because this design is completely inductor-less, the IC has an extremely small size of 0.37 X 0.38 mm<sup>2</sup> including all bound pads. The lowest available metal layer was placed below the large-sized bonding pads to provide a ground shield, as otherwise the noise figure may be deteriorated by the substrate

Noise figure / dB

**Figure 14.** Measurement results of the S-parameter magnitudes and single-ended and extracted

One drawback of the differential configuration is a complex measurement setup. Two identical passive microstrip line UWB baluns are used for differential S-parameter

3 4 5 6 7 8 9 10 11

Measured Fsingle Extracted Fdiff

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451

Frequency / GHz

(b) Noise figures

capacitors are used to improve the buffer bandwidth.

3 4 5 6 7 8 9 10 11

Measured |S11| Measured |S22| Measured |S21|

Frequency / GHz

(a) S-parameters

amplifier.

noise pick-up.

−60 −50 −40 −30 −20 −10 0 10 20

differential noise figures.

S−parameter magnitudes / dB

**Figure 12.** Measured time-domain results of biphase modulated impulses with different applied voltages at the data port and the spectrum information of a 200 MHz impulse train with the data port connected to ground.

6.7 GHz from 3.1 - 9.8 GHz, which complies well with the FCC spectral mask for indoor UWB systems.
