**3. Measurement system and dedicated hardware**

A measurement system has been built in order to verify the approach. Particular components, like the UWB-antennas used and the working principle of the whole set-up, are discussed in this section. A dedicated time domain transmission oscilloscope is presented. It is capable of transmitting and receiving UWB pulses with several gigahertz of bandwidth. It is specially tailored to the measurement problem and therefore functions with less hardware and software, is far more compact, and is cheaper compared to a universal laboratory instrument.

### **3.1. Compact ultra-wideband antennas and arrays**

The sensor system should be able to transmit and receive ultra-wideband (UWB) signals in two orthogonal polarizations. This is of great importance if the orientation of the object

4 Will-be-set-by-IN-TECH 326 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications ISOPerm: Non-Contacting Measurement of Dielectric Properties of Irregular Shaped Objects <sup>5</sup>

(a) Dual polarised dipole. (b) Dual polarised array.

antenna matching. The radiating elements have a maximum diameter of 30mm. The feed is provided through a coaxial-slotline transition, so that no other components are necessary. The distance between two dipoles having an equal polarization is 80mm in each direction; between dipoles with the same polarization it is <sup>√</sup><sup>2</sup> <sup>×</sup> 80mm. The dimensions are overall 160mm × 320mm × 100mm. The matching is better than 10dB between 1.6GHz and 4.2GHz.

ISOPerm: Non-Contacting Measurement of Dielectric Properties of Irregular Shaped Objects 327

The core of the measurement system is the signal generation and sampling. The requirements for the system are that the step signals generated have a rise time in the range of 100ps, and a large amplitude. These signals are transmitted through the measurement path after which, the step response has to be sampled. The system should have a high bandwidth, low noise and low jitter. Furthermore it should be as compact and affordable as possible. Therefore, an impulse technique using equivalent time sampling is the method of choice. Classical swept sine wave techniques, as used in network analysers or real time digital sampling oscilloscopes,

Equivalent time sampling is based on the repetitive stimulation by a measurement signal with a cycle duration of *T*0. At every repetition of the measurement signal the moment of sampling is shifted by Δ*T*. The cycle of the sampling clock is then *T*<sup>1</sup> = *T*<sup>0</sup> + Δ*T*. The

conversion and the data transport and storage is greatly reduced compared to a real time oscilloscope. This technique has been used for decades; with recently available MMICs, both high performance and cheap hardware can be achieved. It is the method of choice for the sampling of signals with huge bandwidth in the GHz range with a high resolution. Laboratory instruments employing this technique have been used for preliminary investigations. In order to demonstrate the system performance and accuracy under practical conditions, dedicated hardware was developed. Two specially tailored systems were investigated. They differed

<sup>Δ</sup>*<sup>T</sup> T*1. The effort regarding analog-to-digital

The crosstalk between the single elements is maximum −20dB at 10GHz.

**3.2. Signal generation and sampling**

are too expensive, complex, and bulky.1

track and hold amplifier

XTAL

HP

f2 f2

output generator

reference path

input

sampled signal is therefore stretched to *T*<sup>2</sup> = *<sup>T</sup>*<sup>0</sup>

ADC FIFO

f2

clock generation

step

f1

memory

microcontroller

computer

**Figure 3.** Time domain transmission measurement system with a microcontroller.

measurements carried out with this method which lead to excellent results [20].

(a) Block diagram. (b) Prototype: integrated on one PCB except microcontroller and computer.

<sup>1</sup> A rather exotic concept employing M-sequences can be found in this book (see chapter HALOS). There are also

under test (e.g. prolate ellipsoid) is not known. There are several polarization set-ups for receiving co- and cross-polarized parts of the scattered wave. In the set-up used latterly the polarization of the transmitting antenna is tilted at 45◦ with respect to the receiving array so that ideally, without an object under test, the receiving antennas receive the transmitted pulse in equal proportions for each polarization. Therefore, dual orthogonal UWB antennas have been developed. Furthermore the single antennas should be as compact as possible while operating in the lower gigahertz range. The arrays presented later are not group antennas as commonly used. The received signal of every antenna is sampled separately in order not to confound the information at different locations with respect to the object under test. Operating at frequencies in the lower GHz range is anticipated because the electromagnetic waves have a higher penetration depth (especially for lossy dielectrics with a high permittivity) and therefore will be more affected.

There are many UWB antennas suitable for building dual polarized antennas [21–25]. Dual polarized dipole (see [26–31]) and horn antennas (see [32–35]) can already be found in the literature. Most of these antennas have the disadvantage that they have crossed feed points. It is possible to overcome this problem with the configuration depicted in Figure 2. Both the four radiation elements and the whole configuration are two-fold symmetric. Two dipoles are excited in even mode in horizontal as well as in vertical polarization. Ideally, due to the symmetry, both planes are decoupled. The diameter of the radiation elements is 24mm; the whole PCB containing the four radiating elements measures 50mm × 50mm. The antenna is equipped with two orthogonal feed networks; each is composed of a two stage Wilkinson divider and two tapered baluns. The horizontal and vertical polarization feed networks have dimensions of 50mm × 142mm and 50mm × 103mm, respectively. In order to avoid backward radiation, absorber material is mounted appropriately (the absorber material was removed for the photograph shown in Figure 2).

The matching at both ports is better than 7dB in a frequency range from 2GHz to 5.7GHz; the isolation is better than 30dB. At 5GHz the radiated crosspolar portion is suppressed by more than 25dB (boresight); the gain is 5.2dB. Measurements with a line containing four of these antennas lead to excellent results but the whole antenna structure is rather complex. Since a scenario with an object moving orthogonally to the line array is anticipated (e.g. a conveyor belt), a simpler arrangement can be used. The antenna array shown in Figure 2 consists of eight single linear broadband dipoles. The geometry is optimized regarding the antenna matching. The radiating elements have a maximum diameter of 30mm. The feed is provided through a coaxial-slotline transition, so that no other components are necessary. The distance between two dipoles having an equal polarization is 80mm in each direction; between dipoles with the same polarization it is <sup>√</sup><sup>2</sup> <sup>×</sup> 80mm. The dimensions are overall 160mm × 320mm × 100mm. The matching is better than 10dB between 1.6GHz and 4.2GHz. The crosstalk between the single elements is maximum −20dB at 10GHz.

### **3.2. Signal generation and sampling**

4 Will-be-set-by-IN-TECH

(a) Dual polarised dipole. (b) Dual polarised array.

under test (e.g. prolate ellipsoid) is not known. There are several polarization set-ups for receiving co- and cross-polarized parts of the scattered wave. In the set-up used latterly the polarization of the transmitting antenna is tilted at 45◦ with respect to the receiving array so that ideally, without an object under test, the receiving antennas receive the transmitted pulse in equal proportions for each polarization. Therefore, dual orthogonal UWB antennas have been developed. Furthermore the single antennas should be as compact as possible while operating in the lower gigahertz range. The arrays presented later are not group antennas as commonly used. The received signal of every antenna is sampled separately in order not to confound the information at different locations with respect to the object under test. Operating at frequencies in the lower GHz range is anticipated because the electromagnetic waves have a higher penetration depth (especially for lossy dielectrics with a high permittivity) and

There are many UWB antennas suitable for building dual polarized antennas [21–25]. Dual polarized dipole (see [26–31]) and horn antennas (see [32–35]) can already be found in the literature. Most of these antennas have the disadvantage that they have crossed feed points. It is possible to overcome this problem with the configuration depicted in Figure 2. Both the four radiation elements and the whole configuration are two-fold symmetric. Two dipoles are excited in even mode in horizontal as well as in vertical polarization. Ideally, due to the symmetry, both planes are decoupled. The diameter of the radiation elements is 24mm; the whole PCB containing the four radiating elements measures 50mm × 50mm. The antenna is equipped with two orthogonal feed networks; each is composed of a two stage Wilkinson divider and two tapered baluns. The horizontal and vertical polarization feed networks have dimensions of 50mm × 142mm and 50mm × 103mm, respectively. In order to avoid backward radiation, absorber material is mounted appropriately (the absorber material was removed for

The matching at both ports is better than 7dB in a frequency range from 2GHz to 5.7GHz; the isolation is better than 30dB. At 5GHz the radiated crosspolar portion is suppressed by more than 25dB (boresight); the gain is 5.2dB. Measurements with a line containing four of these antennas lead to excellent results but the whole antenna structure is rather complex. Since a scenario with an object moving orthogonally to the line array is anticipated (e.g. a conveyor belt), a simpler arrangement can be used. The antenna array shown in Figure 2 consists of eight single linear broadband dipoles. The geometry is optimized regarding the

**Figure 2.** Ultra-wideband antenna structures.

therefore will be more affected.

the photograph shown in Figure 2).

The core of the measurement system is the signal generation and sampling. The requirements for the system are that the step signals generated have a rise time in the range of 100ps, and a large amplitude. These signals are transmitted through the measurement path after which, the step response has to be sampled. The system should have a high bandwidth, low noise and low jitter. Furthermore it should be as compact and affordable as possible. Therefore, an impulse technique using equivalent time sampling is the method of choice. Classical swept sine wave techniques, as used in network analysers or real time digital sampling oscilloscopes, are too expensive, complex, and bulky.1

Equivalent time sampling is based on the repetitive stimulation by a measurement signal with a cycle duration of *T*0. At every repetition of the measurement signal the moment of sampling is shifted by Δ*T*. The cycle of the sampling clock is then *T*<sup>1</sup> = *T*<sup>0</sup> + Δ*T*. The sampled signal is therefore stretched to *T*<sup>2</sup> = *<sup>T</sup>*<sup>0</sup> <sup>Δ</sup>*<sup>T</sup> T*1. The effort regarding analog-to-digital conversion and the data transport and storage is greatly reduced compared to a real time oscilloscope. This technique has been used for decades; with recently available MMICs, both high performance and cheap hardware can be achieved. It is the method of choice for the sampling of signals with huge bandwidth in the GHz range with a high resolution. Laboratory instruments employing this technique have been used for preliminary investigations. In order to demonstrate the system performance and accuracy under practical conditions, dedicated hardware was developed. Two specially tailored systems were investigated. They differed

and computer.

**Figure 3.** Time domain transmission measurement system with a microcontroller.

<sup>1</sup> A rather exotic concept employing M-sequences can be found in this book (see chapter HALOS). There are also measurements carried out with this method which lead to excellent results [20].

0 100 200 300 400 500

TDS8000 TDT

0 100 200 300 400 500

TDS8000 TDT

t [ps]

(b) Monocycle.

−1

−0.5

0

normalized amplitude

**Figure 5.** Comparison between the proposed dedicated hardware (TDT) and a Textronix TDS8000 (no

TDT

(a) Block diagram. (b) Photograph.

of Δ*T* = 4ps. For the measurements presented later a much lower resolution is sufficient, e.g.

The entire measurement system is shown is Figure 6. The step signal of the TDT is amplified by a power amplifier (PA) before being transmitted through a horn antenna. The object under test is illuminated by the transmitted signal and portions of the scattered signal are received by an array of eight receiving antennas, which are arranged in two orthogonal polarizations. In order to emulate a 2-dimensional array, the movable table is moved orthogonally to the orientation of the receiving array. It is crucial to the multivariate calibration, which is applied

LNA

PA

combiner or switch

0.5

1

ISOPerm: Non-Contacting Measurement of Dielectric Properties of Irregular Shaped Objects 329

t [ps]

(a) Pulse.

movable table

0 0.2 0.4 0.6 0.8 1

normalized amplitude

averaging).

object under test

Tx

Rx

Δ*T* = 40ps.

**Figure 6.** Entire measurement set-up.

**3.3. Entire measurement setup**

**Figure 4.** Time domain transmission measurement system with a FPGA.

mostly in the digital part behind the analog-digital converter (ADC); the front end was very similar.

A block diagram and a photograph of the first system using a microcontroller are shown in Figure 3. Two slightly detuned clocks are synthesized from a crystal. One clock triggers a step generator, which transmits a step signal with about 30ps rise time (20% to 80%) and up to 3V amplitude. The other triggers the track and hold amplifier (having an input bandwidth of 13GHz), the 12bit ADC, and the asynchronous FIFO (first in first out) memory with a width of 18bit and a depth of 32k. A microcontroller and a computer control the system. The FIFO memory can be read out, and the clock generation can be programmed via the I2C-bus. Clock signals up to hundreds of megahertz can be chosen with an accuracy of 1ppm. The dimensions of the RF-PCB are 90mm × 60mm. A reference path is necessary to measure time differences, because the phase of the two clock signals is not captured. A FIFO has to be used because the microcontroller is not able to read in 12bit words at tens of megahertz. Therefore, one waveform of up to 32k is captured at a time and is then transferred to the computer. Averaging, in order to improve the SNR, is carried out on the computer.

An improvement is the application of a field programmable gate array (FPGA). A modular system is shown in Figure 4. There is no more need for a fast external FIFO because data can be read into the FPGA directly up to about 66MHz. Furthermore, no reference path is needed, because the FPGA is able to discriminate the phase between both clock signals. Up to 212 = 4096 sampling points are possible while an averaging of up to 213 = 8192 waveforms can be carried out on the FPGA (this is restricted due to the internal resources of the FPGA used2).

Since both front ends are similar they offer a comparable performance. The RMS noise level of the receiver is < 0.9mV, the RMS jitter is < 0.7ps (no averaging). Figure 5 shows a comparison of sampled test signals between the proposed hardware and a Tektronix TDS8000. The frequencies are chosen as *f*<sup>1</sup> = 50.01MHz and *f*<sup>2</sup> = 50MHz, which leads to a resolution

<sup>2</sup> A Spartan 6 from Xilinx is used.

**Figure 5.** Comparison between the proposed dedicated hardware (TDT) and a Textronix TDS8000 (no averaging).

6 Will-be-set-by-IN-TECH

FPGA

computer

mostly in the digital part behind the analog-digital converter (ADC); the front end was very

A block diagram and a photograph of the first system using a microcontroller are shown in Figure 3. Two slightly detuned clocks are synthesized from a crystal. One clock triggers a step generator, which transmits a step signal with about 30ps rise time (20% to 80%) and up to 3V amplitude. The other triggers the track and hold amplifier (having an input bandwidth of 13GHz), the 12bit ADC, and the asynchronous FIFO (first in first out) memory with a width of 18bit and a depth of 32k. A microcontroller and a computer control the system. The FIFO memory can be read out, and the clock generation can be programmed via the I2C-bus. Clock signals up to hundreds of megahertz can be chosen with an accuracy of 1ppm. The dimensions of the RF-PCB are 90mm × 60mm. A reference path is necessary to measure time differences, because the phase of the two clock signals is not captured. A FIFO has to be used because the microcontroller is not able to read in 12bit words at tens of megahertz. Therefore, one waveform of up to 32k is captured at a time and is then transferred to the computer.

An improvement is the application of a field programmable gate array (FPGA). A modular system is shown in Figure 4. There is no more need for a fast external FIFO because data can be read into the FPGA directly up to about 66MHz. Furthermore, no reference path is needed, because the FPGA is able to discriminate the phase between both clock signals. Up to 212 = 4096 sampling points are possible while an averaging of up to 213 = 8192 waveforms can be carried out on the FPGA (this is restricted due to the internal resources of the FPGA

Since both front ends are similar they offer a comparable performance. The RMS noise level of the receiver is < 0.9mV, the RMS jitter is < 0.7ps (no averaging). Figure 5 shows a comparison of sampled test signals between the proposed hardware and a Tektronix TDS8000. The frequencies are chosen as *f*<sup>1</sup> = 50.01MHz and *f*<sup>2</sup> = 50MHz, which leads to a resolution

(a) Block diagram. (b) Prototype: modular system except computer.

track and hold amplifier

XTAL

f2 f2

output generator

ADC

clock generation f1 f2

step

**Figure 4.** Time domain transmission measurement system with a FPGA.

Averaging, in order to improve the SNR, is carried out on the computer.

f1

input

similar.

used2).

<sup>2</sup> A Spartan 6 from Xilinx is used.

of Δ*T* = 4ps. For the measurements presented later a much lower resolution is sufficient, e.g. Δ*T* = 40ps.

### **3.3. Entire measurement setup**

The entire measurement system is shown is Figure 6. The step signal of the TDT is amplified by a power amplifier (PA) before being transmitted through a horn antenna. The object under test is illuminated by the transmitted signal and portions of the scattered signal are received by an array of eight receiving antennas, which are arranged in two orthogonal polarizations. In order to emulate a 2-dimensional array, the movable table is moved orthogonally to the orientation of the receiving array. It is crucial to the multivariate calibration, which is applied

#### 8 Will-be-set-by-IN-TECH 330 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications ISOPerm: Non-Contacting Measurement of Dielectric Properties of Irregular Shaped Objects <sup>9</sup>

later, that the received signals of each antenna placed at its unique position are sampled separately. Therefore, the antennas are switched sequentially onto the input of the TDT and are sampled there, after being amplified by a low noise amplifier (LNA). The MOSFET switch used requires relatively high hard- and software effort and it has to be synchronised with the other components included in the system. Furthermore, these switches have a latency of some nanoseconds. The switch has an insertion loss of 4dB at 2GHz and 8.6dB at 8GHz.

A simpler solution is to use a broadband combiner. The signals received by the individual antennas are combined using a broadband eight-way Wilkinson divider. Prior to combining there is a delay of *τ* = 2ns between two adjacent inputs. It is possible to separate the individual pulses in time. Compared to using a receiver having more channels, or using a switch, the hardware effort is greatly reduced and the instantaneous sampling of all pulses is possible in about 20ns. One of the disadvantages that has to be considered is that there is an increased insertion loss of ideally 9dB; at 1GHz and 5GHz the measured insertion losses are 9.3dB and 10.7dB, respectively. There are also losses and temperature dependencies due to the delay lines, which have to be taken into account.

(a) One of the moulds manufactured from

40 50 60

4.88 % moisture 20.93 % moisture

**Figure 8.** Examples of received pulses from two objects having different moisture content.

multivariate calibration methods which will be explained in the next section 5.

of liquid, and another ten with about 140g. Therefore, overall 20 objects were measured. A photograph of one of the measured objects containing 113.15g ethanol and 27.94g of water is

The time domain signals are gated in order to extract the time interval of interest: about 10 to 30 equidistant amplitude values per pulse are empirically chosen and subjected to the

The sequence of the eight sampled pulses consists of 217 sampling points. There are four positions of the movable table which then leads to 4 × 217 = 868 data points per object. These pulses are modified by many factors: the shape, the position, the rotation, and the intrinsic variables of the material under test. However, as can be seen in Figure 8 the value to be measured has only a relatively low influence on the apparent shape of the pulses, and

−0.15 −0.1 −0.05 0 0.05 0.1 0.15

amplitude [V]

t [ns]

(a) Instantaneous sampled pulses received by the eight antennas using a broadband Wilkinson divider and

**5. Multivariate data processing**

(b) Mould filled with moist clay granules. (c) A bottle filled

ISOPerm: Non-Contacting Measurement of Dielectric Properties of Irregular Shaped Objects 331

with an ethanol-water

mixture.

40.5 41 41.5

t [ns]

(b) Closer look at the first received pulse.

Styrofoam.

−0.2

delay lines.

shown in Figure 7.

−0.1

amplitude [V]

0

0.1

0.2

**Figure 7.** Objects under test.
