*2.3.2. Dielectrically scaled antennas*

For the process of detecting early stage breast cancer by means of microwave imaging (Section 5), the antenna size, the effective radiation of electromagnetic energy into the body, and the operational bandwidth turn out to be the main constraints regarding the design of the antenna.

The miniaturization of the antenna is of main concern to meet the requirements of the devised imaging technique (Section 5) of placing an array of many antennas surrounding the target under investigation (i.e. the human breast), considering also the small anatomic dimensions on the scale of the wavelengths of operation. In general, electrically small antennas are mismatched or narrowband [10], [11]. One possibility to overcome these obstacles is to use the antenna in *contact mode*, i.e., placing the antennas in contact with the target under investigation (e.g., the human body). With this *modus operandi* the antenna will radiate into a dielectric material (the human body), and it can be geometrically scaled by a factor of about √ε, where ε represents the dielectric permittivity of the target, without changing its electrical dimensions and, therefore, its radiative properties [12].

The *contact mode* presents advantages also with respect to the constraint of the effective radiation of electromagnetic energy into the body. In fact, it will not suffer from reflections occurring at the air-skin interface, due to the dielectric mismatch between the two grossly different media. This will also simplify the imaging processing since it prevents the need of surface reconstruction [10], [12]. Though, for practical and hygienic reasons, it is less convenient to put the array of antennas in direct contact with the patient's skin. However, the addition of a further layer, e.g. a disposable thin dielectric film, could spoil the effective radiation into the body1. Electromagnetic simulations (confirmed by measurements, Section 5.4) have shown that even the addition of a thin layer (~ 0.5 mm) can reduce the radiated power to less than half compared to the direct contact case. The implementation of a matching layer (with similar permittivity to human body tissues) can cure this effect (Fig. 5).

Eventually, particular attention has to be paid to the operational bandwidth of the antenna, especially to the lower cut-off frequency, which limits the penetration depth into the target. Based on a specific 14-layer model mimicking a trans-thoratic slice from the visual human data set, we have computed the penetration of electromagnetic waves into a human body, as shown in Fig. 6 [13], [14] and [15]. A strong increase of the signal attenuation with increasing frequency is clearly seen. Therefore, the lower cut-off frequency has to be set between 1 GHz and 2 GHz.

<sup>1</sup> The relevance of this phenomenon depends also on the antenna type used

264 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

separately.

the antenna.

and 2 GHz.

*2.3.2. Dielectrically scaled antennas* 

system (Section 4). These two goals provide different challenges in terms of antenna design, implementation, and experimental evaluation, both with respect to mechanical and electrical constraints [10]. As both applications involve different approaches, they will be treated

For the process of detecting early stage breast cancer by means of microwave imaging (Section 5), the antenna size, the effective radiation of electromagnetic energy into the body, and the operational bandwidth turn out to be the main constraints regarding the design of

The miniaturization of the antenna is of main concern to meet the requirements of the devised imaging technique (Section 5) of placing an array of many antennas surrounding the target under investigation (i.e. the human breast), considering also the small anatomic dimensions on the scale of the wavelengths of operation. In general, electrically small antennas are mismatched or narrowband [10], [11]. One possibility to overcome these obstacles is to use the antenna in *contact mode*, i.e., placing the antennas in contact with the target under investigation (e.g., the human body). With this *modus operandi* the antenna will radiate into a dielectric material (the human body), and it can be geometrically scaled by a factor of about √ε, where ε represents the dielectric permittivity of the target, without

The *contact mode* presents advantages also with respect to the constraint of the effective radiation of electromagnetic energy into the body. In fact, it will not suffer from reflections occurring at the air-skin interface, due to the dielectric mismatch between the two grossly different media. This will also simplify the imaging processing since it prevents the need of surface reconstruction [10], [12]. Though, for practical and hygienic reasons, it is less convenient to put the array of antennas in direct contact with the patient's skin. However, the addition of a further layer, e.g. a disposable thin dielectric film, could spoil the effective radiation into the body1. Electromagnetic simulations (confirmed by measurements, Section 5.4) have shown that even the addition of a thin layer (~ 0.5 mm) can reduce the radiated power to less than half compared to the direct contact case. The implementation of a matching layer (with similar permittivity to human body tissues) can cure this effect (Fig. 5).

Eventually, particular attention has to be paid to the operational bandwidth of the antenna, especially to the lower cut-off frequency, which limits the penetration depth into the target. Based on a specific 14-layer model mimicking a trans-thoratic slice from the visual human data set, we have computed the penetration of electromagnetic waves into a human body, as shown in Fig. 6 [13], [14] and [15]. A strong increase of the signal attenuation with increasing frequency is clearly seen. Therefore, the lower cut-off frequency has to be set between 1 GHz

1 The relevance of this phenomenon depends also on the antenna type used

changing its electrical dimensions and, therefore, its radiative properties [12].

**Figure 5.** Simulated scenarios to investigate the effective radiation of electromagnetic energy into the body (on the left). The antenna used is a bow-tie excited by a Gaussian pulse of a duration of around 80 ps FWHM. The *Phantom* material is a homogenous dispersive material simulating the dielectric behavior of the human body tissues. The results (on the right) represent the time-dependent co-polar component of the electric field evaluated at a distance of 44.5 mm from the phantom interface (the green spot in the figure). The examined cases are, from top to bottom: the antenna in direct contact with the phantom; with the implementation of a thin dielectric layer; with the implementation of a matching layer plus a thin dielectric layer, respectively.

ultraMEDIS – Ultra-Wideband Sensing in Medicine 267

**Figure 7.** The left-hand panel shows the completely processed ceramic body of the antenna. The center panel depicts the final antenna including metallization, feed line and dimensions. The right-hand panel

In order to properly manufacture the antenna and obtain good yield (of around 75 %) and reproducibility, specific manufacturing steps have to be devised, as indicated in the right image of Fig. 8 [22]. First, the white-colored ceramic powder (Fig. 8a) is pressed into the specific pyramidal shape; a cubic base accommodates the asymmetric feed (Fig. 8b). The dimensions of the raw body are slightly enlarged in order to respect the shrinkage upon sintering. The sintered body is complemented by grooves, which form the ridges (Fig. 8c), and is subsequently galvanically metalized with copper or gold (Fig. 8d). Due to the high permittivity of the ceramic body and a feed impedance of 50 Ω, the ridges are nearly linear in geometry, in contrast to the markedly curved shapes found in antennas for operation in air [12][15]. The feed is provided by a coaxial cable whose center conductor is fed through a small bore to the narrow end (diameter about 1.2 mm) of the ridged waveguide. A plastic housing and epoxy fixture provide a compact and mechanically rigid construction, to protect the ceramic body and the coaxial feed against torque and damages due to improper handling. It also provides a mechanical fixture to mount the antennas in an array of complex

geometry (Fig. 8e). Further details of the manufacturing processes are given in [23].

(c) (d)

(b)

**Figure 8.** The left-top picture shows a dry pressed green body of the antenna with some lubricant on it inside the dismantled mold. The left-bottom picture depicts the milling process to work in the ridges into the sintered ceramic body. The right picture illustrates the manufacturing steps for the sintered horn antenna: powder raw material (a), pressed raw (b), sintered (c), metallized (d), and fully packaged

(a)

(e)

1 cm

displays the profile of the ridges.

versions (e).

**Figure 6.** Penetration depth into a 14-layer model computed for different frequencies (see legend) [13].

The dielectric scaling of traveling-wave antennas, like tapered slot antennas and horn antennas, has to consider all three factors - the phase constant, the wave impedance, and the feeding structure [16]. Accordingly, extended iterative full-wave simulations were performed. The key parameters of the wideband radiation properties of double-ridged horn (DRH) antennas turned out to be the curvature of the ridges, the spacing between them, and the geometry of the pyramidal casing itself. Traveling-wave antennas have the benefit that the dielectric medium used to scale the antenna will serve as an embedded *matching medium*.

Our initial design was based on the immersion of dielectric liquids into the sinkhole of a dielectrically scaled DRH antenna [15]. The antenna could successfully be manufactured, using acetone as dielectric medium, with a scaling factor of about 4, but it was still insufficient to obtain a sufficiently compact antenna. The straight-forward approach to solve this problem was to replace the acetone by alternative high-permittivity dielectric materials, like a solid sintered ceramic. The ceramic powder is the commercial product LF-085 manufactured by MRA Laboratories based on neodymium titanate [17].

As the complex permittivity ε of the ceramic body of the antenna plays an important role not only in terms of antenna design but also in terms of *matching medium*, we performed measurements to access the complex permittivity following two different strategies: employing a split-post dielectric resonator (SPDR, [18]) and a dielectric resonator (DR) technique [19][20]. Both techniques are resonant methods and, hence, limit the experimental studies to a small set of discrete frequencies, because specific sample geometries are required for each measured frequency. The results showed that the sintered ceramic presents low frequency dispersion with a mean value of the real part of the permittivity ε' 72 [21], offering the potential for a scale factor of around 8.

The exploitation of the full potential of dielectric scaling leads to an aperture size of only 11 mm × 16 mm, but also to the reduction of the input impedance by the same scaling factor as by which the geometrical dimensions are scaled, resulting in a low value below 10 . This value implies a large mismatch in terms of standard electronic equipment, which is usually designed for a characteristic impedance of 50 Ω. In order to maintain the compatibility with standard electronic equipment, the antenna retains an aperture size of 24 mm × 24 mm, and a frequency bandwidth ranging from 1.5 to 5.5 GHz (Fig. 7).

266 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

manufactured by MRA Laboratories based on neodymium titanate [17].

ε' 72 [21], offering the potential for a scale factor of around 8.

a frequency bandwidth ranging from 1.5 to 5.5 GHz (Fig. 7).

**Figure 6.** Penetration depth into a 14-layer model computed for different frequencies (see legend) [13].

The dielectric scaling of traveling-wave antennas, like tapered slot antennas and horn antennas, has to consider all three factors - the phase constant, the wave impedance, and the feeding structure [16]. Accordingly, extended iterative full-wave simulations were performed. The key parameters of the wideband radiation properties of double-ridged horn (DRH) antennas turned out to be the curvature of the ridges, the spacing between them, and the geometry of the pyramidal casing itself. Traveling-wave antennas have the benefit that the dielectric medium used to scale the antenna will serve as an embedded *matching medium*. Our initial design was based on the immersion of dielectric liquids into the sinkhole of a dielectrically scaled DRH antenna [15]. The antenna could successfully be manufactured, using acetone as dielectric medium, with a scaling factor of about 4, but it was still insufficient to obtain a sufficiently compact antenna. The straight-forward approach to solve this problem was to replace the acetone by alternative high-permittivity dielectric materials, like a solid sintered ceramic. The ceramic powder is the commercial product LF-085

As the complex permittivity ε of the ceramic body of the antenna plays an important role not only in terms of antenna design but also in terms of *matching medium*, we performed measurements to access the complex permittivity following two different strategies: employing a split-post dielectric resonator (SPDR, [18]) and a dielectric resonator (DR) technique [19][20]. Both techniques are resonant methods and, hence, limit the experimental studies to a small set of discrete frequencies, because specific sample geometries are required for each measured frequency. The results showed that the sintered ceramic presents low frequency dispersion with a mean value of the real part of the permittivity

The exploitation of the full potential of dielectric scaling leads to an aperture size of only 11 mm × 16 mm, but also to the reduction of the input impedance by the same scaling factor as by which the geometrical dimensions are scaled, resulting in a low value below 10 . This value implies a large mismatch in terms of standard electronic equipment, which is usually designed for a characteristic impedance of 50 Ω. In order to maintain the compatibility with standard electronic equipment, the antenna retains an aperture size of 24 mm × 24 mm, and

**Figure 7.** The left-hand panel shows the completely processed ceramic body of the antenna. The center panel depicts the final antenna including metallization, feed line and dimensions. The right-hand panel displays the profile of the ridges.

In order to properly manufacture the antenna and obtain good yield (of around 75 %) and reproducibility, specific manufacturing steps have to be devised, as indicated in the right image of Fig. 8 [22]. First, the white-colored ceramic powder (Fig. 8a) is pressed into the specific pyramidal shape; a cubic base accommodates the asymmetric feed (Fig. 8b). The dimensions of the raw body are slightly enlarged in order to respect the shrinkage upon sintering. The sintered body is complemented by grooves, which form the ridges (Fig. 8c), and is subsequently galvanically metalized with copper or gold (Fig. 8d). Due to the high permittivity of the ceramic body and a feed impedance of 50 Ω, the ridges are nearly linear in geometry, in contrast to the markedly curved shapes found in antennas for operation in air [12][15]. The feed is provided by a coaxial cable whose center conductor is fed through a small bore to the narrow end (diameter about 1.2 mm) of the ridged waveguide. A plastic housing and epoxy fixture provide a compact and mechanically rigid construction, to protect the ceramic body and the coaxial feed against torque and damages due to improper handling. It also provides a mechanical fixture to mount the antennas in an array of complex geometry (Fig. 8e). Further details of the manufacturing processes are given in [23].

**Figure 8.** The left-top picture shows a dry pressed green body of the antenna with some lubricant on it inside the dismantled mold. The left-bottom picture depicts the milling process to work in the ridges into the sintered ceramic body. The right picture illustrates the manufacturing steps for the sintered horn antenna: powder raw material (a), pressed raw (b), sintered (c), metallized (d), and fully packaged versions (e).

A further issue of dielectrically scaled antennas is related to their measurement and characterization. As common measurement techniques and equipment cannot be applied, we followed three different strategies: measurements made in the frequency domain, measurements made in the time domain, and basic tests with volunteers.

ultraMEDIS – Ultra-Wideband Sensing in Medicine 269

**Figure 10.** Time domain reflections caused by the aperture plane due to dielectric mismatch between

The measurement of the radiating behavior is more complicated. To evaluate standard antenna parameters (e.g. gain, radiation pattern, etc.), the antennas should be placed in the Fraunhofer region. However, due to the high dielectric losses of water, the antenna could be placed at a maximum distance of 10 cm, which is not sufficient to meet the Fraunhofer region (starting from around 35 cm), but still is large enough to let the antennas operate in the Fresnel region. Near-field measurements are of main concern since the antennas are designed to be used in contact mode for biomedical imaging applications. The results show that the antennas present a flat frequency response (measured along the boresight direction), after the compensation of the frequency dispersion of the water (left diagram in Fig. 11), and 3 dB beam widths of nearly 20° for the E-Plane, and nearly 28° for the H-Plane

**Figure 11.** Transmission behavior between two identical antennas operating in de-ionized water at a distance of 10 cm. The frequency response, compensated for water frequency dispersion is shown in the left diagram, while the near-field pattern is represented on the right. The blue and red curves illustrate a

the antenna and water and skin, respectively (top and bottom).

(right diagram in Fig. 11).

cut along the E-Plane and the H-Plane, respectively.

In order to provide dielectrically matched surrounding conditions for the antenna body, the antennas were tested in de-ionized water. The results were then compared with data obtained in a more realistic environment, i.e. with the antenna put into contact with phantoms mimicking human tissues. The phantoms consisted of oil, water and some additives [24]. The dielectric permittivity ε' and the loss tangent ε''/ε' of the phantoms can be controlled by changing the percentage of oil [24], [25] (Section 5.2).

The frequency behavior of the reflection coefficient is shown in Fig. 9. It can be observed that while the reflection coefficient for the test against the phantom (with 40% oil, Section 5.2) approaches levels around –8 dB, the antenna is even better matched in water, leading to a further decrease of the reflection coefficient by 4 dB in the frequency range of interest. The compromise between input matching to a certain medium and the geometrical dimensions of the antenna denotes the key trade-off exploited for our design. In order to study the reflection occurring at the aperture plane, which is influenced by the permittivity matching between the dielectric medium composing the antenna body and the human skin, we performed time domain reflectometry (TDR) measurements by having the antenna radiate into different media [22].

**Figure 9.** Reflection coefficient measured in water (left) and on a skin mimicking phantom (right).

The results for water and skin (in-vivo) are shown in Fig. 10. The amplitude of the reflected signal with the antenna operating in water is significantly smaller than the one with the antenna operating on skin. This result indicates, in agreement with the frequency domain measurements, that the antenna is better matched to water than to skin. We note from Fig. 10 that the reflection occurring at the aperture due to impedance mismatch results in a signal with a longer decaying time. The larger the impedance mismatch is the longer the decaying time is. This feature is due to the fact that part of the reflected energy does not leave the antenna through the well-matched feed towards the signal source but remains within the antenna body.

268 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

measurements made in the time domain, and basic tests with volunteers.

controlled by changing the percentage of oil [24], [25] (Section 5.2).

radiate into different media [22].

within the antenna body.

A further issue of dielectrically scaled antennas is related to their measurement and characterization. As common measurement techniques and equipment cannot be applied, we followed three different strategies: measurements made in the frequency domain,

In order to provide dielectrically matched surrounding conditions for the antenna body, the antennas were tested in de-ionized water. The results were then compared with data obtained in a more realistic environment, i.e. with the antenna put into contact with phantoms mimicking human tissues. The phantoms consisted of oil, water and some additives [24]. The dielectric permittivity ε' and the loss tangent ε''/ε' of the phantoms can be

The frequency behavior of the reflection coefficient is shown in Fig. 9. It can be observed that while the reflection coefficient for the test against the phantom (with 40% oil, Section 5.2) approaches levels around –8 dB, the antenna is even better matched in water, leading to a further decrease of the reflection coefficient by 4 dB in the frequency range of interest. The compromise between input matching to a certain medium and the geometrical dimensions of the antenna denotes the key trade-off exploited for our design. In order to study the reflection occurring at the aperture plane, which is influenced by the permittivity matching between the dielectric medium composing the antenna body and the human skin, we performed time domain reflectometry (TDR) measurements by having the antenna

**Figure 9.** Reflection coefficient measured in water (left) and on a skin mimicking phantom (right).

The results for water and skin (in-vivo) are shown in Fig. 10. The amplitude of the reflected signal with the antenna operating in water is significantly smaller than the one with the antenna operating on skin. This result indicates, in agreement with the frequency domain measurements, that the antenna is better matched to water than to skin. We note from Fig. 10 that the reflection occurring at the aperture due to impedance mismatch results in a signal with a longer decaying time. The larger the impedance mismatch is the longer the decaying time is. This feature is due to the fact that part of the reflected energy does not leave the antenna through the well-matched feed towards the signal source but remains

**Figure 10.** Time domain reflections caused by the aperture plane due to dielectric mismatch between the antenna and water and skin, respectively (top and bottom).

The measurement of the radiating behavior is more complicated. To evaluate standard antenna parameters (e.g. gain, radiation pattern, etc.), the antennas should be placed in the Fraunhofer region. However, due to the high dielectric losses of water, the antenna could be placed at a maximum distance of 10 cm, which is not sufficient to meet the Fraunhofer region (starting from around 35 cm), but still is large enough to let the antennas operate in the Fresnel region. Near-field measurements are of main concern since the antennas are designed to be used in contact mode for biomedical imaging applications. The results show that the antennas present a flat frequency response (measured along the boresight direction), after the compensation of the frequency dispersion of the water (left diagram in Fig. 11), and 3 dB beam widths of nearly 20° for the E-Plane, and nearly 28° for the H-Plane (right diagram in Fig. 11).

**Figure 11.** Transmission behavior between two identical antennas operating in de-ionized water at a distance of 10 cm. The frequency response, compensated for water frequency dispersion is shown in the left diagram, while the near-field pattern is represented on the right. The blue and red curves illustrate a cut along the E-Plane and the H-Plane, respectively.

### 270 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

Eventually, in order to demonstrate the functionality of the ceramic antennas under realistic conditions, we have performed preliminary transmission measurements through a breast of a female volunteer, and the monitoring of the heartbeat, as illustrated in Figs. 12 and 13. The dynamic range of the achievable signal can be determined from Fig. 12 in comparison with the ideal transmission through a 4 cm path inside distilled water by a face-to-face arrangement of the antennas. The monitoring of the heartbeat was performed on a 35 years old healthy male volunteer. During measurement, the volunteer was sitting still and was holding breath in the state of maximal breathing in. The measurement was performed with an M-sequence radar in a bi-static configuration [Section 2.2]. Figure 13 shows the heartbeat signals as monitored. Upon Fourier transformation, we extracted a beat rate of nearly 75 beats per minute, which is considered normal for an adult. The results display very clearly the characteristic feature of heartbeat, thus manifesting a favorable dynamic range. This opens up promising applications for realistic monitoring and imaging tasks. Further details of these tests can be found in [26] and [22].

ultraMEDIS – Ultra-Wideband Sensing in Medicine 271

**Figure 13.** Monitored heartbeat signal of a 35 years old healthy male volunteer. Upon Fourier

Magnetic resonance imaging (MRI) systems are among the most sensitive diagnostic methods in medicine for the visualization of soft tissue [10]. At present, more than ten million MRI examinations of patients are performed per year worldwide. Given such a progressive development, further improvements of this diagnostic technique are under way. However, MRI systems are not *per se* capable of creating focused images of moving objects like the human heart or the thorax of the patient while breathing. Instead, additional techniques like breath holding, ECG triggering, or MR navigation methods are required. Such techniques either cause some inconvenience for the patient, or they are even not applicable for upcoming generations of MR scanners. A novel approach which overcomes these obstacles is the use of low-power multi-static UWB radar as a contactless navigator technology for MR tomography [29], [30]. To devise such navigators, the design of antennas compatible with MRI systems, i.e. antennas which do not interact neither electrically nor mechanically with the operation of the MR scan, is needed. The strategy to follow when designing an MRI compatible antenna is the minimization of mutual interaction between the metallized antennas and the strong static and gradient magnetic fields. Several experiments with conventional wideband antennas showed strong mechanical interactions, pointing out the need for special antenna designs [31]. The operational conditions inside an MR scanner are determined by three different types of fields. First, a static magnetic field *Bsta*t = 1.5 to 7 T provides a reference orientation of the nuclear spins of the regions under inspection. Furthermore, gradient magnetic fields with a slope of d*Bgrad*/d*t* = 50 T/s at the rising edge are switched during diagnostic measurements to allow for spatial (depth) information of the acquired molecular information. Finally, the nuclear spins are set into precession by a strong (KW range) RF signal at 42.58 MHz/T. The gradient fields induce eddy currents in the metallized sections of the antenna which, in turn, interact with the static magnetic field by exerting a mechanical torque on the antenna structure. The torque can reach peak values of the order of 0.045 Nm for a contiguous metallized area of 20 mm × 30 mm. This value is high enough to result in mechanical amplitudes of several millimeters, deforming or moving the antenna structure, especially in the case of mechanical resonances. Furthermore, the magnetic fields of the eddy currents can lead to artifacts of the MR-image. These interactions inhibit the beneficial application of UWB navigation for magnetic resonance imaging and,

transformation, we extracted a beat rate of nearly 75 beats per minute.

*2.3.3. MRI compatible antennas* 

The full dielectric scaling, as previously stated, offers the potential for a further size reduction of the antenna. Accordingly, we continued our research and succeeded in developing a ceramic DRH antenna with an aperture of only 16 mm × 11 mm and a lower cut-off frequency around 1.5 GHz. Due to the input impedance of the antenna below 10 , active receive and transmit versions are under development in the framework of HALOS (Chapter 14), employing an UWB low-noise subtraction circuitry and power amplification [27]. The manufacture of the tiny antenna followed similar production steps as for the previous version. It proved quite challenging because of the reduced size, requiring additional specific production steps and iterative testing procedures [28].

**Figure 12.** Transmission measurements through a female breast performed with the ceramic doubleridged horn antennas in comparison with a reference measurement of 4 cm distilled water (upper curve).

**Figure 13.** Monitored heartbeat signal of a 35 years old healthy male volunteer. Upon Fourier transformation, we extracted a beat rate of nearly 75 beats per minute.

### *2.3.3. MRI compatible antennas*

270 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

of these tests can be found in [26] and [22].

(upper curve).

Eventually, in order to demonstrate the functionality of the ceramic antennas under realistic conditions, we have performed preliminary transmission measurements through a breast of a female volunteer, and the monitoring of the heartbeat, as illustrated in Figs. 12 and 13. The dynamic range of the achievable signal can be determined from Fig. 12 in comparison with the ideal transmission through a 4 cm path inside distilled water by a face-to-face arrangement of the antennas. The monitoring of the heartbeat was performed on a 35 years old healthy male volunteer. During measurement, the volunteer was sitting still and was holding breath in the state of maximal breathing in. The measurement was performed with an M-sequence radar in a bi-static configuration [Section 2.2]. Figure 13 shows the heartbeat signals as monitored. Upon Fourier transformation, we extracted a beat rate of nearly 75 beats per minute, which is considered normal for an adult. The results display very clearly the characteristic feature of heartbeat, thus manifesting a favorable dynamic range. This opens up promising applications for realistic monitoring and imaging tasks. Further details

The full dielectric scaling, as previously stated, offers the potential for a further size reduction of the antenna. Accordingly, we continued our research and succeeded in developing a ceramic DRH antenna with an aperture of only 16 mm × 11 mm and a lower cut-off frequency around 1.5 GHz. Due to the input impedance of the antenna below 10 , active receive and transmit versions are under development in the framework of HALOS (Chapter 14), employing an UWB low-noise subtraction circuitry and power amplification [27]. The manufacture of the tiny antenna followed similar production steps as for the previous version. It proved quite challenging because of the reduced size, requiring

**Figure 12.** Transmission measurements through a female breast performed with the ceramic double-

ridged horn antennas in comparison with a reference measurement of 4 cm distilled water

additional specific production steps and iterative testing procedures [28].

Magnetic resonance imaging (MRI) systems are among the most sensitive diagnostic methods in medicine for the visualization of soft tissue [10]. At present, more than ten million MRI examinations of patients are performed per year worldwide. Given such a progressive development, further improvements of this diagnostic technique are under way. However, MRI systems are not *per se* capable of creating focused images of moving objects like the human heart or the thorax of the patient while breathing. Instead, additional techniques like breath holding, ECG triggering, or MR navigation methods are required. Such techniques either cause some inconvenience for the patient, or they are even not applicable for upcoming generations of MR scanners. A novel approach which overcomes these obstacles is the use of low-power multi-static UWB radar as a contactless navigator technology for MR tomography [29], [30]. To devise such navigators, the design of antennas compatible with MRI systems, i.e. antennas which do not interact neither electrically nor mechanically with the operation of the MR scan, is needed. The strategy to follow when designing an MRI compatible antenna is the minimization of mutual interaction between the metallized antennas and the strong static and gradient magnetic fields. Several experiments with conventional wideband antennas showed strong mechanical interactions, pointing out the need for special antenna designs [31]. The operational conditions inside an MR scanner are determined by three different types of fields. First, a static magnetic field *Bsta*t = 1.5 to 7 T provides a reference orientation of the nuclear spins of the regions under inspection. Furthermore, gradient magnetic fields with a slope of d*Bgrad*/d*t* = 50 T/s at the rising edge are switched during diagnostic measurements to allow for spatial (depth) information of the acquired molecular information. Finally, the nuclear spins are set into precession by a strong (KW range) RF signal at 42.58 MHz/T. The gradient fields induce eddy currents in the metallized sections of the antenna which, in turn, interact with the static magnetic field by exerting a mechanical torque on the antenna structure. The torque can reach peak values of the order of 0.045 Nm for a contiguous metallized area of 20 mm × 30 mm. This value is high enough to result in mechanical amplitudes of several millimeters, deforming or moving the antenna structure, especially in the case of mechanical resonances. Furthermore, the magnetic fields of the eddy currents can lead to artifacts of the MR-image. These interactions inhibit the beneficial application of UWB navigation for magnetic resonance imaging and,

### 272 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

therefore, must be avoided. We used a 3-T MR scanner with the resulting RF frequency of 127.8 MHz, which is ten times smaller than the lower cut-off frequency of the UWB antennas employed. As the frequency response of a typical antenna corresponds to a high-pass filter of first order, the stop-band attenuation amounts to 20 dB per decade, indicating the risk of collecting RF power even in the presence of path-loss and shadowing.

ultraMEDIS – Ultra-Wideband Sensing in Medicine 273

part of the antenna showed little frequency dependence. Based on these observations, a compromise was sought to reduce the plane metallization with the minimal possible distortion of the broadband current distribution. As a result, the conductor faces of the horn sections were separated into strip lines, straight and elliptically shaped, separated by 1 mm, and oriented parallel to the most common current paths, with plain connections at the face edges only. The central part of the right image of Fig. 14 illustrates the resulting geometric arrangement of the slots, while the normalized surface current of the modified antenna at 5 GHz is shown in the right-most part. The main features of the current distribution could be sustained qualitatively both on the pyramidal faces and the backward cubical part of the antenna. Differences occurred mainly for the currents oriented perpendicular to the slots. It is this minor change in current distribution which causes the modified radiation properties discussed above. The ridges themselves required special attention. A grid of holes was eventually identified as the proper solution to reduce the metallization area of the ridges without disturbing the high-frequency current distribution too much. In order to reduce the maximal loop size for low-frequency currents, the outer contour of the horn section was cut

**Figure 14.** MR-compatible double-ridged horn antenna for a lower cut-off frequency of 1.5 GHz (top). The lower image shows the simulated, normalized current distribution of an unmodified DRH antenna at 5 GHz (left-hand part), the layout of the resulting MR-compatible DRH antenna (center part), and the

current distribution of the modified DRH antenna at 5 GHz (right-hand part).

and shortened by standard surface-mounted device capacitors.

The minimization of contiguous metallized area and, hence, eddy currents, is therefore the main issue to be addressed by the antenna design. Additional constraints arise from the intended applications in biomedical diagnostics: weakly frequency-dependent radiation patterns over the entire operational bandwidth, good decoupling between neighboring antennas, and a lower cut-off frequency around 1 GHz. The DRH antenna was identified to be a suitable UWB antenna type to accomplish these requirements. Due to the functional principle of DRH antennas, the minimization of contiguous metallized areas and the realization of a weakly frequency-dependent radiation pattern are in conflict with each other. Horn antennas are typically made entirely out of metallic parts of high electrical conductivity σ, thus suffering from the induction of eddy currents under MR-scanner conditions. Therefore, the major challenge was to modify the double-ridged horn antenna to achieve MR-compatibility, without compromising the favorable radiation properties.

Inspired by commercial counterparts of DRH antennas, we removed the H-plane sidewalls of the pyramidal horn, leaving just a thin wire in the plane of the aperture, as illustrated by the left picture in Fig. 14. As a result, the lower cut-off frequency could be reduced from 2.6 GHz to 1.5 GHz for otherwise unchanged dimensions and operation in air. The comparison with a conventional double-ridged horn antenna with a similar bandwidth revealed that this improvement was achieved at the expense of increased beam width, sidelobes and backward radiation, predominantly at frequencies below 3 GHz, due to the modified aperture distribution and diffraction at the edges of the open construction. The increased beam width led to a slightly increased crosstalk [32]. It can easily be compensated for by re-orienting the antennas relative to each other. While the crosstalk for conventional DRH antennas becomes small for an H-plane alignment, the MR-compatible versions have to be aligned along the E-plane due to the removed H-plane sidewalls and, thus, reduced shielding.

The thickness of the metallization was also reduced in order to exploit the skin effect for a decoupling of the low-frequency eddy current paths. The metal planes were replaced by metallized dielectric boards with a metallization thickness of 12 μm (Fig. 14). This value corresponds to about twice the skin depth at the lowest frequency used. The high-frequency currents determining the radiation of the antenna remain essentially undisturbed while the eddy currents in the KHz range are strongly attenuated by the high sheet resistance. For further optimization of the remaining metallized areas, the distribution of surface currents in the UWB frequency range was inspected by electromagnetic simulations (right image of Fig. 14). Typical results for the normalized surface current are illustrated at 5 GHz (leftmost). The surface current is concentrated near the position of the ridge and the edges of the pyramidal frame. According to our expectations, the number of current loops was found to increase with frequency; in contrast, the current distribution across the backward cubical part of the antenna showed little frequency dependence. Based on these observations, a compromise was sought to reduce the plane metallization with the minimal possible distortion of the broadband current distribution. As a result, the conductor faces of the horn sections were separated into strip lines, straight and elliptically shaped, separated by 1 mm, and oriented parallel to the most common current paths, with plain connections at the face edges only. The central part of the right image of Fig. 14 illustrates the resulting geometric arrangement of the slots, while the normalized surface current of the modified antenna at 5 GHz is shown in the right-most part. The main features of the current distribution could be sustained qualitatively both on the pyramidal faces and the backward cubical part of the antenna. Differences occurred mainly for the currents oriented perpendicular to the slots. It is this minor change in current distribution which causes the modified radiation properties discussed above. The ridges themselves required special attention. A grid of holes was eventually identified as the proper solution to reduce the metallization area of the ridges without disturbing the high-frequency current distribution too much. In order to reduce the maximal loop size for low-frequency currents, the outer contour of the horn section was cut and shortened by standard surface-mounted device capacitors.

272 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

collecting RF power even in the presence of path-loss and shadowing.

therefore, must be avoided. We used a 3-T MR scanner with the resulting RF frequency of 127.8 MHz, which is ten times smaller than the lower cut-off frequency of the UWB antennas employed. As the frequency response of a typical antenna corresponds to a high-pass filter of first order, the stop-band attenuation amounts to 20 dB per decade, indicating the risk of

The minimization of contiguous metallized area and, hence, eddy currents, is therefore the main issue to be addressed by the antenna design. Additional constraints arise from the intended applications in biomedical diagnostics: weakly frequency-dependent radiation patterns over the entire operational bandwidth, good decoupling between neighboring antennas, and a lower cut-off frequency around 1 GHz. The DRH antenna was identified to be a suitable UWB antenna type to accomplish these requirements. Due to the functional principle of DRH antennas, the minimization of contiguous metallized areas and the realization of a weakly frequency-dependent radiation pattern are in conflict with each other. Horn antennas are typically made entirely out of metallic parts of high electrical conductivity σ, thus suffering from the induction of eddy currents under MR-scanner conditions. Therefore, the major challenge was to modify the double-ridged horn antenna to

achieve MR-compatibility, without compromising the favorable radiation properties.

shielding.

Inspired by commercial counterparts of DRH antennas, we removed the H-plane sidewalls of the pyramidal horn, leaving just a thin wire in the plane of the aperture, as illustrated by the left picture in Fig. 14. As a result, the lower cut-off frequency could be reduced from 2.6 GHz to 1.5 GHz for otherwise unchanged dimensions and operation in air. The comparison with a conventional double-ridged horn antenna with a similar bandwidth revealed that this improvement was achieved at the expense of increased beam width, sidelobes and backward radiation, predominantly at frequencies below 3 GHz, due to the modified aperture distribution and diffraction at the edges of the open construction. The increased beam width led to a slightly increased crosstalk [32]. It can easily be compensated for by re-orienting the antennas relative to each other. While the crosstalk for conventional DRH antennas becomes small for an H-plane alignment, the MR-compatible versions have to be aligned along the E-plane due to the removed H-plane sidewalls and, thus, reduced

The thickness of the metallization was also reduced in order to exploit the skin effect for a decoupling of the low-frequency eddy current paths. The metal planes were replaced by metallized dielectric boards with a metallization thickness of 12 μm (Fig. 14). This value corresponds to about twice the skin depth at the lowest frequency used. The high-frequency currents determining the radiation of the antenna remain essentially undisturbed while the eddy currents in the KHz range are strongly attenuated by the high sheet resistance. For further optimization of the remaining metallized areas, the distribution of surface currents in the UWB frequency range was inspected by electromagnetic simulations (right image of Fig. 14). Typical results for the normalized surface current are illustrated at 5 GHz (leftmost). The surface current is concentrated near the position of the ridge and the edges of the pyramidal frame. According to our expectations, the number of current loops was found to increase with frequency; in contrast, the current distribution across the backward cubical

**Figure 14.** MR-compatible double-ridged horn antenna for a lower cut-off frequency of 1.5 GHz (top). The lower image shows the simulated, normalized current distribution of an unmodified DRH antenna at 5 GHz (left-hand part), the layout of the resulting MR-compatible DRH antenna (center part), and the current distribution of the modified DRH antenna at 5 GHz (right-hand part).

ultraMEDIS – Ultra-Wideband Sensing in Medicine 275

The MRI compatible DRH antennas were implemented as part of a UWB MR navigator, by means of which it was possible to take images of the myocardium for the first time without using an ECG as navigator. The quality achieved was comparable with the one achievable

**Figure 17.** Two-dimensional representation of the measured time domain response of the MRcompatible DRH antenna for the E-plane (left) and the H-plane (right) through the main beam. The

The electrical properties of biological tissues and cell suspensions have been of interest for over a century for many reasons. They determine the pathways of current flow through the body and are very important for the analysis of a wide range of biomedical applications such as functional electrical stimulation and the diagnosis and treatment of various physiological conditions with weak electric currents, radio-frequency hyperthermia, electrocardiography, and body composition. On a more fundamental level, the knowledge of these electrical properties can lead to an understanding of the underlying basic biological processes. Indeed, biological impedance studies have long been an important issue in electrophysiology and biophysics; interestingly, one of the first demonstrations of the existence of the cell membrane was based on dielectric studies on cell suspensions [33].

Biological tissues are a mixture of water, ions, and organic molecules organized in cells, subcellular structures, and membranes, and its dielectric properties are highly frequencydependent in the range from Hz to GHz. The spectrum is characterized by three main dispersion regions referred to as α, β, and γ regions at low, intermediate, and high frequencies [34]. Biological materials can show large dispersions, especially at low frequencies (Fig. 18). Low frequencies are mainly caused by interfacial polarizations at the surfaces between the different materials of which a cell is composed [35]. Reviews of the dielectric properties of cells and the different dispersions are given in the literature [36], [37].

scales indicate the normalized impulse response of the antenna.

**3.1. Impact on living cells** 

**3. Weak electromagnetic fields and biological tissue** 

with the ECG navigator (see Section 4).

**Figure 15.** Measured reflection coefficient of the MR-compatible double-ridged horn antenna (lower curve) and the measured antenna gain (upper curve) versus frequency.

Fig. 15 displays measured results for the reflection coefficient and the gain of the modified DRH antenna. A return loss above 10 dB was achieved over the frequency range from 1.5 to 12 GHz. Radiation measurements in an anechoic chamber yielded the radiation patterns illustrated in Fig. 16 for two orthogonal cuts with respect to the plane of the ridges. The halfpower beam width, indicated as the black contour line in Fig. 16, was found to vary between 30 and 50 degrees, thus covering a range suitable for the envisaged applications. Except for frequencies around 2 GHz, the main lobe showed little spectral variation. The corresponding frequency variation of the antenna gain is displayed in Fig. 16. These results were found in good agreement with the numerical simulations.

**Figure 16.** Two-dimensional representation of the measured radiation pattern of the MR-compatible double-ridged horn antenna for the E-plane (left) and the H-plane (right) through the main beam. The scales indicate the antenna gain in dBi. The black and white contour lines illustrate the corresponding beam widths at 3 and 10 dB below the frequency-dependent maximum gain, respectively.

The transient response of the antenna is shown in Fig. 17. Despite the open geometry of the MR-compatible antenna, a low signal distortion could be sustained. The slight angular dependence of the time responses can be attributed to an offset between the phase centers of the antennas and the center of rotation of the antenna positioning system.

The MRI compatible DRH antennas were implemented as part of a UWB MR navigator, by means of which it was possible to take images of the myocardium for the first time without using an ECG as navigator. The quality achieved was comparable with the one achievable with the ECG navigator (see Section 4).

**Figure 17.** Two-dimensional representation of the measured time domain response of the MRcompatible DRH antenna for the E-plane (left) and the H-plane (right) through the main beam. The scales indicate the normalized impulse response of the antenna.
