**3.1. Signal definition**

Fig. 1 depicts a pulsed angle modulated UWB signal consisting of a sequence of short pulses (width *Td*, period *Ts*), in which each pulse is an oscillation with the frequency *ωosc* and the modulated initial phase *ϕi*:

$$s(t) = \sum\_{i=0}^{N} \cos\left(\omega\_{0\text{sc}}\left(t - i \cdot T\_s + \frac{T\_d}{2}\right) + \varphi\_i\right) \cdot \text{rect}\left(\frac{t - i \cdot T\_s}{T\_d}\right) \tag{1}$$

with

2 UKoLoS

mentioned mm-wave UWB bands, the permitted maximum mean power density is at least 38

Most of the mm-wave UWB communication and ranging systems published so far use a simple pulse generator as signal source. In the simplest case, a mm-wave CW carrier is modulated with an ASK (s. e.g [17]) or BPSK (s. e.g. [18])) sequence. A very interesting low-power approach that is somewhat related to the approach in this work is shown in [6] and [7]. Here, a 60 GHz oscillator itself is switched on and off. To guarantee a stable startup phase and to improve the phase noise, the oscillator is injection locked to a spurious harmonic of the switching signal. The benefit of the pulsed injection locking approach with respect to power consumption was impressively shown in this work. The general approach to obtain a stable pulse to pulse phase condition by injecting a spurious harmonic of the switching pulse into the oscillator is well known for a long time from low-power and low-cost microwave primary pulse radar systems. This basic principle can be extended in a way that frequency modulated signals can be generated based on a switched injection locked oscillator [19]. In this work, it is generalized for synthesizing arbitrarily phase modulated signals for integrated local positioning and communication. The fusion of positioning and communication capability is especially needed for future wireless devices applied in the "internet of things" or for advanced multimedia / augmented reality applications, for robot control and for vehicle2X /

Most existing UWB communication and ranging systems - especially those dedicated to low power consumption and mm-wave frequencies - employ simple impulse radios (IR). Popular IR-UWB modulation techniques include on-off keying (OOK), pulse-position modulation (PPM), pulse-amplitude modulation (PAM) and binary phase shift keying (BPSK) [5, 17, 18]. Their waveform can be synthesized using low complexity impulse generators and control circuitry, which comes at the cost of low spectral efficiency and severely limited control over spectral properties of the synthesized signals. Consequently, these transmitter cannot exhaust regulatory boundaries in all operation modes. High data rate synthesizers are often average power limited whereas low data rate implementations may be peak power limited [20].

In order to overcome these issues, pulsed angle modulated UWB signals are proposed to provide greater flexibility and better control over the spectral properties of the synthesized signals. Additionally, this signal type is well suited for both ranging and communication, since it allows synthesizing pulsed frequency modulated chirps that are attractive for ranging as well as digital phase modulation schemes for data transmission with the same hardware. Since classic architectures containing VCOs, PLLs, mixer, linear amplifiers and switches are not suited for low complexity, low power systems, the switched injection-locked oscillator is suggested for signal synthesis. It regenerates and amplifies a weak phase-modulated signal. Consequently, the high frequency RF signal can be generated from a high power but efficiently synthesized low frequency phase modulated baseband signal in two simple stages - a lossy passive or low power frequency multiplier (harmonic generator) and a switched

In this work, it is demonstrated that this approach allows synthesizing pulsed, arbitrarily phase modulated signals using the switched injection-locked harmonic sampling principle. The theory of this concept was investigated thoroughly and verified experimentally for the synthesis of phase shift keying (PSK) modulated communication signals and pulsed frequency modulated (PFM) radar signals with the same hardware. Regarding the switched

injection-locked oscillator as single stage pulsed high gain (> 50 dB) amplifier.

dB higher than in the UWB bands below 30 GHz.

**2. Proposed concepts and components**

car2X applications.

$$\text{rect}\left(\mathbf{x}\right) = \begin{cases} 1 & \text{for } -0.5 < \mathbf{x} < 0.5\\ 0 & \text{else} \end{cases}$$

For flexible signal synthesis, initial phase modulation, pulse period, pulse width and oscillation frequency can be tuned.

**Figure 1.** Pulsed angle modulated UWB signal - the modulated parameter is the initial phase *ϕ<sup>i</sup>* of each pulse

### **3.2. SILO operation principle**

The switched injection-locked oscillator (SILO) is basically a normal oscillator which is turned on and off while a weak reference signal is injected into its feedback loop (see Fig. 2). During startup of the oscillator, the injection signal provides an initial condition in the oscillator's resonator instead of noise like in oscillators without injection signal. This way, the

#### 4 UKoLoS 346 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems <sup>5</sup>

instantaneous phase of the injection signal is adopted though the oscillator runs with its own natural frequency, which may differ from the injection signal's frequency. Since the power level of the injection signal is far too low to influence the oscillation as soon as the oscillator has reached its final amplitude, it performs only phase, but no frequency locking.

**3.3. Phase sampling theory**

+∞ ∑ *i*=−∞

transform *F* {·} of (4) leads to:

 *a* · *e j* 

*S*(*ω*) = *A* ·

less than half pulse repetition frequency).

**Figure 3.** SILO output spectrum according to (5)

*-*

*-*

*<sup>j</sup>ϕinj <sup>t</sup>*<sup>−</sup> *Td* 2 

·*F*{*e*

*s*(*t*) =

In [3, 4, 19], the SILO's phase sampling principle and its applications have been investigated thoroughly. The most important results will be summarized and discussed in the following. Starting from equations (2) and (3), the SILO's output signal can be expressed by (disregarding

Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems 347

This expression still suggests an oscillation with *ωosc* - the presence of the injection signal regeneration feature that includes the frequency is not obvious. According to [4], the Fourier

> ·

*<sup>ω</sup>* <sup>−</sup> *<sup>ω</sup>inj*

*<sup>i</sup>*·*Ts*<sup>−</sup> *Td* 2 

*<sup>e</sup>j*(*ωinj*−*ωosc* ) *Td*

∗ X <sup>1</sup> *Ts ω* 2*π*


*-*



2

· rect *<sup>t</sup>* <sup>−</sup> *<sup>i</sup>* · *Ts Td*

. (4)

. (5)

negative frequencies and finite time domain waveform length for sake of simplicity):

 *<sup>i</sup>*·*Ts*<sup>−</sup> *Td* 2 · *e <sup>j</sup>ϕinj*

sinc (*<sup>ω</sup>* <sup>−</sup> *<sup>ω</sup>osc*) · *Td* 2

}(*ω*) ∗ *δ*

The SILO output spectrum according to (5) consists of a convolution of the user-defined phase modulation spectrum with its center / carrier frequency signal and the sampling process' aliasing signal (Dirac comb, X), see Fig. 3. It is weighted with a sinc envelope centered at the oscillator's natural frequency *ωosc*. Since this frequency only affects the envelope and a constant phase offset, the SILO can be regarded as a highly effective aliased regenerative amplifier. In consequence, an injected user-defined constant envelope phase modulated signal is reproduced correctly even with a free running oscillator with (in certain bounds) unknown natural frequency as long as Nyquist's sampling theorem is fulfilled (modulation bandwidth

In general, this signal synthesis principle is not limited to phase modulated / constant envelope signal synthesis. For amplitude modulation, e.g. an electronically tuned attenuator at the SILO's output can be employed to manipulate the amplitude of each pulse synchronously to the pulse rate, which leads to a polar modulator. Since efficient pulse

*-*

*-*


*<sup>ω</sup>osc <sup>t</sup>*+(*ωinj*−*ωosc* )·

**Figure 2.** SILO principle

This behavior can be described theoretically by:

$$s(t) = \sum\_{i=0}^{N} \cos\left(\omega\_{osc}\left(t - i \cdot T\_s + \frac{T\_d}{2}\right) + \arg\left\{s\_{inj}(t)\right\}\right) \cdot \text{rect}\left(\frac{t - i \cdot T\_s}{T\_d}\right),\tag{2}$$

with the injection signal (center/reference frequency *ωinj*, phase modulation *ϕ*(*t*))

$$s\_{\rm inj}(t) = \cos\left(\omega\_{\rm inj}t + \varphi\_{\rm inj}(t)\right), \quad \text{arg}\left\{s\_{\rm inj}(t)\right\} = \omega\_{\rm inj}t + \varphi\_{\rm inj}(t). \tag{3}$$

In spite of the fact that this model only describes the fundamental principle, the physical behavior of the oscillator is very similar in most operation modes. The most important disregarded physical effects observed in real implementations are:


Thus, the simplifications of the proposed ideal model mainly affect time and frequency domain amplitude shape, which makes this model suitable for the analysis of the phase sampling process.

346 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems <sup>5</sup> Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems 347

### **3.3. Phase sampling theory**

4 UKoLoS

instantaneous phase of the injection signal is adopted though the oscillator runs with its own natural frequency, which may differ from the injection signal's frequency. Since the power level of the injection signal is far too low to influence the oscillation as soon as the oscillator

> *-*-

 

 

· rect

 *<sup>t</sup>* <sup>−</sup> *<sup>i</sup>* · *Ts Td*

= *ωinjt* + *ϕinj*(*t*). (3)

, (2)

*-*-

 

 

*Td* 2 + arg *sinj*(*t*)

 , arg *sinj*(*t*) 

In spite of the fact that this model only describes the fundamental principle, the physical behavior of the oscillator is very similar in most operation modes. The most important

• Due to balancing imperfections e.g. in differential oscillators, high order harmonics of the startup pulse turning on the circuit cause self-locking effects that degrade the SILO's performance at low injection levels. Hence, the rise time of the oscillator should not be too short in order to reduce the harmonic power level. Obviously, this leads to a trade-off with spectral bandwidth, minimum pulse width and maximum achievable pulse repetition rate. • The phase sampling process is affected by the amplitude of the injection signal. In consequence, amplitude variations of the injection signal are converted into phase distortions. Therefore, constant amplitude injection signals should be used to mitigate these effects. Then there is only a constant phase offset between injection and regenerated

• If the rise time of the oscillator is configured to be relatively long compared to the pulse width, there will be a noticeable dependence between the injection signal's power level and pulse width. With a large amplitude injection signal, the oscillator settles much faster than when starting from noise level. Again, constant amplitude injection signals are the

Thus, the simplifications of the proposed ideal model mainly affect time and frequency domain amplitude shape, which makes this model suitable for the analysis of the phase

with the injection signal (center/reference frequency *ωinj*, phase modulation *ϕ*(*t*))

*t* − *i* · *Ts* +

*ωinjt* + *ϕinj*(*t*)

disregarded physical effects observed in real implementations are:

preferred countermeasure to avoid pulse width jitter.

has reached its final amplitude, it performs only phase, but no frequency locking.

 

This behavior can be described theoretically by:

**Figure 2.** SILO principle

*s*(*t*) =

signal.

sampling process.

*N* ∑ *i*=0 cos *ωosc* 

*sinj*(*t*) = cos

In [3, 4, 19], the SILO's phase sampling principle and its applications have been investigated thoroughly. The most important results will be summarized and discussed in the following.

Starting from equations (2) and (3), the SILO's output signal can be expressed by (disregarding negative frequencies and finite time domain waveform length for sake of simplicity):

$$s(t) = \sum\_{i = -\infty}^{+\infty} \left[ a \cdot e^{j\left(\omega\_{\rm{ok}}t + \left(\omega\_{\rm{ini}} - \omega\_{\rm{ok}}\right) \cdot \left(i \cdot T\_s - \frac{T\_d}{2}\right)\right)} \cdot e^{j\varphi\_{\rm{inj}}\left(i \cdot T\_s - \frac{T\_d}{2}\right)} \cdot \text{rect}\left(\frac{t - i \cdot T\_s}{T\_d}\right) \right]. \tag{4}$$

This expression still suggests an oscillation with *ωosc* - the presence of the injection signal regeneration feature that includes the frequency is not obvious. According to [4], the Fourier transform *F* {·} of (4) leads to:

$$S(\omega) = A \cdot \left[ \text{sinc}\left(\frac{(\omega - \omega\_{\text{os}\varepsilon}) \cdot T\_d}{2}\right) \cdot \left(e^{j\left(\omega\_{\text{inj}} - \omega\_{\text{os}\varepsilon}\right) \frac{T\_d}{2}}\right) \right. \tag{5}$$

$$\cdot F\{e^{j\varrho\_{\text{inj}}\left(t - \frac{T\_d}{2}\right)}\}(\omega) \* \delta\left(\omega - \omega\_{\text{inj}}\right) \* \text{III}\_1\left(\frac{\omega}{2\pi}\right)\}\Big]. \tag{5}$$

The SILO output spectrum according to (5) consists of a convolution of the user-defined phase modulation spectrum with its center / carrier frequency signal and the sampling process' aliasing signal (Dirac comb, X), see Fig. 3. It is weighted with a sinc envelope centered at the oscillator's natural frequency *ωosc*. Since this frequency only affects the envelope and a constant phase offset, the SILO can be regarded as a highly effective aliased regenerative amplifier. In consequence, an injected user-defined constant envelope phase modulated signal is reproduced correctly even with a free running oscillator with (in certain bounds) unknown natural frequency as long as Nyquist's sampling theorem is fulfilled (modulation bandwidth less than half pulse repetition frequency).

In general, this signal synthesis principle is not limited to phase modulated / constant envelope signal synthesis. For amplitude modulation, e.g. an electronically tuned attenuator at the SILO's output can be employed to manipulate the amplitude of each pulse synchronously to the pulse rate, which leads to a polar modulator. Since efficient pulse

**Figure 3.** SILO output spectrum according to (5)

amplitude modulation is feasible for a long time in contrast to complex phase modulation and can be added independently, this work is concentrated on the latter aspect.

### **3.4. Phase modulated UWB communication signals**

For the synthesis of communication signals [4], any phase modulated constant envelope signal that is bandwidth limited to half pulse repetition frequency can be chosen. The maximum possible symbol rate leads to one symbol per pulse.

Demodulation can be achieved similar to existing approaches that allow quadrature pulse demodulation (e.g. [11]). Basically, the phase of each pulse has to be sampled synchronously to the pulse sequence (i.e. during pulse duration), which can be realized e.g. by quadrature baseband down-conversion and synchronized sample acquisition. In this case, the sequence of received samples is given by

$$s\_{\rm recv}(k) = s\left(k \cdot T\_s + \Delta t\_{\rm sync}\right) \cdot e^{-j\omega\_{\rm inj}\left(k \cdot T\_s + \Delta t\_{\rm sync}\right)}, \quad k \in \mathbb{N},\tag{6}$$

given that transmitter and receiver were precisely synchronized, which can be achieved

Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems 349

 -


Strictly speaking, the sampling theorem is not met for a sweep bandwidth larger than the pulse repetition frequency. Though, aliasing can be exploited to minimize the ramp synthesis effort (see Fig. 4). The injected and regenerated signal is configured to represent a short chirp within the sampling bandwidth that is repeated continuously. Considering aliasing, the resulting signal appears to be continuous at the receiver when sweeping through all aliases. The required effort can even be further reduced: Since the SILO only samples certain phase values, it is not necessary to actually generate continuous sweeps as intermediate signal. Instead, a CW injection signal with stepped phase modulation is sufficient as long as its phase (modulus 2*π*) equals (8) at sampling time. This approach results according to [3] in a short

> *pBT*<sup>2</sup> *s*

The only restriction that results from exploiting aliases is a limitation in unambiguous range,

*dmax* <sup>=</sup> *cpT BTs*

Considering a sampling period of 100 ns (*Ts* = 10 MHz), which is convenient for low power implementations, a sufficient maximum range of over 1 km can be achieved even at a high

In the following, concepts and implementations for the pulsed angle modulated signal synthesis principle are presented. Firstly, the harmonic sampling approach is presented,

**Figure 4.** Exploiting sampling aliases to synthesize frequency modulated UWB radar signals with

periodic sequence of samples (period *<sup>p</sup>* <sup>∈</sup> **<sup>N</sup>**+) under the condition that the term

is whole-number and *p* even. The sequence features a minimum period of

*pmin* <sup>=</sup> *<sup>T</sup>*

*BT*<sup>2</sup> *s*

 *-*

*-*

*-*

*<sup>T</sup>* (12)

, *pmin* <sup>∈</sup> **<sup>N</sup>**+. (13)

. (14)

*-*

through two-way synchronization like in [16].


 

> - -

minimal effort


 

i.e. maximum distance (phase velocity *cp*):

bandwidth of 2 GHz in 1 ms.

**4. System concepts**

where Δ*tsync* denotes a modestly (uncertainty less than half pulse width) unknown synchronization error that has to be taken into account in practice. Inserting (4) in (6) leads to:

$$s\_{\rm rect}(k) = A\_I \cdot e^{j\left(\varphi\_{\rm inj}\left(i \cdot T\_s - \frac{T}{2}\right) + \left(\omega\_{\rm osc} - \omega\_{\rm inj}\right) \cdot \left(\Delta t\_{sync} + \frac{T}{2}\right)\right)}.\tag{7}$$

Accordingly, the original phase modulation *ϕinj* is reconstructed correctly aside from a constant phase offset. Its constancy is guaranteed as long as the natural frequency of the unstabilized oscillator does not drift too fast, which is mostly given due to relatively slow changes in environmental parameters like temperature. For compensation, e.g. differential modulation schemes or short frames can be applied.

### **3.5. Frequency modulated UWB radar signals**

Since the SILO based synthesizer is capable of generating any constant envelope phase modulated signals (within the bandwidth limit), even a frequency modulated radar signal with the bandwidth *B*, sweep duration *T* and phase

$$
\varphi\_{\rm inj,FM}(t) = 2\pi \frac{B}{2T} t^2 \tag{8}
$$

can be transmitted. At the receiver, the time delayed transmit signal *s*(*t*) is mixed with a FMCW signal:

$$s\_{\text{rec}\upsilon, FM}(t) = s(t - t\_d) \cdot e^{-j\left(\omega\_{ln}t + \pi \frac{B}{T} t^2\right)}.\tag{9}$$

According to [3], the approximate resulting beat frequency spectrum (disregarding envelope)

$$S\_{\rm recv,FM}(\omega) = \underline{A} \cdot \delta\left(\omega + 2\pi \frac{B}{T} \left(t\_d + \frac{T\_d}{2}\right)\right) \* \text{III}\_{\frac{1}{T\_s}}(\omega) \tag{10}$$

is equivalent to the conventional FMCW spectrum except for the aliases resulting from switched operation and a constant phase offset *A*. The (one way) distance can be calculated from

$$f\_b = \frac{B}{T} \left( t\_d + \frac{T\_d}{2} \right) \tag{11}$$

given that transmitter and receiver were precisely synchronized, which can be achieved through two-way synchronization like in [16].

**Figure 4.** Exploiting sampling aliases to synthesize frequency modulated UWB radar signals with minimal effort

Strictly speaking, the sampling theorem is not met for a sweep bandwidth larger than the pulse repetition frequency. Though, aliasing can be exploited to minimize the ramp synthesis effort (see Fig. 4). The injected and regenerated signal is configured to represent a short chirp within the sampling bandwidth that is repeated continuously. Considering aliasing, the resulting signal appears to be continuous at the receiver when sweeping through all aliases.

The required effort can even be further reduced: Since the SILO only samples certain phase values, it is not necessary to actually generate continuous sweeps as intermediate signal. Instead, a CW injection signal with stepped phase modulation is sufficient as long as its phase (modulus 2*π*) equals (8) at sampling time. This approach results according to [3] in a short periodic sequence of samples (period *<sup>p</sup>* <sup>∈</sup> **<sup>N</sup>**+) under the condition that the term

$$\frac{pBT\_s^2}{T} \tag{12}$$

is whole-number and *p* even. The sequence features a minimum period of

$$p\_{\rm min} = \frac{T}{BT\_{\rm s}^2}, \quad p\_{\rm min} \in \mathbb{N}^+. \tag{13}$$

The only restriction that results from exploiting aliases is a limitation in unambiguous range, i.e. maximum distance (phase velocity *cp*):

$$d\_{\max} = \frac{c\_p T}{B T\_s}.\tag{14}$$

Considering a sampling period of 100 ns (*Ts* = 10 MHz), which is convenient for low power implementations, a sufficient maximum range of over 1 km can be achieved even at a high bandwidth of 2 GHz in 1 ms.

### **4. System concepts**

6 UKoLoS

amplitude modulation is feasible for a long time in contrast to complex phase modulation

For the synthesis of communication signals [4], any phase modulated constant envelope signal that is bandwidth limited to half pulse repetition frequency can be chosen. The maximum

Demodulation can be achieved similar to existing approaches that allow quadrature pulse demodulation (e.g. [11]). Basically, the phase of each pulse has to be sampled synchronously to the pulse sequence (i.e. during pulse duration), which can be realized e.g. by quadrature baseband down-conversion and synchronized sample acquisition. In this case, the sequence

> · *e*

where Δ*tsync* denotes a modestly (uncertainty less than half pulse width) unknown synchronization error that has to be taken into account in practice. Inserting (4) in (6) leads to:

Accordingly, the original phase modulation *ϕinj* is reconstructed correctly aside from a constant phase offset. Its constancy is guaranteed as long as the natural frequency of the unstabilized oscillator does not drift too fast, which is mostly given due to relatively slow changes in environmental parameters like temperature. For compensation, e.g. differential

Since the SILO based synthesizer is capable of generating any constant envelope phase modulated signals (within the bandwidth limit), even a frequency modulated radar signal

can be transmitted. At the receiver, the time delayed transmit signal *s*(*t*) is mixed with a

According to [3], the approximate resulting beat frequency spectrum (disregarding envelope)

is equivalent to the conventional FMCW spectrum except for the aliases resulting from switched operation and a constant phase offset *A*. The (one way) distance can be calculated

*B T td* + *Td* 2

*ω* + 2*π*

*ϕinj*,*FM*(*t*) = 2*π*

*srecv*,*FM*(*t*) = *s*(*t* − *td*) · *e*

*fb* <sup>=</sup> *<sup>B</sup> T td* + *Td* 2 

+(*ωosc*−*ωinj*)·

*B* 2*T t*

<sup>−</sup>*j*(*ωinjt*+*<sup>π</sup> <sup>B</sup>*

*T t* 2

∗ X <sup>1</sup> *Ts*

 <sup>Δ</sup>*tsync*+ *Td* 2 

<sup>−</sup>*jωinj*(*k*·*Ts*+Δ*tsync* ), *<sup>k</sup>* <sup>∈</sup> **<sup>N</sup>**, (6)

<sup>2</sup> (8)

). (9)

(*ω*) (10)

(11)

. (7)

and can be added independently, this work is concentrated on the latter aspect.

**3.4. Phase modulated UWB communication signals**

possible symbol rate leads to one symbol per pulse.

*srecv*(*k*) = *s*

*srecv*(*k*) = *Ar* · *e*

modulation schemes or short frames can be applied.

**3.5. Frequency modulated UWB radar signals**

with the bandwidth *B*, sweep duration *T* and phase

*Srecv*,*FM*(*ω*) = *A* · *δ*

FMCW signal:

from

*k* · *Ts* + Δ*tsync*

*j ϕinj <sup>i</sup>*·*Ts*<sup>−</sup> *Td* 2 

of received samples is given by

In the following, concepts and implementations for the pulsed angle modulated signal synthesis principle are presented. Firstly, the harmonic sampling approach is presented,

#### 8 UKoLoS 350 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems <sup>9</sup>

which is used to take advantage of all benefits of the switched injection-locked oscillator concept by generating a high power, high frequency signal efficiently from a low frequency intermediate signal (4.1). Secondly, a frequency modulated direct digital synthesis (DDS) based upconversion approach for radar applications from the preceding project (PFM-USR) is presented as starting point for the subsequent development (4.2). Thirdly, the recent hardware concept and implementation for phase stepped modulation is described, which allows for synthesizing both frequency modulated radar signals and phase modulated communication signals with the same simple communication signal generator hardware for integrated communication and ranging.

Regarding maximum baseband modulation bandwidth, there exists a limit for the frequency multiplication factor *n* in order to guarantee spectral separation, since the bandwidth increases with the harmonic order whereas the spacing of the harmonics' center frequencies is equidistant. According to [2] (see also Fig. 5 right), the upper boundary for the multiplication

Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems 351

*<sup>n</sup>* <sup>&</sup>lt; *fc*

*<sup>B</sup>* <sup>−</sup> <sup>1</sup> 2

The "classic" approach towards synthesizing linear frequency modulated signals (see Fig. 6) consists of a DDS generating a low frequency reference chirp, a PLL and VCO loop and a linear power amplifier. By adding a pulsed switch at the output, pulsed frequency modulation can be realized similar to section 4.2 as long as the pulse width is short enough (the latter signal has constant phase during the pulse, the first one features slight frequency modulation). Obviously, this classic approach has several disadvantages at high frequencies, especially power consuming linear amplifiers and a switch that dissipates more than 90% of the RF

. (15)

factor is (harmonic center frequency *fc*, harmonic modulation bandwidth *B*):

**4.2. Frequency modulated baseband upconversion**

power at common pulse sequence duty cycles of less than 1:10.

 


" ", , - .""



 

 0 1

 **- -**

 

 

 

**Figure 7.** Harmonic sampling concept for FMCW baseband upconversion

 **-**

.#" ." ." .-"

**- -**

 /




 

**Figure 6.** Comparison of classic and SILO based pulsed frequency modulated signal synthesis [2]

 /

." .-" "  


\$

 0 1 - <sup>2</sup> , ! # <sup>3</sup> " ."

 

\$ 

 

Therefore, a harmonic sampling approach was proposed to directly synthesize the ramp from a DDS signal while avoiding PLLs and linear amplifiers at high frequencies [2]. Due to the

 

 

!"#"  -%&'

 *- -*

)) 

( (

\*+ 

### **4.1. Harmonic sampling approach**

**Figure 5.** SILO based harmonic sampling; left: concept, right: spectrum of bandwidth limited signal after harmonic generator (here: FMCW sweep from *f*<sup>0</sup> to *f*1) [2]

When synthesizing a high frequency pulsed angle modulated signal, classic approaches based on VCO, PLL, linear amplifier and pulsed switch are not suitable to meet goals like low complexity and low power hardware. Instead, a baseband modulator is proposed for signal generation that generates much lower frequencies than at the system's RF output, e.g. 5.8 GHz instead of 63.8 GHz. At lower frequency ranges, analog RF circuits are usually more efficient than their high frequency counterparts. The baseband signal is then applied to the input of a passive or low power non-linear element that generates harmonics, e.g. a diode or transistor (see Fig. 5). Finally, a SILO is used to amplify the upconverted signal by typically more than 50 dB (within pulse duration). Considering an instantaneous output power level of 0 to 5 dBm, an injection level of less than −45 dBm is sufficient, which allows for high losses and low power consumption in the preceding frequency multiplier stage.

In order to avoid strong intermodulation products caused by the baseband modulation, it should be "slow" compared to the center frequency of the baseband signal so that the non-linear element's instantaneous input and filtered output signal can be considered approximately single tone. This requirement is needed for the SILO, which can itself only correctly regenerate constant envelope signals (apart from the fact that intermodulation products are undesirable) that are stable during the startup phase of the oscillator, e.g. FMCW signals with low ramp slope or rectangular shaped PSK with symbol rate / pulse repetition frequency much smaller than RF frequency.

Regarding maximum baseband modulation bandwidth, there exists a limit for the frequency multiplication factor *n* in order to guarantee spectral separation, since the bandwidth increases with the harmonic order whereas the spacing of the harmonics' center frequencies is equidistant. According to [2] (see also Fig. 5 right), the upper boundary for the multiplication factor is (harmonic center frequency *fc*, harmonic modulation bandwidth *B*):

$$m < \frac{f\_c}{B} - \frac{1}{2}.\tag{15}$$

### **4.2. Frequency modulated baseband upconversion**

8 UKoLoS

which is used to take advantage of all benefits of the switched injection-locked oscillator concept by generating a high power, high frequency signal efficiently from a low frequency intermediate signal (4.1). Secondly, a frequency modulated direct digital synthesis (DDS) based upconversion approach for radar applications from the preceding project (PFM-USR) is presented as starting point for the subsequent development (4.2). Thirdly, the recent hardware concept and implementation for phase stepped modulation is described, which allows for synthesizing both frequency modulated radar signals and phase modulated communication signals with the same simple communication signal generator hardware for integrated

communication and ranging.


 

 

**4.1. Harmonic sampling approach**



**Figure 5.** SILO based harmonic sampling; left: concept, right: spectrum of bandwidth limited signal

When synthesizing a high frequency pulsed angle modulated signal, classic approaches based on VCO, PLL, linear amplifier and pulsed switch are not suitable to meet goals like low complexity and low power hardware. Instead, a baseband modulator is proposed for signal generation that generates much lower frequencies than at the system's RF output, e.g. 5.8 GHz instead of 63.8 GHz. At lower frequency ranges, analog RF circuits are usually more efficient than their high frequency counterparts. The baseband signal is then applied to the input of a passive or low power non-linear element that generates harmonics, e.g. a diode or transistor (see Fig. 5). Finally, a SILO is used to amplify the upconverted signal by typically more than 50 dB (within pulse duration). Considering an instantaneous output power level of 0 to 5 dBm, an injection level of less than −45 dBm is sufficient, which allows for high losses and low

In order to avoid strong intermodulation products caused by the baseband modulation, it should be "slow" compared to the center frequency of the baseband signal so that the non-linear element's instantaneous input and filtered output signal can be considered approximately single tone. This requirement is needed for the SILO, which can itself only correctly regenerate constant envelope signals (apart from the fact that intermodulation products are undesirable) that are stable during the startup phase of the oscillator, e.g. FMCW signals with low ramp slope or rectangular shaped PSK with symbol rate / pulse repetition

 

**-**

**-**

**-**

 

**-**

**-**

**-**

**-**

 - 

 

power consumption in the preceding frequency multiplier stage.

frequency much smaller than RF frequency.

after harmonic generator (here: FMCW sweep from *f*<sup>0</sup> to *f*1) [2]

The "classic" approach towards synthesizing linear frequency modulated signals (see Fig. 6) consists of a DDS generating a low frequency reference chirp, a PLL and VCO loop and a linear power amplifier. By adding a pulsed switch at the output, pulsed frequency modulation can be realized similar to section 4.2 as long as the pulse width is short enough (the latter signal has constant phase during the pulse, the first one features slight frequency modulation). Obviously, this classic approach has several disadvantages at high frequencies, especially power consuming linear amplifiers and a switch that dissipates more than 90% of the RF power at common pulse sequence duty cycles of less than 1:10.

**Figure 6.** Comparison of classic and SILO based pulsed frequency modulated signal synthesis [2]

**Figure 7.** Harmonic sampling concept for FMCW baseband upconversion

Therefore, a harmonic sampling approach was proposed to directly synthesize the ramp from a DDS signal while avoiding PLLs and linear amplifiers at high frequencies [2]. Due to the

#### 10 UKoLoS 352 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems <sup>11</sup>

bandwidth restrictions with harmonic sampling (see section 4.1), a single non-linear stage is not sufficient to generate a 7-8 GHz ramp with a commercially available 1 GS/s DDS circuit. Hence, a Nyquist image from the DDS is used to shift the baseband output frequency range to 1.4-1.6 GHz (see Fig. 7).

**5. SILO concept and implementation**

*var*,*min C*<sup>∗</sup>

**Table 1.** VCO component parameters

suitable for implementation of the SILO.

an integrated circuit.

shown in Fig. 9.

output switch. Parameter *C*∗

switch

is given in the following.

Consider the signal displayed in Fig. 1 and the basic SILO model depicted in Fig. 2. As a pulse width *Td* of 1 ns and shorter was to be accomplished, the large parasitic capacitances associated with discrete components made it clear that only an integrated solution would be

Concepts and Components for Pulsed Angle Modulated Ultra Wideband Communication and Radar Systems 353

As a benchmark for the novel circuit concept of the SILO, some key components of a more conservative concept of generating pulsed frequency modulated signals were developed in

All integrated circuits were designed in Cadence Virtuoso and simulated using the Cadence Virtuoso Spectre Circuit Simulator (Cadence, Spectre and Virtuoso are registered trademarks of Cadence Design Systems, Inc). The transmission lines and passive baluns used in the

To evaluate the efficiency of the SILO approach, a conventional circuit using a VCO with wide tuning range and an output switch was designed. The system with the manufactured IC is

The schematic of the VCO can be seen in Fig. 10, together with the half-circuit of the designed

Value 65 fF 200 fF 145 fF 455 fF 0.41 nH 200 Ω 200 Ω Parameter *C*<sup>1</sup> *C*<sup>2</sup> *C*<sup>3</sup> *C*<sup>4</sup> *VBias VCC Vtune* Value 700 fF 200 fF 300 fF 300 fF 1.8 V 3.3 V 0 to 4 V

**Figure 9.** Pulsed frequency modulated continuous wave synthesizer system concept using an output

The VCO is based on a common collector Colpitts oscillator design, including a second varactor diode pair at the transistor base. It is described in detail in [8]. A short overview

A bipolar current mirror is used to drive the oscillator core. The emitter follower output buffer from [8] was replaced by a differential pair to increase common-mode rejection. The

<sup>√</sup>*LbCin*

(16)

*fres* <sup>=</sup> <sup>1</sup> 2*π*

VCO frequency defining series resonant circuit consists of *LB* and *Cin*:

*var*,*max Cvar*,*min Cvar*,*max LB RE RCC*

63.8 GHz-IC were simulated in the Sonnet Professional 2.5D field simulator.

**5.1. The benchmark circuit: VCO with integrated switch**

The main advantage of this concept is that the generated pulsed frequency modulated signal features a very good linearity in comparison to simple PLL control loops and that the only active component at output frequency is a simple, efficient oscillator (SILO). Despite the simplicity of this concept, its hardware design is quite challenging, since the amplitude of a wideband sweep is subject to many inherent sources of frequency dependent amplitude behavior like DDS spectral envelope, insufficient filter flatness and the non-linear element, which increases existing amplitude variations notably.
