**2.3. UWB pseudo-Noise Concept**

378 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

will be gambled away if sounding signals of large crest factors are applied.

*2.2.2. Sensors of narrow instantaneous bandwidth* 

DUT

Stimulus DUT reaction

*Y ft* cos 2

 <sup>0</sup> 

**Figure 3.** Principle of sine wave measurement.

<sup>0</sup> *X ft* cos 2

response function of the DUT:

estimation.

considered.

*Analog correlator*: Due to the lag of programmable analog wideband delay lines, one applies two wideband sources (pulse or PN-sequence) providing two identical signals which are shifted in time. The time shift may be controlled by the same approaches as mentioned above. One of these signals stimulates the DUT, and the other one acts as reference in a correlator. Even if the mixer and the integrator do not waste signal energy, the correlator has about the same efficiency as a sequential sampling receiver as long as one does not deal with parallel correlation stages. We can find from eq. (7) that the correlation principle will provide the best dynamic range due to the large time-bandwidth product. But this benefit

*Sub-sampling correlator*: Here, we can use also random noise as stimulus. The time lag between measurement and reference signal is performed by shifting the sampling time as explained before. The correlation is done in the numerical domain. The approach is quite time consuming since the averaging time must be high in order to achieve a stable

Strictly spoken, such sensors do not belong to UWB systems but they are doing the same job as real UWB devices if they are applied for sensing. Hence, they are worth being

> homodyne or heterodyne vector

Fig. 3 depicts the very basic principle. The signal source is a sine wave generator which steps or sweeps the signal frequency over the wanted bandwidth. Depending on the requirements (signal purity, frequency stability, frequency axis linearity, settling time etc.), free-running VCO's, synthesizers or DDS-circuits are in use. The receivers are based on homodyne or heterodyne down-conversion providing the complex valued frequency

*IFFT I f jQ f G f g t R f*

(10)

*I f XY* <sup>0</sup> 0.5 cos

*Q f XY* <sup>0</sup> 0.5 sin

<sup>0</sup> *R f X* 0.5

receiver <sup>2</sup>

Under the assumption of Pseudo-Noise (PN)-codes for sounding, Nyquist sampling for data capture and embedded pre-processing for data reduction, the principle depicted on the top of Fig. 2 seems to be the most promising if one trades the pros and cons of the various UWB principles with respect to monolithic integration, system performance, MIMO-capability and power consumption. Fig. 4 represents the modified structure adapted to the conditions mentioned above. The use of two receiver channels yields the best performance with respect to different application aspects like synchronous measurement of stimulus and reaction signal, opportunity of device calibration, difference or interferometric measurements as well as long term sensor stability.

**Figure 4.** Basic structure of UWB PN-device.

A stable microwave oscillator controls the whole system. It has to provide only a single frequency *<sup>c</sup> f* which allows the use of simple and stable generator concepts. The oscillator pushes a high-speed shift register. Depending on its feedback structure, it provides any binary sequence. Preferentially, M-sequences are used due to their favorable autocorrelation function. Other options could be Golay-codes [8] or Gold-codes if crosscorrelation properties are in the foreground of interest.

### 380 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

For the sake of brevity, we restrict ourselves here solely to M-sequences. These codes have a sinc2-spectrum with the first zero located at *<sup>c</sup> f* and they have concentrated about 80% of their energy in the spectral band from DC to 2 *<sup>c</sup> f* . Hence, we will not provoke a dramatic performance loss if we limit the upper frequency to 2 *<sup>c</sup> f* . On the contrary, the band limitation avoids disproportionate growth of noise against ever decreasing signal power.

HaLoS – Integrated RF-Hardware Components for Ultra-Wideband Localization and Sensing 381

The sensor principle depicted in Fig. 4 is basically also able to deal with short subnanosecond pulses. However, this would greatly degrade the performance of the system which is largely determined by the amount of signal energy accumulated in the receiver. In the case of pulse signals, this requires amplifiers of high compression points and high resolution ADCs since the whole signal energy is concentrated in a short moment. Furthermore, the measurement object may be exposed to strong fields in the case of nearfield measurements. The application of PN-codes avoids all these flaws since it carries enough energy even with small signal magnitudes. As the impulse compression (leading to high crest factor signals) is performed in the digital domain, the analog sensor components

**Figure 5.** Stability of time position measurement. The delay time of a 50 cm long RF cable was measured at constant temperature. The RF clock was provided either by a sophisticated sine wave

It is well known that the impulse compression of time-extended wideband signals largely improves the dynamic range. As shown in [2] (chapter 4.7.3), it also reduces the jitter susceptibility. The impulse compression distributes the jitter power evenly over the whole signal like additive noise. However, a noise increase above the "natural" level of electronic and quantization noise cannot be observed since the jitter-induced perturbations remain quite low due to the measures described above. Hence, the edges of the impulse response of a DUT measured by the PN-principle are not affected by jitter as usually in pulse

<sup>0</sup> <sup>5</sup> <sup>10</sup> <sup>15</sup> <sup>20</sup> <sup>25</sup> <sup>30</sup> <sup>35</sup> <sup>40</sup> <sup>45</sup> <sup>50</sup> -10

Standard deviation: 2.4 fs

0 5 10 15 20 25 30 35 40 45

Standard deviation: 1.4 fs

observation time [min]

observation time [min]

generator (SMP04 from Rhode & Schwarz) (top) or by a free running DRO (bottom).

and test objects are spared from high voltage peaks.


0

5

10


0

5

fluctuation of pulse position [fs]

fluctuation of pulse position [fs]

The band limitation to half the clock rate touches several performance-relevant issues:


The embedded pre-processing is mainly aimed at data reduction by synchronous averaging (often the measurement rate is much higher than required by the time variance of the test scenarios), static background removal or signal transformations. It should, however, be respected that impulse compression (in order to get the impulse response) performed at this point will increase the data throughput toward the main processor since the word length of the data samples increases.

<sup>1</sup> Here, we refer to general measurement conditions. We disregard sparse sampling which is largely dependent on the measurement objects.

The sensor principle depicted in Fig. 4 is basically also able to deal with short subnanosecond pulses. However, this would greatly degrade the performance of the system which is largely determined by the amount of signal energy accumulated in the receiver. In the case of pulse signals, this requires amplifiers of high compression points and high resolution ADCs since the whole signal energy is concentrated in a short moment. Furthermore, the measurement object may be exposed to strong fields in the case of nearfield measurements. The application of PN-codes avoids all these flaws since it carries enough energy even with small signal magnitudes. As the impulse compression (leading to high crest factor signals) is performed in the digital domain, the analog sensor components and test objects are spared from high voltage peaks.

380 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

dramatic performance loss if we limit the upper frequency to 2 *<sup>c</sup>*

a sinc2-spectrum with the first zero located at *<sup>c</sup>*

1. We need a sampling rate of at least *<sup>c</sup>*

2. Both, the signal edges of the microwave clock *<sup>c</sup>*

3. The time axis of the receiver is defined by the sampling clock *<sup>s</sup>*

within one period.

axis representation.

sampling theorem.

the data samples increases.

measurement objects.

efficiency *<sup>r</sup>*

drift.

power.

of their energy in the spectral band from DC to 2 *<sup>c</sup>*

For the sake of brevity, we restrict ourselves here solely to M-sequences. These codes have

band limitation avoids disproportionate growth of noise against ever decreasing signal

words, it is sufficient to capture one sample per chip of the M-sequence. As the Msequence is periodic, we can do this by sub-sampling. It is easy to show that a binary divider is sufficient for timing control since the number *N* of chips in the sequence is always one less than a power of two ( 2 1 *<sup>m</sup> N* ). If the order *m* of the shift register and the order *n* of the binary divider are identical, we have sequential sampling. For *n m* , one speaks about interleaved sampling which takes more than one sample

quite steep. Hence, the trigger events activated by them are robust against jitter and

from a stable RF-generator and a digital frequency divider which has to run trough all its states before it can launch a new impulse. Hence, any internal deviations between the involved flip-flops have no effect on the divided signal. Therefore, apart from the remaining jitter, we can expect exact equidistant sampling i.e. an absolutely linear time

4. The principle of interleaved sampling allows the sampling rate to be varied by keeping the sensor concept. Thus, one can reduce the sampling rate in favor of reduced power consumption and device costs or it can also be increased to improve the receiver

The embedded pre-processing is mainly aimed at data reduction by synchronous averaging (often the measurement rate is much higher than required by the time variance of the test scenarios), static background removal or signal transformations. It should, however, be respected that impulse compression (in order to get the impulse response) performed at this point will increase the data throughput toward the main processor since the word length of

1 Here, we refer to general measurement conditions. We disregard sparse sampling which is largely dependent on the

 depending on the development state of high-speed electronics. 5. Nyquist sampling provides the lowest possible data throughput1 without violation of

The band limitation to half the clock rate touches several performance-relevant issues:

*f* and they have concentrated about 80%

*f* in order to meet the Nyquist theorem. In other

*f* as well as of the sampling clock *<sup>s</sup>*

*f* . Hence, we will not provoke a

*f* . On the contrary, the

*f* are

*f* . This clock originates

**Figure 5.** Stability of time position measurement. The delay time of a 50 cm long RF cable was measured at constant temperature. The RF clock was provided either by a sophisticated sine wave generator (SMP04 from Rhode & Schwarz) (top) or by a free running DRO (bottom).

It is well known that the impulse compression of time-extended wideband signals largely improves the dynamic range. As shown in [2] (chapter 4.7.3), it also reduces the jitter susceptibility. The impulse compression distributes the jitter power evenly over the whole signal like additive noise. However, a noise increase above the "natural" level of electronic and quantization noise cannot be observed since the jitter-induced perturbations remain quite low due to the measures described above. Hence, the edges of the impulse response of a DUT measured by the PN-principle are not affected by jitter as usually in pulse

### 382 Ultra-Wideband Radio Technologies for Communications, Localization and Sensor Applications

measurements. This favors the PN-sensor concept for applications dealing with super resolution techniques or micro-Doppler problems, particularly if weak scattering targets are overwhelmed by strong ones. Chapter11 *ultraMedis* gives some examples of related problems, and Fig. 5 illustrates the achieved short-time stability of an M-sequence sensor having a bandwidth of about 8 GHz. The short-time variance of the pulse position measurement was in the lower fs range corresponding to a distance variation below 1 µm.

HaLoS – Integrated RF-Hardware Components for Ultra-Wideband Localization and Sensing 383

In order to support system design in its individual stages, different amplifier versions have been developed. Classical ultra-wideband low-noise amplifiers (UWB-LNAs) have been implemented first to ensure early availability and to assess the SiGe BiCMOS technology applied. Then, new receiving components have been considered to address the requirements discovered in system design. This way, a new subtraction amplifier has been made available which allows for practical evaluation of the feedback sampling approach. **Fig. 7** illustrates

**Figure 7.** Transmit and receive path of the basic UWB PN-device from Fig. 4 (solid lines) and proposed

In this figure, the basic M-Sequence system is indicated by solid lines. Within this system, classical UWB-LNAs (the LNA component in the dash-dotted box) are used to support the way of operation presented in subsection 2.3. Possible implementations are covered in subsections 3.2.1 to 3.2.3. In addition, a system extension is indicated by the dashed lines, which tries to overcome the limitations imposed by the analog-to-digital converter. For this reason, a digitally calculated prediction signal is provided by a digital-to-analog converter and subtracted from the receive signal to generate a difference signal of highly reduced signal swing close to sampling time instances (see sections 5.1 and 6.2.3). Subtraction is performed by an amplifier with integrated signal subtraction capability, practical

extension to allow for feedback sampling (dashed lines – see subsection 6.2.3). Either the LNA component or the amplifier with integrated signal subtraction capability are present (dash-dotted box)

the way in which LNAs and subtraction amplifiers are used as part of the system.

**3. Analog wideband receiver circuits** 

according to the system state at which it is focused.

realizations of which are presented in sections 3.4.1 and 3.4.2.

**3.1. Introduction** 

The simple timing concept of the PN-sensors enables the implementation of large MIMOarrays at which the number of cascaded measurement units is basically not limited. The principle is shown in Fig. 6. However, the data handling will be increasingly demanding with a rising number of channels. In a typical operation mode, the transmitters are sequentially activated while the receivers of all channels work in parallel. Some details of implemented MIMO-systems can be found in chapter 11 *ultraMedis*.

**Figure 6.** Creation of a MIMO-system by cascading M-sequence sensors.

The receiver of the UWB PN-sensor applies sub-sampling for data capture. Hence, its efficiency gives some potential of further improvements. This would, however, be connected with a considerable increase of the sampling rate *sf* . As we can see from (8), the elevation of the sampling rate has to be done at the expense of the ADC resolution since the FoM-value is primarily fixed by the semi-conductor technology, while the maximum power is limited by the achievable heat transport. However, simply increasing the sampling rate based on low bit ADCs will not bring any profit with respect to the sensor performance, i.e. the opposite will happen.

As, however, the update rate of UWB PN-sensors is much higher than required by the time variance of the test object, the difference between two consecutive measurements is very low so that low resolution ADCs are sufficient for capturing these deviations. Anyway, this supposes a fast control loop and a (less power hungry) DAC of sufficient resolution which provides the captured signals from previous measurements for reference. Some basic considerations related to this type of feedback sampling can be found in [2]. Details of the layout and implementation of related sub-components are discussed in sections 3.4, 5 and 6.2.
