**4. Multiple band-notched characteristics using meander lines (MLs)**

#### **4.1. Design multiple band-notched antennas using MLs**

Meander line (ML), also known as serpentine line, consisting of a number of transmission lines closely packed and jointed to each others, as shown in Figure 24, is an effective way for size reduction of a transmission line [29-36]. The idea behind meandering is to increase the electrical length per unit area of circuit board space when the signal is propagating along the ML.

252 Ultra Wideband – Current Status and Future Trends

These results indicate that the

*3.6.2. Time-domain performance* 

received pulses for both antennas.

achieve the fidelity of more than 94%.

characteristic for the antenna.

about 2.5 dBi to -5.4 dBi and the radiation efficiency is reduced from about 80% to 15%.

**Figure 22.** Simulated and measured (a) peak gains and (b) efficiencies of proposed antenna

**Figure 23.** Measured pulse responses for (a) reference and (b) single-band notched antennas

The fidelities *F* of the time responses using (1) are computed and shown in Table 4. In both the face-to-face and side-by-side arrangements, the fidelities are about the same. As expected, the reference UWB antenna has the fidelity of more than 97%, very close to the source signal. The results in Table 4 show that the antenna with a single-band notch can

The measurement procedure for the time-domain performance is described in the previous section. For comparison, the pulse responses of the reference UWB antenna (without having the notched characteristic) are also measured and shown in Figure 23. It can be seen that, the pulse responses in the face-to-face arrangements have larger amplitudes than those in the side-by-side arrangements. This agrees with the radiation patterns shown in Figure 21 where radiations in the x-direction are slightly larger than those in the y-direction in most of the frequencies tested. The pulse responses for the reference antenna are only slightly larger than those for the notched antenna in the same arrangements. Late time ringing (or distortion) and lower power are observed in the

/4-resonators effectively generate a single band-notched

λ

To design multiple band-notched characteristics for compact UWB antennas, the λ/4 resonator used in section 3 is too large to be used. By folding the λ/4-resonator into a ML, we can obtain a compact structure. Due to the mutual coupling between the adjacent segments of the ML, the total physical resonator length is no longer λ/4 long. Studies have shown that we can make the electrical length of the MLs to be λ/4 with a smaller size. With the compact structures of MLs, we can easily place several pairs of MLs in different positions of the antenna to obtain a multiple band-notched characteristic. In our proposed design, two different types of feeding techniques, known here as parallel-coupled feeding (PCF) and direct-connected feeding (DCF), as shown in Figures 25(a) and 25(b), respectively, are employed. In the PCF ML, the signal is coupled from the transmission line to the ML, while in the DCF ML, the signal is fed directly to the ML. Details of the different band-notched designs for a compact UWB antenna using MLs are described in the following sections.

**Figure 25.** MLs with (a) parallel-coupled fed and (b) direct-connected fed

**Figure 24.** Typical ML with 8 segments

The design procedure for our band-notched antennas is as follows. An UWB antenna without any ML is designed and used as a reference antenna for comparison. Several pairs of MLs are then added to the reference antenna to make it a multiple band-notched antenna. (A single notched-band antenna requires only one pair of MLs.) The dimensions of the individual MLs are adjusted to achieve the desirable center frequencies and bandwidths for the notches. For convenience, we use the same antenna in section 2 as the reference antenna for our band-notched antennas design here.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 255

In the single and dual band-notched designs shown in Figures 26(a) and 26(b), respectively [10, 11, 37], the notches are designed at the frequencies of existing communication systems such as the WiMax system, upper and lower bands of the WLAN systems. In the triple band-notched design of Figure 26(c), the notches are designed at 3.5, 5.25 and 5.775 GHz. In the quadruple band-notched antenna design of Figure 26(d), we select four notch frequencies with around 1 GHz separation to show the flexibility in designing the notch frequencies. To avoid duplicated description, the quadruple band-notched antenna is used here to explain creating notches using MLs. In Figure 26(d), the four pairs of MLs, labeled as MLs #1, #2, #3 and #4, are designed to create the notches at 3.14, 4.34, 5.4 and 6.4 GHz, respectively. Detailed dimensions of the MLs for single, dual, triple and quadruple band-

*ML d1 d2 d3 d4 #1 0.3 0.3 0.4 1.9* 

*ML d5 d6 d7 d8 w3 #1 0.25 0.25 0.25 2.9 5 #2 0.25 0.25 0.25 2.6 8.5* 

*ML d1 d2 d3 d4 ML d5 d6 d7 d8 w3 #1 0.3 0.3 0.25 2.2 #3 0.3 0.3 0.3 2.7 5.2* 

*ML d1 d2 d3 d4 ML d5 d6 d7 d8 w3 #1 0.3 0.3 0.25 3 #3 0.2 0.2 0.2 2.5 5.2 #2 0.3 0.3 0.25 2.1 #4 0.2 0.2 0.2 2.1 8.5* 

The PCF MLs, i.e., MLs #1 and #2 in Figure 26(d), on both sides of the feed line have open circuit at one end and short circuit to the ground through a via at the other end. With an

feed line is coupled to the MLs and then flowing through the vias to the ground plane. This creates high impedance for the signal and prevents the signal from flowing into the radiator.

*<sup>g</sup>*/4-resonators with parallel-coupled fed. At resonant frequencies, the signal on the

*<sup>g</sup>* is the guided wavelength approximately given by (2), the

λ

*g*/4-length standing wave

**Table 5.** Dimensions of PCF MLs and DCF MLs for Single Band-Notched Antenna

**Table 6.** Dimensions of PCF MLs and DCF MLs for Dual Band-Notched Antenna

**Table 7.** Dimensions of PCF MLs and DCF MLs for Triple Band-Notched Antenna

**Table 8.** Dimensions of PCF MLs and DCF MLs for Quadruple Band-Notched Antenna

**4.2. Parallel-coupled fed (PCF) and direct-connected fed (DCF) MLs** 

notched antennas are listed in Tables 5 to 8, respectively.

*#2 0.3 0.3 0.3 2* 

λ

*g*/4, where

λ

*g*/4-resonator operating in the fundamental mode will have a

electrical length of

λ

MLs are

A λ

As previously described and shown in Figure 14, the surface current concentrates most at the edges of the feed line, the ground plane and the radiator. So we can place the resonators implemented using MLs along these edges as shown in Figures 26(a), 26(b), 26(c) and 26(d) to create a single-, dual-, triple- and quadruple-band notches, respectively, for the antenna. Two different types of feeding techniques, i.e. PCF and DCF, are used. The MLs with PCF are etched on the same side as the radiator on the substrate and connected to ground through vias, while the MLs with DCF are etched on the ground plane on the other side of the substrate.

**Figure 26.** Top views and side views of (a) single, (b) dual, (c) triple, (d) quadruple band-notched antennas, (e) PCF and (f) DCF MLs

In the single and dual band-notched designs shown in Figures 26(a) and 26(b), respectively [10, 11, 37], the notches are designed at the frequencies of existing communication systems such as the WiMax system, upper and lower bands of the WLAN systems. In the triple band-notched design of Figure 26(c), the notches are designed at 3.5, 5.25 and 5.775 GHz. In the quadruple band-notched antenna design of Figure 26(d), we select four notch frequencies with around 1 GHz separation to show the flexibility in designing the notch frequencies. To avoid duplicated description, the quadruple band-notched antenna is used here to explain creating notches using MLs. In Figure 26(d), the four pairs of MLs, labeled as MLs #1, #2, #3 and #4, are designed to create the notches at 3.14, 4.34, 5.4 and 6.4 GHz, respectively. Detailed dimensions of the MLs for single, dual, triple and quadruple bandnotched antennas are listed in Tables 5 to 8, respectively.



254 Ultra Wideband – Current Status and Future Trends

for our band-notched antennas design here.

the substrate.

antennas, (e) PCF and (f) DCF MLs

The design procedure for our band-notched antennas is as follows. An UWB antenna without any ML is designed and used as a reference antenna for comparison. Several pairs of MLs are then added to the reference antenna to make it a multiple band-notched antenna. (A single notched-band antenna requires only one pair of MLs.) The dimensions of the individual MLs are adjusted to achieve the desirable center frequencies and bandwidths for the notches. For convenience, we use the same antenna in section 2 as the reference antenna

As previously described and shown in Figure 14, the surface current concentrates most at the edges of the feed line, the ground plane and the radiator. So we can place the resonators implemented using MLs along these edges as shown in Figures 26(a), 26(b), 26(c) and 26(d) to create a single-, dual-, triple- and quadruple-band notches, respectively, for the antenna. Two different types of feeding techniques, i.e. PCF and DCF, are used. The MLs with PCF are etched on the same side as the radiator on the substrate and connected to ground through vias, while the MLs with DCF are etched on the ground plane on the other side of

**Figure 26.** Top views and side views of (a) single, (b) dual, (c) triple, (d) quadruple band-notched


**Table 6.** Dimensions of PCF MLs and DCF MLs for Dual Band-Notched Antenna


**Table 7.** Dimensions of PCF MLs and DCF MLs for Triple Band-Notched Antenna


**Table 8.** Dimensions of PCF MLs and DCF MLs for Quadruple Band-Notched Antenna

#### **4.2. Parallel-coupled fed (PCF) and direct-connected fed (DCF) MLs**

The PCF MLs, i.e., MLs #1 and #2 in Figure 26(d), on both sides of the feed line have open circuit at one end and short circuit to the ground through a via at the other end. With an electrical length of λ*g*/4, where λ*<sup>g</sup>* is the guided wavelength approximately given by (2), the MLs are λ*<sup>g</sup>*/4-resonators with parallel-coupled fed. At resonant frequencies, the signal on the feed line is coupled to the MLs and then flowing through the vias to the ground plane. This creates high impedance for the signal and prevents the signal from flowing into the radiator. A λ*g*/4-resonator operating in the fundamental mode will have a λ*g*/4-length standing wave

formed along it. Computer simulation has been carried out to study the surface current on ML #2 at the resonant frequency of 4.34 GHz and results show that, at any instance, the currents on the whole ML always have the same phase (in the same direction), i.e. a standing wave is formed on the ML which is expected for a λ*<sup>g</sup>*/4-resonator operating in the fundamental mode. This confirmed that MLs #1 & #2 are λ*<sup>g</sup>*/4-resonators. Figure 27(a) shows a snap-shot from the simulation result on the surface current of ML #2. It can be seen that, the current is quite small at the open end of the ML but substantially larger near the via, which is typical for a λ*<sup>g</sup>*/4-resonator.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 257

*<sup>g</sup>*/4=9.27 and 7.825 mm when computed using (2).

*<sup>f</sup> L C* <sup>∝</sup> (3)

*f* proportional to

6.4 GHz computed using Table 8 are 16.2 and 15 mm, respectively, which again are very

These results show that (2) cannot be used in our MLs to determine the notch frequencies for

(2) is derived using the distributed-elements model and can be applied quite accurately to a straight microstrip line. For complicated structures which are many times, say 10, less than the free space wavelength, the lumped-elements model should be used [38]. In our design, the sizes of MLs #1, #2, #3 and #4 are 3×4.5 mm2, 2.1×4.5 mm2, 2.8×2.7 mm2, 2.8×2.3 mm2, respectively. The maximum lengths of these structures are less than 1/10th of the corresponding guided wavelengths of 51.18, 38.82, 29.76 and 25.11 mm at the resonant frequencies of 3.14, 4.34, 5.4 and 6.4 GHz, respectively, thus both the distributed-elements model and the lumped-elements model together should be used [38-40]. In the lumpedelement model, there will be capacitance formed by the shunt capacitance between the ML and the substrate, series capacitance formed between segments of the ML, inductance formed by the series inductance along the ML plus shunt inductance formed between the segments of the ML, and inductance formed by the via [41], so the models are extremely complicated [35, 42-44]. These models will have different resonant modes of operation. In the fundamental mode, only the largest capacitances and inductances are in effect and form

1

where *Leq* and *Ceq* are the equivalent inductance and capacitance, respectively, resulted from

For the PCF MLs, at resonance, the signals will be electrically coupled from the feed line to MLs #1 and #2 via a mutual coupling capacitor *Cm* between them [38]. This coupling will detune the resonant frequency as will be shown later. For the DCF MLs, the signals will be directly fed to MLs #3 and #4. Since it is not easy, if not impossible at all, to relate *Leq* and *Ceq*

Parametric study of the quadruple band-notched antenna is carried out using computer simulation to explore how the dimensions of the MLs affect the characteristics of the band

For the PCF resonators, ML #1 is used for studies. The smallest dimension we can make for our antenna using the prototype-machine in our laboratory is 0.1 mm. So for convenience in our design process, we fix the segment width *d1* and segment spacing *d2* of the ML to be the same value. Simulation results show that, with *d1* = *d2* < 0.2 mm, the notch bandwidth is too

*eq eq*

λ

an *LC* resonator which will have the fundamental resonant frequency *<sup>r</sup>*

*r*

to the dimensions of the MLs, full-wave simulation needs to be used for the design.

**4.4. Setting the notch center frequencies and bandwidths** 

different from the corresponding lengths

1/ *eq eq L C* , i.e.,

notches.

the antennas. The reason can be explained as follows.

the self inductance and self capacitance of the ML.

The DCF MLs, i.e., MLs #3 and #4 in Figure 26(d), at the upper edges of the ground plane are stubs having open circuit at one end and directly connect to ground at the other end. With an electrical length of λ*g*/4, the MLs are λ*<sup>g</sup>*/4-resonators with direct-connected fed. At resonance, the MLs prevent the signal from passing through, creating high input impedance. This causes severely mismatching to the antenna and reduces the return loss. As mentioned before, a λ*g*/4-resonator operating in the fundamental mode has a λ*<sup>g</sup>*/4-length standing wave formed along it. Computer simulation has also been carried out on the surface current of ML #3 at the resonant frequency of 5.4 GHz. Results show that, at any instance, the currents along the ML all have the same phase (in the same direction), as expected for a λ*<sup>g</sup>*/4-resonator operating in the fundamental mode. This again confirms that MLs #3 & #4 are λ*<sup>g</sup>*/4-resonators. Figure 27(b) shows a snap shot from the simulation result on the surface current of ML #3. The current is substantially smaller at the open end of the ML than that at the connected end, which is expected for a λ*<sup>g</sup>*/4-resonator.

**Figure 27.** Current distribution on (a) ML #2 at 4.34 GHz and (b) ML #3 at 5.4 GHz

#### **4.3. Discussions on electrical lengths of PCF and DCF MLs**

For a λ*g*/4-resonator implemented using a straight microstrip line, if ε*<sup>r</sup>* in (2) is known, the physical length of the microstrip line, for the λ*<sup>g</sup>*/4-resonator to work at a particular frequency, can easily be calculated. Unfortunately, this does not work with the MLs in our design. For example, the total lengths measured along the internal perimeters of MLs #1 and #2 in Figure 26(d) for the resonant frequencies of 3.14 and 4.34 GHz are 26.1 and 17.7 mm, respectively, which are quite different from the computed λ*<sup>g</sup>*/4=15.95 and 11.54 mm using (2). Then for MLs #3 and #4, the internal perimeters for the resonant frequencies of 5.4 and 6.4 GHz computed using Table 8 are 16.2 and 15 mm, respectively, which again are very different from the corresponding lengths λ*<sup>g</sup>*/4=9.27 and 7.825 mm when computed using (2). These results show that (2) cannot be used in our MLs to determine the notch frequencies for the antennas. The reason can be explained as follows.

256 Ultra Wideband – Current Status and Future Trends

which is typical for a

mentioned before, a

expected for a

For a λ

MLs #3 & #4 are

With an electrical length of

λ

λ

standing wave is formed on the ML which is expected for a

fundamental mode. This confirmed that MLs #1 & #2 are

*<sup>g</sup>*/4-resonator.

λ

ML than that at the connected end, which is expected for a

**Figure 27.** Current distribution on (a) ML #2 at 4.34 GHz and (b) ML #3 at 5.4 GHz

*g*/4-resonator implemented using a straight microstrip line, if

**4.3. Discussions on electrical lengths of PCF and DCF MLs** 

physical length of the microstrip line, for the

respectively, which are quite different from the computed

*g*/4, the MLs are

λ

λ

formed along it. Computer simulation has been carried out to study the surface current on ML #2 at the resonant frequency of 4.34 GHz and results show that, at any instance, the currents on the whole ML always have the same phase (in the same direction), i.e. a

a snap-shot from the simulation result on the surface current of ML #2. It can be seen that, the current is quite small at the open end of the ML but substantially larger near the via,

The DCF MLs, i.e., MLs #3 and #4 in Figure 26(d), at the upper edges of the ground plane are stubs having open circuit at one end and directly connect to ground at the other end.

resonance, the MLs prevent the signal from passing through, creating high input impedance. This causes severely mismatching to the antenna and reduces the return loss. As

standing wave formed along it. Computer simulation has also been carried out on the surface current of ML #3 at the resonant frequency of 5.4 GHz. Results show that, at any instance, the currents along the ML all have the same phase (in the same direction), as

on the surface current of ML #3. The current is substantially smaller at the open end of the

λ

*g*/4-resonator operating in the fundamental mode has a

*<sup>g</sup>*/4-resonator operating in the fundamental mode. This again confirms that

λ

λ

frequency, can easily be calculated. Unfortunately, this does not work with the MLs in our design. For example, the total lengths measured along the internal perimeters of MLs #1 and #2 in Figure 26(d) for the resonant frequencies of 3.14 and 4.34 GHz are 26.1 and 17.7 mm,

(2). Then for MLs #3 and #4, the internal perimeters for the resonant frequencies of 5.4 and

*<sup>g</sup>*/4-resonators. Figure 27(b) shows a snap shot from the simulation result

λ

*<sup>g</sup>*/4-resonator.

ε

*<sup>g</sup>*/4-resonator to work at a particular

*<sup>g</sup>*/4=15.95 and 11.54 mm using

*<sup>r</sup>* in (2) is known, the

λ

λ

*<sup>g</sup>*/4-resonator operating in the

λ

*<sup>g</sup>*/4-length

*<sup>g</sup>*/4-resonators. Figure 27(a) shows

*<sup>g</sup>*/4-resonators with direct-connected fed. At

(2) is derived using the distributed-elements model and can be applied quite accurately to a straight microstrip line. For complicated structures which are many times, say 10, less than the free space wavelength, the lumped-elements model should be used [38]. In our design, the sizes of MLs #1, #2, #3 and #4 are 3×4.5 mm2, 2.1×4.5 mm2, 2.8×2.7 mm2, 2.8×2.3 mm2, respectively. The maximum lengths of these structures are less than 1/10th of the corresponding guided wavelengths of 51.18, 38.82, 29.76 and 25.11 mm at the resonant frequencies of 3.14, 4.34, 5.4 and 6.4 GHz, respectively, thus both the distributed-elements model and the lumped-elements model together should be used [38-40]. In the lumpedelement model, there will be capacitance formed by the shunt capacitance between the ML and the substrate, series capacitance formed between segments of the ML, inductance formed by the series inductance along the ML plus shunt inductance formed between the segments of the ML, and inductance formed by the via [41], so the models are extremely complicated [35, 42-44]. These models will have different resonant modes of operation. In the fundamental mode, only the largest capacitances and inductances are in effect and form an *LC* resonator which will have the fundamental resonant frequency *<sup>r</sup> f* proportional to 1/ *eq eq L C* , i.e.,

$$f\_r \approx \frac{1}{\sqrt{L\_{eq}C\_{eq}}}\tag{3}$$

where *Leq* and *Ceq* are the equivalent inductance and capacitance, respectively, resulted from the self inductance and self capacitance of the ML.

For the PCF MLs, at resonance, the signals will be electrically coupled from the feed line to MLs #1 and #2 via a mutual coupling capacitor *Cm* between them [38]. This coupling will detune the resonant frequency as will be shown later. For the DCF MLs, the signals will be directly fed to MLs #3 and #4. Since it is not easy, if not impossible at all, to relate *Leq* and *Ceq* to the dimensions of the MLs, full-wave simulation needs to be used for the design.

#### **4.4. Setting the notch center frequencies and bandwidths**

Parametric study of the quadruple band-notched antenna is carried out using computer simulation to explore how the dimensions of the MLs affect the characteristics of the band notches.

For the PCF resonators, ML #1 is used for studies. The smallest dimension we can make for our antenna using the prototype-machine in our laboratory is 0.1 mm. So for convenience in our design process, we fix the segment width *d1* and segment spacing *d2* of the ML to be the same value. Simulation results show that, with *d1* = *d2* < 0.2 mm, the notch bandwidth is too large (more than 1 GHz). Thus for having a low percentage of tolerance and compact size, we fix *d1* and *d2* to a slightly larger value of 0.3 mm. The segment length *d4* and distance *d3* between the ML and the feed line are then used for parametric studies using computer simulation. Results show that *d4* and *d3* affect the notch frequency and bandwidth, respectively. The effect of *d4* on the notch frequency is shown in Figure 28(a), indicating that *d4* is inversely proportional to the notch frequency. This is because increasing the segment length *d4* increases the series inductance and capacitance along the ML and hence lowering down the resonant frequency through (3). With *d4* increased from 2.8 to 3.2 mm, the resonant frequency shifts from 2.85 to 3.23 GHz, which is quite significant. ML #1 is parallelcoupled fed via the mutual capacitance between the feed line and the ML. At the resonant frequency, the signal travelling along the feed line is coupled to the ML. When *d3* is increased, the coupling effect (or the mutual capacitance) between them is reduced, leading to a lower reactance and hence a lower quality factor *Q*. For a given resonant frequency *fr*, the bandwidth is given by *fr/Q*. Thus *d3* is inversely proportional to the notch bandwidth and so can be used to adjust the notch bandwidth. The effect of *d3* on the notch bandwidth is shown in Figure 28(b). With *d3* increased from 0.1 to 0.5 mm, the 10-dB bandwidth changes from 0.31 to 0.11 GHz. It should be noted that the notch frequency is dominated by the ML's equivalent self-capacitance *Ceq* and inductance *Leq* given by (3) and so *d3* will only slightly affect the notch frequency, as can be seen in Figure 28(b).

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 259

close to the feed line, e.g. *w3 =* 2 mm, the notch bandwidth will be too wide and, together with the adjacent notches, form a very wide stopband. Moreover, the small distance *gap* is very critical for impedance matching of the antenna. Placing the ML too close to this gap will affect the matching performance and reduce the impedance bandwidth, as evident in

**Figure 28.** Return loss with different values of (a) *d4* and (b) *d3* for ML #1, and (c) *d8* and (d) *w3* for ML

first spurious response always occurs at three times the resonant frequency [46]. ML #1 is a parallel-coupled fed resonator at around 3 GHz. Figures 28(a) and 28(b) show that it has a spurious response at about 8.5 GHz, slightly less than three times the resonant frequency. In this respect, ML #1 again behaves like a parallel-coupled fed resonator. Note that ML #2 also has the first spurious response at about three times the resonant frequency, which is not

These results also reveal that when the values of *d4*, *d3*, *d8* and *w3* are changed, the return loss in the rest of the UWB band remains about the same. This provides the designers with a

The return loss, efficiency and peak gain across the UWB, and the radiation patterns at the passband and also the notch frequencies of all the four band-notched antennas are studied

great freedom to select the notched-band frequency and bandwidth for the antenna.

*<sup>g</sup>*/4-resonator implemented using a straight microstrip line, the

Figure 28(d).

#3 of the quadruple band-notched antenna

shown in Figure 28(a) due to the small scale.

**4.5. Results and discussions** 

*4.5.1. Frequency-domain performance* 

λ

For a parallel-coupled fed

For the DCF resonators, ML #3 in Figure 28(d) is used for parametric studies. Again, for convenience, we fix the segment width *d6* and segment spacing *d5* of the ML to have the same value. Simulation results show that a smaller value of 0.2 mm for the segment width *d6* and segment spacing *d5* can be used without having a too large notch bandwidth, so we fix *d6* = *d5* = 0.2 mm for having a smaller size. The segment length *d8*, the distance *d7* between the ML and the ground, and the distance *w3* between the center of the feed line and the ML are used for parametric studies. Simulation results show that *d8* affects the notch frequency, for the same reason as described for the PCF resonators. The effect of *d8* on the notch frequency is shown in Figure 28(c). With *d8* increased from 2.3 to 2.7 mm, the notch frequency shifts from 5.71 to 5.04 GHz.

Simulation results show that both *d7* and *w3* in ML #3 affects the notch bandwidth, with *w3* having significantly higher effects, so *w3* is used for studies and results are shown in Figure 26(d). MLs #3 and #4 are λ*<sup>g</sup>*/4-resonators with DCF. There is series inductance on the ground plane between the feed line and the ML. This inductance is connected in series with the ML and increases with the current flowing through it [45]. Since the current density is higher in the region closer to the feed line than far away from it, the inductance is larger with a shorter distance *w3*, leading to a larger reactance and a higher quality factor *Q*. For a given resonant frequency *fr*, the bandwidth given by *fr/Q* is therefore smaller. This is confirmed by the simulation results of Figure 28(d) which shows that, with *w3* increased from 4.6 to 5.8 mm, the 10-dB bandwidth changes from 0.74 to 0.28 GHz. Thus *w3* can be used to adjust the notch bandwidth. Similar to the case for the PCF resonators, the notch frequency is dominated by the self-capacitance and inductance of the ML, so *w3* will only slightly affect the notch frequency as can be seen in Figure 28(d). It should be noted that, if ML #3 is too close to the feed line, e.g. *w3 =* 2 mm, the notch bandwidth will be too wide and, together with the adjacent notches, form a very wide stopband. Moreover, the small distance *gap* is very critical for impedance matching of the antenna. Placing the ML too close to this gap will affect the matching performance and reduce the impedance bandwidth, as evident in Figure 28(d).

**Figure 28.** Return loss with different values of (a) *d4* and (b) *d3* for ML #1, and (c) *d8* and (d) *w3* for ML #3 of the quadruple band-notched antenna

For a parallel-coupled fed λ*<sup>g</sup>*/4-resonator implemented using a straight microstrip line, the first spurious response always occurs at three times the resonant frequency [46]. ML #1 is a parallel-coupled fed resonator at around 3 GHz. Figures 28(a) and 28(b) show that it has a spurious response at about 8.5 GHz, slightly less than three times the resonant frequency. In this respect, ML #1 again behaves like a parallel-coupled fed resonator. Note that ML #2 also has the first spurious response at about three times the resonant frequency, which is not shown in Figure 28(a) due to the small scale.

These results also reveal that when the values of *d4*, *d3*, *d8* and *w3* are changed, the return loss in the rest of the UWB band remains about the same. This provides the designers with a great freedom to select the notched-band frequency and bandwidth for the antenna.

#### **4.5. Results and discussions**

258 Ultra Wideband – Current Status and Future Trends

affect the notch frequency, as can be seen in Figure 28(b).

λ

from 5.71 to 5.04 GHz.

26(d). MLs #3 and #4 are

large (more than 1 GHz). Thus for having a low percentage of tolerance and compact size, we fix *d1* and *d2* to a slightly larger value of 0.3 mm. The segment length *d4* and distance *d3* between the ML and the feed line are then used for parametric studies using computer simulation. Results show that *d4* and *d3* affect the notch frequency and bandwidth, respectively. The effect of *d4* on the notch frequency is shown in Figure 28(a), indicating that *d4* is inversely proportional to the notch frequency. This is because increasing the segment length *d4* increases the series inductance and capacitance along the ML and hence lowering down the resonant frequency through (3). With *d4* increased from 2.8 to 3.2 mm, the resonant frequency shifts from 2.85 to 3.23 GHz, which is quite significant. ML #1 is parallelcoupled fed via the mutual capacitance between the feed line and the ML. At the resonant frequency, the signal travelling along the feed line is coupled to the ML. When *d3* is increased, the coupling effect (or the mutual capacitance) between them is reduced, leading to a lower reactance and hence a lower quality factor *Q*. For a given resonant frequency *fr*, the bandwidth is given by *fr/Q*. Thus *d3* is inversely proportional to the notch bandwidth and so can be used to adjust the notch bandwidth. The effect of *d3* on the notch bandwidth is shown in Figure 28(b). With *d3* increased from 0.1 to 0.5 mm, the 10-dB bandwidth changes from 0.31 to 0.11 GHz. It should be noted that the notch frequency is dominated by the ML's equivalent self-capacitance *Ceq* and inductance *Leq* given by (3) and so *d3* will only slightly

For the DCF resonators, ML #3 in Figure 28(d) is used for parametric studies. Again, for convenience, we fix the segment width *d6* and segment spacing *d5* of the ML to have the same value. Simulation results show that a smaller value of 0.2 mm for the segment width *d6* and segment spacing *d5* can be used without having a too large notch bandwidth, so we fix *d6* = *d5* = 0.2 mm for having a smaller size. The segment length *d8*, the distance *d7* between the ML and the ground, and the distance *w3* between the center of the feed line and the ML are used for parametric studies. Simulation results show that *d8* affects the notch frequency, for the same reason as described for the PCF resonators. The effect of *d8* on the notch frequency is shown in Figure 28(c). With *d8* increased from 2.3 to 2.7 mm, the notch frequency shifts

Simulation results show that both *d7* and *w3* in ML #3 affects the notch bandwidth, with *w3* having significantly higher effects, so *w3* is used for studies and results are shown in Figure

plane between the feed line and the ML. This inductance is connected in series with the ML and increases with the current flowing through it [45]. Since the current density is higher in the region closer to the feed line than far away from it, the inductance is larger with a shorter distance *w3*, leading to a larger reactance and a higher quality factor *Q*. For a given resonant frequency *fr*, the bandwidth given by *fr/Q* is therefore smaller. This is confirmed by the simulation results of Figure 28(d) which shows that, with *w3* increased from 4.6 to 5.8 mm, the 10-dB bandwidth changes from 0.74 to 0.28 GHz. Thus *w3* can be used to adjust the notch bandwidth. Similar to the case for the PCF resonators, the notch frequency is dominated by the self-capacitance and inductance of the ML, so *w3* will only slightly affect the notch frequency as can be seen in Figure 28(d). It should be noted that, if ML #3 is too

*<sup>g</sup>*/4-resonators with DCF. There is series inductance on the ground

#### *4.5.1. Frequency-domain performance*

The return loss, efficiency and peak gain across the UWB, and the radiation patterns at the passband and also the notch frequencies of all the four band-notched antennas are studied

using computer simulation. To validate the simulation results, the antennas are also fabricated on PCBs with PTFE substrate with parameters described previously, as shown in Figure 29, and measured using the antenna measurement system, Satimo Starlab.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 261

The simulated and measured radiation patterns of the single band-notched antenna at the frequencies of 3, 5.5, 7 and 10 GHz in the two principle planes, the x-y and x-z planes, are shown in Figure 31. At 4, 7 and 10 GHz, Figures 31(a), 31(e) and 31(g) show that the antenna has approximately omnidirectional radiation patterns in the x-y plane. In the x-z plane, Figures. 31(b), 31(f) and 31(h) show that there are two nulls at the z-direction, typical for monopole antennas. At the notch frequency of 5.5 GHz, the radiation patterns in Figures 31(c) and 31(d) indicate that the gain is almost evenly suppressed in all directions by the

The simulated and measured peak gains of the single band-notched antenna are shown in Figure 32(a) and the efficiencies are shown in Figure 32(b). The average gain over the UWB, computed by excluding the notched band, is about 3.5 dBi. However, at the notched band of about 5.5 GHz, the antenna gain is suppressed from about 3 dBi to -7.1 dBi and the radiation efficiency is reduced from about 85% to 8.7%. These phenomena indicate that the MLs work

**Figure 31.** Simulated and measured radiation patterns of single band-notched antenna. (a) 3 GHz in xy plane; (b) 3 GHz in x-z plane; (c) 5.5 GHz in x-y plane; (d) 5.5 GHz in x-z plane; (e) 7 GHz in x-y plane;

(f) 7 GHz in x-z plane; (g) 10 GHz in x-y plane; and (h) 10 GHz in x-z plane

effectively to introduce a single band-notched characteristic for the antenna.

pair of MLs and the average gain drops to about -10 dBi.

**Figure 29.** Top and bottom views of prototyped (a) single, (b) dual, (c) triple and (d) quadruple bandnotched antennas

#### *4.5.2. Single band-notched antenna*

The simulated and measured return losses of the single band-notched antenna of Figure 29(a) are shown in Figure 30. It can be seen that, the antenna can operate from 2.76 GHz to over 12 GHz with return loss ≥ 10 dB which fully satisfies the UWB requirement. In the WLAN band from 4.98 to 5.93 GHz, the measured return loss is substantially lower than 10 dB.

**Figure 30.** Simulated and measured return losses of single band-notched antenna

The simulated and measured radiation patterns of the single band-notched antenna at the frequencies of 3, 5.5, 7 and 10 GHz in the two principle planes, the x-y and x-z planes, are shown in Figure 31. At 4, 7 and 10 GHz, Figures 31(a), 31(e) and 31(g) show that the antenna has approximately omnidirectional radiation patterns in the x-y plane. In the x-z plane, Figures. 31(b), 31(f) and 31(h) show that there are two nulls at the z-direction, typical for monopole antennas. At the notch frequency of 5.5 GHz, the radiation patterns in Figures 31(c) and 31(d) indicate that the gain is almost evenly suppressed in all directions by the pair of MLs and the average gain drops to about -10 dBi.

260 Ultra Wideband – Current Status and Future Trends

notched antennas

*4.5.2. Single band-notched antenna* 

using computer simulation. To validate the simulation results, the antennas are also fabricated on PCBs with PTFE substrate with parameters described previously, as shown in

**Figure 29.** Top and bottom views of prototyped (a) single, (b) dual, (c) triple and (d) quadruple band-

The simulated and measured return losses of the single band-notched antenna of Figure 29(a) are shown in Figure 30. It can be seen that, the antenna can operate from 2.76 GHz to over 12 GHz with return loss ≥ 10 dB which fully satisfies the UWB requirement. In the WLAN band

from 4.98 to 5.93 GHz, the measured return loss is substantially lower than 10 dB.

**Figure 30.** Simulated and measured return losses of single band-notched antenna

Figure 29, and measured using the antenna measurement system, Satimo Starlab.

The simulated and measured peak gains of the single band-notched antenna are shown in Figure 32(a) and the efficiencies are shown in Figure 32(b). The average gain over the UWB, computed by excluding the notched band, is about 3.5 dBi. However, at the notched band of about 5.5 GHz, the antenna gain is suppressed from about 3 dBi to -7.1 dBi and the radiation efficiency is reduced from about 85% to 8.7%. These phenomena indicate that the MLs work effectively to introduce a single band-notched characteristic for the antenna.

**Figure 31.** Simulated and measured radiation patterns of single band-notched antenna. (a) 3 GHz in xy plane; (b) 3 GHz in x-z plane; (c) 5.5 GHz in x-y plane; (d) 5.5 GHz in x-z plane; (e) 7 GHz in x-y plane; (f) 7 GHz in x-z plane; (g) 10 GHz in x-y plane; and (h) 10 GHz in x-z plane

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 263

**Figure 34.** Simulated and measured radiation patterns of dual band-notched antenna. (a) 3.5 GHz in xy plane; (b) 3.5 GHz in x-z plane; (c) 5.25 GHz in x-y plane; (d) 5.25 GHz in x-z plane; (e) 5.775 GHz in x-

The measured peak gain and efficiency of the dual band-notched antenna are shown in Figure 35. The average antenna gain over the UWB, computed by excluding the notched bands, is about 3 dBi. While at the notch frequencies, a significant gain and radiation efficiency reductions can be seen. At the notch frequencies of 5.25 and 5.83 GHz, the peak gain drops to -6 dBi and -2 dBi and the radiation efficiency reduces to below 20% and 45%, respectively. These phenomena indicate that by introducing two pairs of MLs, a dual band-

y plane; (f) 5.775 GHz in x-z plane; (g) 9 GHz in x-y plane; and (h) 9 GHz in x-z plane

**Figure 35.** Measured peak gain and efficiency of dual band-notched antenna

notched characteristic can be achieved.

**Figure 32.** Simulated and measured (a) peak gains and (b) efficiencies of single band-notched antenna

#### *4.5.3. Dual band-notched antenna*

The simulated and measured return losses of the dual band-notched antenna of Figure 29(b) are shown in Figure 33. It can be seen that, the antenna can operate from 2.95 GHz to about 11.8 GHz with return loss ≥ 10 dB. In the lower and upper WLAN bands from 5.01 to 5.48 GHz and 5.63 to 5.92 GHz, respectively, the measured return loss for the lower WLAN band is much lower than 10 dB. In the upper WLAN band, the return loss is larger. This is because the MLs are further away from the center feed line and so couple less energy compared to the MLs for the lower WLAN band.

The simulated and measured radiation patterns of the antenna at 3.5 and 9 GHz in the two principle planes, the x-z and x-y planes, are shown in Figure 34. The antenna has approximately omnidirectional radiation patterns in the x-y plane. The x-z plane patterns show two nulls at the z-direction, which are similar to that of a typical monopole antenna.

**Figure 33.** Simulated and measured return losses of dual band-notched antenna

#### Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 263

262 Ultra Wideband – Current Status and Future Trends

*4.5.3. Dual band-notched antenna* 

antenna.

compared to the MLs for the lower WLAN band.

**Figure 32.** Simulated and measured (a) peak gains and (b) efficiencies of single band-notched antenna

The simulated and measured return losses of the dual band-notched antenna of Figure 29(b) are shown in Figure 33. It can be seen that, the antenna can operate from 2.95 GHz to about 11.8 GHz with return loss ≥ 10 dB. In the lower and upper WLAN bands from 5.01 to 5.48 GHz and 5.63 to 5.92 GHz, respectively, the measured return loss for the lower WLAN band is much lower than 10 dB. In the upper WLAN band, the return loss is larger. This is because the MLs are further away from the center feed line and so couple less energy

The simulated and measured radiation patterns of the antenna at 3.5 and 9 GHz in the two principle planes, the x-z and x-y planes, are shown in Figure 34. The antenna has approximately omnidirectional radiation patterns in the x-y plane. The x-z plane patterns show two nulls at the z-direction, which are similar to that of a typical monopole

**Figure 33.** Simulated and measured return losses of dual band-notched antenna

**Figure 34.** Simulated and measured radiation patterns of dual band-notched antenna. (a) 3.5 GHz in xy plane; (b) 3.5 GHz in x-z plane; (c) 5.25 GHz in x-y plane; (d) 5.25 GHz in x-z plane; (e) 5.775 GHz in xy plane; (f) 5.775 GHz in x-z plane; (g) 9 GHz in x-y plane; and (h) 9 GHz in x-z plane

The measured peak gain and efficiency of the dual band-notched antenna are shown in Figure 35. The average antenna gain over the UWB, computed by excluding the notched bands, is about 3 dBi. While at the notch frequencies, a significant gain and radiation efficiency reductions can be seen. At the notch frequencies of 5.25 and 5.83 GHz, the peak gain drops to -6 dBi and -2 dBi and the radiation efficiency reduces to below 20% and 45%, respectively. These phenomena indicate that by introducing two pairs of MLs, a dual bandnotched characteristic can be achieved.

**Figure 35.** Measured peak gain and efficiency of dual band-notched antenna

#### *4.5.4. Triple band-notched antenna*

The simulated and measured return losses of the triple band-notched antenna of Figure 28(c) are shown in Figure 36. The antenna can operate from 2.68 GHz to 11.15 GHz with return loss ≥ 10 dB. In the three desired notched bands, the measured return loss is substantially less than 10 dB.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 265

**Figure 37.** Simulated and measured radiation patterns of triple band-notched antenna. (a) 2 GHz in x-y plane; (b) 2 GHz in x-z plane; (c) 3.45 GHz in x-y plane; (d) 3.45 GHz in x-z plane; (e) 5.25 GHz in x-y plane; (f) 5.25 GHz in x-z plane; (g) 5.775 GHz in x-y plane; (h) 5.775 GHz in x-z plane; (i) 11 GHz in x-y

**Figure 38.** Simulated and measured (a) peak gains and (b) efficiencies of triple band-notched antenna

plane; and (j) 11 GHz in x-z plane

The simulated and measured radiation patterns of the antenna at the frequencies of 2, 3.45, 5.25, 5.775 and 11 GHz in the x-y and x-z planes are shown in Figure 37. At 2 and 11 GHz, Figures 37(a) and 37(i) show that the antenna has approximately omnidirectional radiation patterns in the x-y plane. In the x-z plane patterns, Figures 37(b) and 37(j) show that there are two nulls in the positive and negative z directions. The radiation patterns in Figures 37(c) and 37(d) for 3.45 GHz, in Figures. 37(e) and 37(f) for 5.25 GHz, and in Figures 37(g) and 37(h) for 5.775 GHz indicate that the gain is almost evenly suppressed in all directions in the three notched bands by the MLs and the average gain is about -10 dBi.

The simulated and measured peak gains and efficiencies of the antenna are shown in Figures 38(a) and 38(b), respectively. The measured peak gain is between 2 to 4.75 dBi over the UWB, except in the notched bands. In the three notched bands, i.e., the WiMax, lower WLAN and upper WLAN frequency bands, the peak gain is suppressed to -3.4 dBi, -2.3 dBi and -2.1 dBi, respectively. The radiation efficiency is between 55% - 99%, but substantially reduced to 14.6%, 15.2% and 22.8%, respectively, in these frequency bands. These results indicate that the MLs work effectively to introduce a triple band-notched characteristic for the UWB antenna.

**Figure 36.** Simulated and measured return losses of the triple band-notched antenna

*4.5.4. Triple band-notched antenna* 

substantially less than 10 dB.

the UWB antenna.

The simulated and measured return losses of the triple band-notched antenna of Figure 28(c) are shown in Figure 36. The antenna can operate from 2.68 GHz to 11.15 GHz with return loss ≥ 10 dB. In the three desired notched bands, the measured return loss is

The simulated and measured radiation patterns of the antenna at the frequencies of 2, 3.45, 5.25, 5.775 and 11 GHz in the x-y and x-z planes are shown in Figure 37. At 2 and 11 GHz, Figures 37(a) and 37(i) show that the antenna has approximately omnidirectional radiation patterns in the x-y plane. In the x-z plane patterns, Figures 37(b) and 37(j) show that there are two nulls in the positive and negative z directions. The radiation patterns in Figures 37(c) and 37(d) for 3.45 GHz, in Figures. 37(e) and 37(f) for 5.25 GHz, and in Figures 37(g) and 37(h) for 5.775 GHz indicate that the gain is almost evenly suppressed in all directions

The simulated and measured peak gains and efficiencies of the antenna are shown in Figures 38(a) and 38(b), respectively. The measured peak gain is between 2 to 4.75 dBi over the UWB, except in the notched bands. In the three notched bands, i.e., the WiMax, lower WLAN and upper WLAN frequency bands, the peak gain is suppressed to -3.4 dBi, -2.3 dBi and -2.1 dBi, respectively. The radiation efficiency is between 55% - 99%, but substantially reduced to 14.6%, 15.2% and 22.8%, respectively, in these frequency bands. These results indicate that the MLs work effectively to introduce a triple band-notched characteristic for

in the three notched bands by the MLs and the average gain is about -10 dBi.

**Figure 36.** Simulated and measured return losses of the triple band-notched antenna

**Figure 37.** Simulated and measured radiation patterns of triple band-notched antenna. (a) 2 GHz in x-y plane; (b) 2 GHz in x-z plane; (c) 3.45 GHz in x-y plane; (d) 3.45 GHz in x-z plane; (e) 5.25 GHz in x-y plane; (f) 5.25 GHz in x-z plane; (g) 5.775 GHz in x-y plane; (h) 5.775 GHz in x-z plane; (i) 11 GHz in x-y plane; and (j) 11 GHz in x-z plane

**Figure 38.** Simulated and measured (a) peak gains and (b) efficiencies of triple band-notched antenna

#### *4.5.5. Quadruple band-notched antenna*

The return losses of the reference and quadruple band-notched antennas are shown in Figure 39. The simulation and measurement return losses agree well and are larger than 10 dB across the UWB. In the notched bands, the return loss is substantially smaller than 10 dB. In the rest of the UWB, the return loss is larger than 10 dB, indicating good radiation performance. Figure 39 also shows that ML #1, being a parallel-coupled resonator at 3.14 GHz, has a spurious response at about 8.65 GHz, which is caused by the harmonic responses of the resonator. The measured results in Figure 39 show that the four notches at the frequencies of 3.14, 4.34, 5.4 and 6.4 have the bandwidth of 178, 374, 495 and 862 MHz, respectively.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 267

**Figure 40.** Simulated and measured (a) peak gains and (b) efficiencies of quadruple band-notched

level is quite low, except at the highest frequency of 11 GHz, as can be seen in Figure 41.

notches, but also the bandwidths and locations of the notches.

For comparison, the studies of time-domain performances are carried out as described in previous sections on the single, dual, triple and quadruple band-notched antennas and results are all plotted in Figure 42. The single, dual and triple band-notched antennas have notches at 5.5 GHz, 5.25 & 5.775 GHz, 3.5 & 5.25 & 5.775 GHz, respectively, while the quadruple band-notched antenna has notches at 3.14 & 4.34 & 5.4 & 6.4 GHz. It can be seen that, as the number of notches increases, more late time ringing (distortion) and lower powers are observed in the received pulses. To evaluate the quality of the received pulses, the fidelity *F* is used. The calculated fidelities *F* for the four band-notched antennas are shown in Table 9. It can be seen that the fidelities in the face-to-face and side-by-side arrangements are about the same, which should be the case for monopole antennas. As expected, the reference UWB antenna has the fidelity of more than 97%, the best among the antennas tested. With quadruple notches, the proposed antenna has the fidelity of more than 85%. It was expected that the fidelities *F* would drop when the number of notches increased. However, Table 9 shows that the fidelity of the single band-notched antenna is less than that of the dual band-notched antenna. This is because the dual band-notched antenna has the two notches in the lower and upper WLAN bands of 5.15-5.35 GHz and 5.725-5.825 GHz, respectively. While the single band-notched antenna has the single notched-band from 5.15 GHz to 5.825 GHz, which is larger than the total of the two notches in the dual band-notched antenna. Thus the fidelity depends not only on the number of

The simulated and measured radiation patterns of the quadruple band-notched antenna at the notch frequencies of 3.14 and 6.4 GHz and passband frequencies of 2.5, 7 and 11 GHz in the x-y and x-z planes are shown in Figure 41. At all the frequencies tested, Figures 41(a), 41(c), 41(e), 41 (g) and 41(i) show that the radiation pattern for co-polarization is approximately omnidirectional and stable in the x-y plane. While Figures 41(b), 41(d), 41(f), 41(h) and 41(j) show two nulls occurring in the positive and negative z-directions, which is typical for monopole antennas. At the notch frequencies of 3.14 and 6.4 GHz, Figures 41(c), 41(d), 41(e) and 41(f) show that the gain is almost evenly suppressed in all directions. The cross-polarization

antenna

*4.5.6. Time-domain performance* 

The simulated and measured results on the peak gain and radiation efficiency of the antenna are shown in Figures 40(a) and 40(b), respectively. A relatively flat peak gain and constant efficiency are observed over the UWB, except in the notch bands. At the notch frequencies of 3.14, 4.34, 5.4 and 6.4 GHz, the measured gain is suppressed to -5.4, -4.1 and - 3.7 and -2.5 dBi, respectively, with the corresponding efficiency substantially reduced to 13.2%, 26.5%, 26.8% and 35.4%. Thus the MLs work effectively to generate a quadruple band-notched characteristic for the UWB antenna. Comparing the PCF MLs to the DCF MLs in terms of peak gain and efficiency reduction, the PCF MLs perform better than the DCF MLs. This is because the PCF MLs are closer to the feed line and thus couple more energy. There are discrepancies between the simulated and measured peak gains and efficiencies, especially in the lower frequency bands. This is mainly due to the small ground plane of the antenna, which results in current flowing back from the ground plane to the outer conductor of the feeding coaxial cable [21, 22].

**Figure 39.** Simulated and measured return losses of reference antenna and quadruple band-notched antenna

**Figure 40.** Simulated and measured (a) peak gains and (b) efficiencies of quadruple band-notched antenna

The simulated and measured radiation patterns of the quadruple band-notched antenna at the notch frequencies of 3.14 and 6.4 GHz and passband frequencies of 2.5, 7 and 11 GHz in the x-y and x-z planes are shown in Figure 41. At all the frequencies tested, Figures 41(a), 41(c), 41(e), 41 (g) and 41(i) show that the radiation pattern for co-polarization is approximately omnidirectional and stable in the x-y plane. While Figures 41(b), 41(d), 41(f), 41(h) and 41(j) show two nulls occurring in the positive and negative z-directions, which is typical for monopole antennas. At the notch frequencies of 3.14 and 6.4 GHz, Figures 41(c), 41(d), 41(e) and 41(f) show that the gain is almost evenly suppressed in all directions. The cross-polarization level is quite low, except at the highest frequency of 11 GHz, as can be seen in Figure 41.

#### *4.5.6. Time-domain performance*

266 Ultra Wideband – Current Status and Future Trends

respectively.

antenna

*4.5.5. Quadruple band-notched antenna* 

conductor of the feeding coaxial cable [21, 22].

The return losses of the reference and quadruple band-notched antennas are shown in Figure 39. The simulation and measurement return losses agree well and are larger than 10 dB across the UWB. In the notched bands, the return loss is substantially smaller than 10 dB. In the rest of the UWB, the return loss is larger than 10 dB, indicating good radiation performance. Figure 39 also shows that ML #1, being a parallel-coupled resonator at 3.14 GHz, has a spurious response at about 8.65 GHz, which is caused by the harmonic responses of the resonator. The measured results in Figure 39 show that the four notches at the frequencies of 3.14, 4.34, 5.4 and 6.4 have the bandwidth of 178, 374, 495 and 862 MHz,

The simulated and measured results on the peak gain and radiation efficiency of the antenna are shown in Figures 40(a) and 40(b), respectively. A relatively flat peak gain and constant efficiency are observed over the UWB, except in the notch bands. At the notch frequencies of 3.14, 4.34, 5.4 and 6.4 GHz, the measured gain is suppressed to -5.4, -4.1 and - 3.7 and -2.5 dBi, respectively, with the corresponding efficiency substantially reduced to 13.2%, 26.5%, 26.8% and 35.4%. Thus the MLs work effectively to generate a quadruple band-notched characteristic for the UWB antenna. Comparing the PCF MLs to the DCF MLs in terms of peak gain and efficiency reduction, the PCF MLs perform better than the DCF MLs. This is because the PCF MLs are closer to the feed line and thus couple more energy. There are discrepancies between the simulated and measured peak gains and efficiencies, especially in the lower frequency bands. This is mainly due to the small ground plane of the antenna, which results in current flowing back from the ground plane to the outer

**Figure 39.** Simulated and measured return losses of reference antenna and quadruple band-notched

For comparison, the studies of time-domain performances are carried out as described in previous sections on the single, dual, triple and quadruple band-notched antennas and results are all plotted in Figure 42. The single, dual and triple band-notched antennas have notches at 5.5 GHz, 5.25 & 5.775 GHz, 3.5 & 5.25 & 5.775 GHz, respectively, while the quadruple band-notched antenna has notches at 3.14 & 4.34 & 5.4 & 6.4 GHz. It can be seen that, as the number of notches increases, more late time ringing (distortion) and lower powers are observed in the received pulses. To evaluate the quality of the received pulses, the fidelity *F* is used. The calculated fidelities *F* for the four band-notched antennas are shown in Table 9. It can be seen that the fidelities in the face-to-face and side-by-side arrangements are about the same, which should be the case for monopole antennas. As expected, the reference UWB antenna has the fidelity of more than 97%, the best among the antennas tested. With quadruple notches, the proposed antenna has the fidelity of more than 85%. It was expected that the fidelities *F* would drop when the number of notches increased. However, Table 9 shows that the fidelity of the single band-notched antenna is less than that of the dual band-notched antenna. This is because the dual band-notched antenna has the two notches in the lower and upper WLAN bands of 5.15-5.35 GHz and 5.725-5.825 GHz, respectively. While the single band-notched antenna has the single notched-band from 5.15 GHz to 5.825 GHz, which is larger than the total of the two notches in the dual band-notched antenna. Thus the fidelity depends not only on the number of notches, but also the bandwidths and locations of the notches.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 269

**Figure 42.** Measured transmit pulses with spectrum from 3.1 to 10.6 GHz and received pulses of (a)

*Face-to-Face 0.9825 0.9444 0.9590 0.9184 0.8918 Side-by-Side 0.9744 0.9463 0.9557 0.9281 0.8538* 

In this chapter, the designs of band-notched characteristics for compact UWB antennas have been presented. The single, dual, triple and quadruple band-notched UWB antennas using

have been designed and studied using computer simulation. For verification of the simulation results, these antennas have been fabricated and measured using the antenna measurement system, Starlab. The bandwidths and center frequencies of all these notches can be controlled independently by adjusting the dimensions of the resonators. In the frequency domain, the simulated and measured return losses of the antennas agree well. The UWB monopole antennas have approximately omnidirectional radiation patterns with

*Dual Band-Notched Antenna* 

*Triple Band-Notched Antenna* 

λ

*Quadruple Band-Notched Antenna* 

/4-resonators and the MLs

*Single Band-Notched Antenna* 

single, (b) dual, (c) triple and (d) quadruple band-notched antennas

**Table 9.** Calculated Fidelity For Different Band-Notched Antennas

different resonator structures including the CPW resonators, the

*Reference Antenna* 

**5. Conclusions** 

**Figure 41.** Simulated and measured radiation patterns of quadruple band-notched antenna. x-y plane: (a) 2.5 GHz, (c) 3.14 GHz, (e) 6.4 GHz, (g) 7 GHz and (i) 11 GHz. x-z plane: (b) 2.5 GHz, (d) 3.14 GHz, (f) 6.4 GHz, (h) 7 GHz and (j) 11 GHz

**Figure 42.** Measured transmit pulses with spectrum from 3.1 to 10.6 GHz and received pulses of (a) single, (b) dual, (c) triple and (d) quadruple band-notched antennas


**Table 9.** Calculated Fidelity For Different Band-Notched Antennas

#### **5. Conclusions**

268 Ultra Wideband – Current Status and Future Trends

6.4 GHz, (h) 7 GHz and (j) 11 GHz

**Figure 41.** Simulated and measured radiation patterns of quadruple band-notched antenna. x-y plane: (a) 2.5 GHz, (c) 3.14 GHz, (e) 6.4 GHz, (g) 7 GHz and (i) 11 GHz. x-z plane: (b) 2.5 GHz, (d) 3.14 GHz, (f) In this chapter, the designs of band-notched characteristics for compact UWB antennas have been presented. The single, dual, triple and quadruple band-notched UWB antennas using different resonator structures including the CPW resonators, the λ/4-resonators and the MLs have been designed and studied using computer simulation. For verification of the simulation results, these antennas have been fabricated and measured using the antenna measurement system, Starlab. The bandwidths and center frequencies of all these notches can be controlled independently by adjusting the dimensions of the resonators. In the frequency domain, the simulated and measured return losses of the antennas agree well. The UWB monopole antennas have approximately omnidirectional radiation patterns with good band-notched characteristics. The simulated and measured results have shown that substantial reductions in efficiency and peak gain can be achieved at the notch frequencies. In the time domain, the pulse responses of these notched antennas have been measured inside the quiet zone of the anechoic chamber in The University of Hong Kong. Fidelity has been used to evaluate the time-domain performance of these antennas. Results have shown that all these band-notched antennas designed have the fidelities of more than 85%, compared to 97% for the reference UWB antenna.

Creating Band-Notched Characteristics for Compact UWB Monopole Antennas 271

[14] Y. J. Cho*, et al.*, "A miniature UWB planar monopole antenna with 5 GHz band-rejection

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