**Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter**

Matthieu Chatras , Stéphane Bila, Sylvain Giraud, Lise Catherinot, Ji Fan, Dominique Cros, Michel Aubourg, Axel Flament, Antoine Frappé, Bruno Stefanelli, Andreas Kaiser, Andreia Cathelin, Jean Baptiste David, Alexandre Reinhardt, Laurent Leyssenne and Eric Kerhervé

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/56026

**1. Introduction** 

322 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

*ferroelectrics, and frequency control, vol. 52, no. 6*, june 2005

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[21] Lukas Baumgartel and Eun Sok Kim. Experimental Optimization of Electrodes for High Q, High Frequency HBAR. *2009 IEEE International Ultrasonics Symposium Proceedings.* 

[22] S.G.Alekseev, I.M.Kotelyanskii, G.D.Mansfeld, N.I.Polzikova. Energy Trapping in HBARs Based on Cubic Crystals. *2006 IEEE Ultrasonics Symposium*. pp. 1478-1480, 2006 [23] Hui Zhang, Zuoqing Wang, and Shu-Yi Zhang. Electrode Effects on Frequency Spectra and Electromechanical Coupling Factors of HBAR. *IEEE transactions on ultrasonics,* 

[24] Hui Zhang, Shu-Yi Zhang, Kai Zheng. Parameter characterization of high-overtone bulk acoustic resonators by resonant spectrum method. *Ultrasonics 43 (2005)* pp. 635–642.

[25] J.J. Campbell, W.R. Jones, A method for estimating crystals cuts and propagation direction for excitation of piezoelectric surface waves. *IEEE Trans. On Sonics and* 

[26] T. Baron, D. Gachon, *et al.*. Temperature Compensated Radio-Frequency Harmonic Bulk

[27] T. Baron, G. Martin, E. Lebrasseur, B. Francois, S. Ballandras, P.-P. Lasagne, A. Reinhardt, L. Chomeloux, D. Gachon, J.-M. Lesage. RF oscillators stabilized by temperature compensated HBARs based on LiNbO3/Quartz combination. *Frequency Control and the European Frequency and Time Forum (FCS), 2011 Joint Conference of the IEEE* 

[28] J.-M Friedt, C. Droit, G. Martin, and S. Ballandras, "A wireless interrogation system exploiting narrowband acoustic resonator for remote physical quantity measurement",

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[31] G.D. Mansfeld. THEORY OF HIGH OVERTONE BULK ACOUSTIC WAVE RESONATOR AS A GAS SENSOR. *13th International Conference on Microwaves, Radar* 

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[34] Hongyu Yu, Chuang-yuan Lee, Wei Pang, Hao Zhang, Alan Brannon, John Kitching, and Eun Sok Kim. HBAR-Based 3.6 GHz Oscillator with Low Power Consumption and Low Phase Noise. *IEEE Transactions on Ultrasonics, Ferroelectrics, and Frequency Control,* 

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Bulk-Acoustic Wave (BAW) resonators and filters are highly integrated devices, which represent an effective alternative for narrow-band components (up to 5% fractional bandwidth) up to few GHz [1].

This chapter presents the integration of a BAW filter and of a BAW duplexer in a UMTS transmitter. The first section details one dimensional and three-dimensional techniques for the modeling and the design of BAW resonators. The second section proposes a synthesis approach for dimensioning BAW filters and the third section illustrates the approach with the characterization of several fabricated prototypes. Finally, The UMTS transmitter incorporating a BAW filter and a BAW duplexer is described with a particular emphasis on the performances of these devices.

## **2. Model and design of bulk acoustic wave resonators**

## **2.1. Modeling of a BAW resonator in 1 dimension**

The proposed method compares the impedance of a piezoelectric resonator obtained by both an electrical equivalent model and a piezoelectric model. By this way, it is possible to obtain the values of the electrical model as functions of all geometrical and material characteristics. The two models and their relation are described in the following sections.

#### *2.1.1. Resonator impedance based on electrical (BVD) model*

A piezoelectric resonator can be modeled by the lossless BVD (Butterworth Van Dick) model as shown in Figure 1(a)) [2], [3]. *Co* is the geometric capacitance of the structure and the *Lm Cm* series circuit (called the motional arm) represents the mechanical resonance (motional behavior). According to the circuit in Figure 1 (a), two resonances are obtained, and the equivalent impedance *ZBVD* of this circuit can be easily derived:

$$f\_s = \frac{1}{2\pi\sqrt{L\_mC\_m}}\tag{1}$$

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 325

(7)

<sup>1</sup> 3 3 T 0 0 ; T z 0 (8)

11 21 11 21 T z T z ; u z u z (9)

22 32 22 32 T z T z ; u z u z (10)

<sup>2</sup> ( ) *<sup>z</sup> U Ez z dz <sup>z</sup>* (11)

I j *wQ* (12)

*Q Dz S* (13)

*I wS Dz* j (14)

 

<sup>2</sup> ( ) ( ) *T z u z*

In the above equations, T is the mechanical stress tensor, S is the strain tensor, E is the electric field, D is the electrical displacement vector (C/m2), ρ is the density, u is the mechanical displacement vector, c is the elastic stiffness tensor calculated at constant electric

The potential difference u on the piezoelectric layer can be obtained by integrating Ez (the

1

By definition, the current I is the temporal derivation of the charge Q at the surface of the

The continuity of the normal component of the electrical displacement vector D at the interface piezoelectric-metal makes it possible to express the charge Q as a function of D and

Consequently, the impedance of the piezoelectric layer can be obtained in function of the thickness of the used materials and the dimensions of the resonator, as written in equation

Where *e2* is the piezoelectric tensor of the piezoelectric layer, *A* is the surface (area) of the electrodes, *ε2* is the permittivity of the piezoelectric layer, and *r11*, *r12*, *r21*, *r22*, *α,* and are

*<sup>U</sup> z z er r er r <sup>Z</sup> I j A* 

2 1 2 21 11 2 22 12 2 [( ) ( ) ( )] /

(15)

field, e is the piezoelectric tensor (C/m2), and ε is the relative permittivity.

Using the boundaries conditions for T (z) and u(z), we have:

electric field) on the thickness of the considered layer:

electrodes, which in sinusoidal mode is equivalent to:

of the surface of the metal electrodes:

(15):

The expression of the current becomes then:

*eq*

expressions in function of known constants.

*z*

$$f\_p = f\_s \sqrt{\frac{\mathcal{C}\_m + \mathcal{C}\_o}{\mathcal{C}\_o}} \tag{2}$$

$$Z\_{\rm BVD} = \frac{j(\rho \mathcal{L}\_m + 1/\rho \mathcal{C}\_m)}{1 - \alpha^2 \mathcal{C}\_o \mathcal{L}\_m + \mathcal{C}\_o / \mathcal{C}\_m} \tag{3}$$

**Figure 1.** (a) Equivalent circuit of the lossless BVD model. (b) One-dimensional structure piezoelectric resonator.

#### *2.1.2. Resonator impedance based on piezoelectric equations*

The description of a piezoelectric resonator, made of a single piezoelectric layer and two thin electrodes as depicted in Figure 1 (b), is considered. Using piezoelectric fundamental equations [4], one can write:

$$T(z) = cS(z) - eE(z) \tag{4}$$

$$D = \varepsilon S(z) + \varepsilon E(z) \tag{5}$$

$$S(z) = \frac{\partial \mu(z)}{\partial z} \tag{6}$$

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 325

$$-\rho^2 u(z) = \frac{\hat{\sigma}T(z)}{\hat{\sigma}z} \tag{7}$$

In the above equations, T is the mechanical stress tensor, S is the strain tensor, E is the electric field, D is the electrical displacement vector (C/m2), ρ is the density, u is the mechanical displacement vector, c is the elastic stiffness tensor calculated at constant electric field, e is the piezoelectric tensor (C/m2), and ε is the relative permittivity.

Using the boundaries conditions for T (z) and u(z), we have:

324 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

A piezoelectric resonator can be modeled by the lossless BVD (Butterworth Van Dick) model as shown in Figure 1(a)) [2], [3]. *Co* is the geometric capacitance of the structure and the *Lm Cm* series circuit (called the motional arm) represents the mechanical resonance (motional behavior). According to the circuit in Figure 1 (a), two resonances are obtained, and the

1

*m m*

*m o*

*o*

( 1/ ) 1 / *m m*

*om o m*

*CL C C*

*L C* (1)

(2)

*T z cS z eE z* () () () (4)

(5)

(6)

(3)

2 *<sup>s</sup>*

*p s*

*BVD*

*2.1.2. Resonator impedance based on piezoelectric equations* 

resonator.

equations [4], one can write:

*C C f f <sup>C</sup>*

2

**Figure 1.** (a) Equivalent circuit of the lossless BVD model. (b) One-dimensional structure piezoelectric

The description of a piezoelectric resonator, made of a single piezoelectric layer and two thin electrodes as depicted in Figure 1 (b), is considered. Using piezoelectric fundamental

> *D eS z E z* () ()

> > ( ) ( ) *u z S z z*

*jL C <sup>Z</sup>*

*f*

*2.1.1. Resonator impedance based on electrical (BVD) model* 

equivalent impedance *ZBVD* of this circuit can be easily derived:

$$\mathbf{T\_1(0)} = \mathbf{0} \; ; \quad \mathbf{T\_3(z\_3)} = \mathbf{0} \tag{8}$$

$$\mathbf{T\_1(z\_1)} = \mathbf{T\_2(z\_1)} \; ; \quad \mathbf{u\_1(z\_1)} = \mathbf{u\_2(z\_1)} \tag{9}$$

$$\mathbf{T\_2(z\_2)} = \mathbf{T\_3(z\_2)} \; ; \quad \mathbf{u\_2(z\_2)} = \mathbf{u\_3(z\_2)} \tag{10}$$

The potential difference u on the piezoelectric layer can be obtained by integrating Ez (the electric field) on the thickness of the considered layer:

$$
\Delta U = -\left[ \right]\_{z2}^{z1} \to \text{z}(z)dz \tag{11}
$$

By definition, the current I is the temporal derivation of the charge Q at the surface of the electrodes, which in sinusoidal mode is equivalent to:

$$\mathbf{I} = \mathbf{j}w\mathbf{Q} \tag{12}$$

The continuity of the normal component of the electrical displacement vector D at the interface piezoelectric-metal makes it possible to express the charge Q as a function of D and of the surface of the metal electrodes:

$$Q = Dz \text{ S} \tag{13}$$

The expression of the current becomes then:

$$I = \text{j}wSDz \tag{14}$$

Consequently, the impedance of the piezoelectric layer can be obtained in function of the thickness of the used materials and the dimensions of the resonator, as written in equation (15):

$$Z\_{eq} = \frac{\mathcal{U}}{I} = \frac{[(z\_2 - z\_1) - \alpha e\_2(r\_{21} - r\_{11}) - \beta e\_2(r\_{22} - r\_{12})] / \varepsilon\_2}{j\alpha A} \tag{15}$$

Where *e2* is the piezoelectric tensor of the piezoelectric layer, *A* is the surface (area) of the electrodes, *ε2* is the permittivity of the piezoelectric layer, and *r11*, *r12*, *r21*, *r22*, *α,* and are expressions in function of known constants.

The impedance can be evaluated knowing the material properties and dimensions. This method can also be used for more complex structures, such as SCF (Stacked Crystal Filters), or CRF (Coupled Resonator Filters) resonators or filters, accounting for all the layers.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 327

With: T: mechanical stress (Pa), E: electric field (V/m), S: mechanical strain, D: electric displacement(C/m²), c: stiffness tensor (Pa), : permittivity tensor (F/m) and e: piezoelectric

Unfortunately, one-dimensional approximation becomes too restrictive when we need to predict spurious modes that may appear with lateral direction mode coupling or with resonators electrical or mechanical cross coupling. Thus a 3D simulation tool is needed to

In the following sections, examples of 3D FEM computation will be proposed and described

We have analyzed the suspended resonator structure presented in Figure 2. The structure is clamped on lateral sides (no mechanical displacement in space directions), the bottom

In order to identify spurious modes, the mechanical displacement for four modes has been

The first mode is the so-called thickness extensional mode. It corresponds to the maximum energy coupling and to minimum mechanical losses. This mode is the one taken into

The second mode is a cavity mode. In FEM, in order to keep the problem size finite, the physical domain needs to be truncated. This truncation introduces artificial boundaries where artificial boundary conditions are considered. Domain truncation causes reflection of the waves on clamped lateral sides. Then a standing wave may appear at certain frequencies. Those spurious modes are modeling errors and will be suppressed by mesh

electrode is grounded and a potential constraint is applied to the top electrode.

tensor (C/m²)

compute the resonator in three dimensions.

*2.2.2. Mechanical displacement modes* 

**Figure 2.** 3D suspended resonator structure.

calculated, as displayed in Figure 3.

account in 1D analytical models.

apodization.

highlighting the advantages of 3D simulations.

D e \* E e \* S k kl l ijk ij (18)

#### *2.1.3. Equivalence of one dimensional models*

Using a least squares method, the two expressions in equations (3) and (15) can be equated. By this way, the values of *Co*, *Cm* and *Lm* for different resonator areas *A* and for the thicknesses of each layer can be obtained.

For example, the expression of *Lm* as a function of the surface *A* and the thickness *t* of the top electrode is presented. For fixed values, *[A1 A2 A3······Ai]* and *[t1 t2 t3······ ti]*, we obtain corresponding values of *Lm*. For each same thickness *ti*, we can consider that *Lm* is only varying according to the surface *A*. As shown in equation (16), we can find a polynomial function in the variable *A* fitting the values of *Lm*. The polynomial coefficients (*Wi*, *Xi*, *Yi* and *Zi*) are now independent of the surface *A* and depend only on the thickness *t*. In a second time, using the same method, the coefficients can be expressed as a function of variable *t*.

$$\begin{aligned} L\_m &= W\_i A^n + X\_i A^{n-1} + \cdots + Y\_i A + Z\_i \\ &= \left(\sum\_{j=1}^N a\_j t^{j-1}\right) A^n + \left(\sum\_{j=1}^N b\_j t^{j-1}\right) A^{n-1} + \cdots + \left(\sum\_{j=1}^N c\_j t^{j-1}\right) A + \left(\sum\_{j=1}^N d\_j t^{j-1}\right) \end{aligned} \tag{16}$$

This identification method can also be applied to the other variables, such as the thickness of the loading layer. The three elements of the BVD model (*Co*, *Cm* and *Lm*) can be determined as functions of the piezoelectric resonator dimensions (thickness of each layer and surface of resonator). According to the expressions of *Co*, *Lm*, and *Cm*, surfaces and thicknesses can be optimized with an electrical software for designing a BAW resonator or a BAW filter.

#### **2.2. Model in 3 dimensions**

Even though Film Bulk Acoustic Resonator simulation with 1D model enables to quickly compute complex frequency response as filters, it becomes too restrictive when spurious modes in lateral dimensions have to be predicted. The 3D Finite Elements Method (FEM) enables to investigate the effect of the electrode shape on the spurious modes that are present in the electrical impedance. In order to reduce or to suppress these modes, solutions have to be investigated.

#### *2.2.1. Finite element model*

Piezoelectricity is a phenomenon which couples electrical and mechanical domains. It can be modeled into coupled equations:

$$\mathbf{T\_{ij}} = \mathbf{c\_{ijkl}} \mathbf{\*}\ \mathbf{S\_{kl}} - \mathbf{e\_{ijk}} \ \mathbf{\*}\ \to\_{\mathbf{k}} \tag{17}$$

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 327

$$\mathbf{D}\_{\mathbf{k}} = \mathbf{e}\_{\mathbf{k}\mathbf{l}}\text{ \* }\to\!\!\mathbf{e}\_{\mathbf{l}}\text{ \* }\mathbf{e}\_{\mathbf{i}\{\mathbf{k}\}}\text{ \* }\mathbf{S}\_{\mathbf{i}\{\mathbf{j}\}}\tag{18}$$

With: T: mechanical stress (Pa), E: electric field (V/m), S: mechanical strain, D: electric displacement(C/m²), c: stiffness tensor (Pa), : permittivity tensor (F/m) and e: piezoelectric tensor (C/m²)

Unfortunately, one-dimensional approximation becomes too restrictive when we need to predict spurious modes that may appear with lateral direction mode coupling or with resonators electrical or mechanical cross coupling. Thus a 3D simulation tool is needed to compute the resonator in three dimensions.

In the following sections, examples of 3D FEM computation will be proposed and described highlighting the advantages of 3D simulations.

## *2.2.2. Mechanical displacement modes*

326 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

1

*mi i i i*

*L WA XA YA Z*

*n n*

*2.1.3. Equivalence of one dimensional models* 

thicknesses of each layer can be obtained.

variable *t*.

**2.2. Model in 3 dimensions** 

have to be investigated.

*2.2.1. Finite element model* 

modeled into coupled equations:

The impedance can be evaluated knowing the material properties and dimensions. This method can also be used for more complex structures, such as SCF (Stacked Crystal Filters),

Using a least squares method, the two expressions in equations (3) and (15) can be equated. By this way, the values of *Co*, *Cm* and *Lm* for different resonator areas *A* and for the

For example, the expression of *Lm* as a function of the surface *A* and the thickness *t* of the top electrode is presented. For fixed values, *[A1 A2 A3······Ai]* and *[t1 t2 t3······ ti]*, we obtain corresponding values of *Lm*. For each same thickness *ti*, we can consider that *Lm* is only varying according to the surface *A*. As shown in equation (16), we can find a polynomial function in the variable *A* fitting the values of *Lm*. The polynomial coefficients (*Wi*, *Xi*, *Yi* and *Zi*) are now independent of the surface *A* and depend only on the thickness *t*. In a second time, using the same method, the coefficients can be expressed as a function of

1 1 1 1 1

(16)

1 1 1 1

optimized with an electrical software for designing a BAW resonator or a BAW filter.

*at A bt A ct A dt*

This identification method can also be applied to the other variables, such as the thickness of the loading layer. The three elements of the BVD model (*Co*, *Cm* and *Lm*) can be determined as functions of the piezoelectric resonator dimensions (thickness of each layer and surface of resonator). According to the expressions of *Co*, *Lm*, and *Cm*, surfaces and thicknesses can be

Even though Film Bulk Acoustic Resonator simulation with 1D model enables to quickly compute complex frequency response as filters, it becomes too restrictive when spurious modes in lateral dimensions have to be predicted. The 3D Finite Elements Method (FEM) enables to investigate the effect of the electrode shape on the spurious modes that are present in the electrical impedance. In order to reduce or to suppress these modes, solutions

Piezoelectricity is a phenomenon which couples electrical and mechanical domains. It can be

ij ijkl kl ijk k T c \* S – e \* E (17)

 

*N N N N j j n n j j j j j j j j j j*

or CRF (Coupled Resonator Filters) resonators or filters, accounting for all the layers.

We have analyzed the suspended resonator structure presented in Figure 2. The structure is clamped on lateral sides (no mechanical displacement in space directions), the bottom electrode is grounded and a potential constraint is applied to the top electrode.

**Figure 2.** 3D suspended resonator structure.

In order to identify spurious modes, the mechanical displacement for four modes has been calculated, as displayed in Figure 3.

The first mode is the so-called thickness extensional mode. It corresponds to the maximum energy coupling and to minimum mechanical losses. This mode is the one taken into account in 1D analytical models.

The second mode is a cavity mode. In FEM, in order to keep the problem size finite, the physical domain needs to be truncated. This truncation introduces artificial boundaries where artificial boundary conditions are considered. Domain truncation causes reflection of the waves on clamped lateral sides. Then a standing wave may appear at certain frequencies. Those spurious modes are modeling errors and will be suppressed by mesh apodization.

Standing wave modes are harmonic thickness extensional modes due to acoustic wave reflection on top electrode edges. They appear as spurious modes in the electrical response. In order to use this electrical response for oscillator or filter applications, we need to suppress these modes. Different solutions can be considered.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 329

**Figure 5.** Apodized top electrode electrical admittance for two apodization angle values

**Figure 6.** Framed top electrode suspended resonator 3D structure

**Figure 7.** Framed top electrode suspended resonator thickness extensional mode

Another solution proposed in [6] is considered. A narrow region is deposited at the edge of the suspended resonator top electrode as described in Figure 6. This thickened region constitutes a frame that matches the acoustic impedance and suppresses reflection on top electrode edges. Within certain optimum range for the edge region width, the resonator operates in a mode where the mechanical displacement is constant on the top electrode

*2.2.3.2. Edge loading* 

surface (Figure 7).

**Figure 3.** Top view of the suspended resonator mode shapes at resonant frequencies (The only top electroded region is the square in the middle).

## *2.2.3. Suppression of standing wave modes*

## *2.2.3.1. Apodization*

The first solution consists of cut a triangular part of the square top electrode and to paste it on another edge as shown in Figure 4. Then a quadrilateral electrode is obtained with no parallel sides [5].

**Figure 4.** Apodization applied to a square electrode

Standing waves cannot appear since multiple reflections are not constructive. One can notice from Figure 5 that standing wave modes are not or weakly coupled as the apodization angle increases. A 3D simulation software enables the analysis of an entire apodized resonator and is very useful in this case for determining an optimum apodization angle.

**Figure 5.** Apodized top electrode electrical admittance for two apodization angle values

#### *2.2.3.2. Edge loading*

328 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

suppress these modes. Different solutions can be considered.

electroded region is the square in the middle).

*2.2.3. Suppression of standing wave modes* 

**Figure 4.** Apodization applied to a square electrode

*2.2.3.1. Apodization* 

parallel sides [5].

Standing wave modes are harmonic thickness extensional modes due to acoustic wave reflection on top electrode edges. They appear as spurious modes in the electrical response. In order to use this electrical response for oscillator or filter applications, we need to

**Figure 3.** Top view of the suspended resonator mode shapes at resonant frequencies (The only top

The first solution consists of cut a triangular part of the square top electrode and to paste it on another edge as shown in Figure 4. Then a quadrilateral electrode is obtained with no

Standing waves cannot appear since multiple reflections are not constructive. One can notice from Figure 5 that standing wave modes are not or weakly coupled as the apodization angle increases. A 3D simulation software enables the analysis of an entire apodized resonator and

is very useful in this case for determining an optimum apodization angle.

Another solution proposed in [6] is considered. A narrow region is deposited at the edge of the suspended resonator top electrode as described in Figure 6. This thickened region constitutes a frame that matches the acoustic impedance and suppresses reflection on top electrode edges. Within certain optimum range for the edge region width, the resonator operates in a mode where the mechanical displacement is constant on the top electrode surface (Figure 7).

**Figure 6.** Framed top electrode suspended resonator 3D structure

**Figure 7.** Framed top electrode suspended resonator thickness extensional mode

It can be observed, in Figure 8, that the electrical response is standing wave modes free. Standing waves on the top electrode are not coupled. Moreover, 3D resonant frequency tends toward 1D result.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 331

We obtained an asymmetrical structure with no parallel sides. The irregular shape avoids phase reflections on the structure edges with constructive interferences, reducing the

One can observe in Figures 11, 12 and 13 that this technique enables to obtain an electric

A full 3D FEM tool is very useful to analyze and to predict the behavior of complex structures. One can take into account non-homogenous structure and non-linear materials. Obviously it is the only method to compute 3D geometries. The shape of the mechanical displacement can be obtained to identify real or non-physical modes. Solutions have been

proposed with apodized shape or with edge loading to remove parasitic modes.

**Figure 10.** Apodized cavity (Top view) and mesh

response free of spurious cavity modes.

**Figure 11.** Suspended resonator electrical admittance

occurrence of standing waves.

**Figure 8.** Framed top electrode suspended resonator admittance

## *2.2.3.3. Cavity modes suppression by resonator apodization*

In order to identify spurious modes due to domain truncation, we have changed the distance from the top electrode edge to the structure edge (A=500 m or 400 m in Figure 9). We have found that resonant conditions and resonant frequencies for cavity modes change with cavity dimensions.

The mechanical displacement for those modes is displayed in Figure 9.

**Figure 9.** Mechanical displacement for two cavity modes (Top view)

While the mode which appears at *f = 224,5 MHz* is obviously a cavity mode, we can have doubts about the one at *f = 228,4 MHz*. In order to remove all doubts, we have applied the apodization technique to the resonator edges (Figure 10).

**Figure 10.** Apodized cavity (Top view) and mesh

**Figure 8.** Framed top electrode suspended resonator admittance

The mechanical displacement for those modes is displayed in Figure 9.

**Figure 9.** Mechanical displacement for two cavity modes (Top view)

apodization technique to the resonator edges (Figure 10).

*2.2.3.3. Cavity modes suppression by resonator apodization* 

tends toward 1D result.

with cavity dimensions.

It can be observed, in Figure 8, that the electrical response is standing wave modes free. Standing waves on the top electrode are not coupled. Moreover, 3D resonant frequency

In order to identify spurious modes due to domain truncation, we have changed the distance from the top electrode edge to the structure edge (A=500 m or 400 m in Figure 9). We have found that resonant conditions and resonant frequencies for cavity modes change

While the mode which appears at *f = 224,5 MHz* is obviously a cavity mode, we can have doubts about the one at *f = 228,4 MHz*. In order to remove all doubts, we have applied the We obtained an asymmetrical structure with no parallel sides. The irregular shape avoids phase reflections on the structure edges with constructive interferences, reducing the occurrence of standing waves.

One can observe in Figures 11, 12 and 13 that this technique enables to obtain an electric response free of spurious cavity modes.

**Figure 11.** Suspended resonator electrical admittance

A full 3D FEM tool is very useful to analyze and to predict the behavior of complex structures. One can take into account non-homogenous structure and non-linear materials. Obviously it is the only method to compute 3D geometries. The shape of the mechanical displacement can be obtained to identify real or non-physical modes. Solutions have been proposed with apodized shape or with edge loading to remove parasitic modes.

332 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 333

*<sup>2</sup>* by:

(20)

(21)

(22)

*f Co Ro* (23)

 1 *p s Cm f f Co*

related to the electromechanical coupling coefficient *kt*

specified target.

ladder network

**Figure 14.** MBVD model of a BAW resonator

and 16.

*fs* and *fp* are known as the series and parallel resonant frequencies and correspond respectively to a minimum and to a maximum of the electrical impedance. *fs* and *fp* are

2

*f f <sup>k</sup>*

*p s*

*p*

*f*

4

Moreover, one can define *Qs* and *Qp* the quality factors of series and parallel resonances:

2 *<sup>s</sup>*

1

*p*

For dimensioning a BAW filter, each resonator is represented by its MBVD model and the synthesis of the BAW filter is carried out by optimizing each resonator with respect to a

BAW filters are usually implemented arranging series and shunt resonators. Basic resonator arrangements, namely ladder and lattice configurations, are defined as shown in Figures 15

**Figure 15.** Electrical impedance of shunt and series resonators for providing a band-pass filter with a

*f Lm <sup>Q</sup> Rm* 

2

*t*

*s*

*Q*

2 *<sup>p</sup>*

**Figure 12.** Suspended resonator electrical admittance around 224,5 MHz

**Figure 13.** Suspended resonator electrical admittance around 228,4 MHz

## **3. Design of bulk acoustic wave filters**

#### **3.1. Implementation**

The MBVD (Modified Butterworth Van Dyke) model, presented in Figure 14, can been used to compute the behavior of a BAW resonator [7]. Compared to the BVD model, the MBVD model incorporates resistances, which take into account losses in the piezoelectric material and in the electrodes.

This equivalent circuit resonates for two particular frequencies:

$$f\_s = \frac{1}{2\pi\sqrt{\text{Lm.Cm}}}\tag{19}$$

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 333

$$f\_p = f\_s \sqrt{1 + \frac{Cm}{Co}}\tag{20}$$

*fs* and *fp* are known as the series and parallel resonant frequencies and correspond respectively to a minimum and to a maximum of the electrical impedance. *fs* and *fp* are related to the electromechanical coupling coefficient *kt <sup>2</sup>* by:

$$k\_t^2 = \frac{\pi^2}{4} \frac{f\_p - f\_s}{f\_p} \tag{21}$$

Moreover, one can define *Qs* and *Qp* the quality factors of series and parallel resonances:

$$Q\_s = \frac{2\pi f\_s \text{ Lm}}{\text{R}m} \tag{22}$$

$$Q\_p = \frac{1}{2\pi f\_p \text{ CoRo}}\tag{23}$$

For dimensioning a BAW filter, each resonator is represented by its MBVD model and the synthesis of the BAW filter is carried out by optimizing each resonator with respect to a specified target.

BAW filters are usually implemented arranging series and shunt resonators. Basic resonator arrangements, namely ladder and lattice configurations, are defined as shown in Figures 15 and 16.

**Figure 14.** MBVD model of a BAW resonator

332 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

**Figure 12.** Suspended resonator electrical admittance around 224,5 MHz

**Figure 13.** Suspended resonator electrical admittance around 228,4 MHz

This equivalent circuit resonates for two particular frequencies:

*f*

The MBVD (Modified Butterworth Van Dyke) model, presented in Figure 14, can been used to compute the behavior of a BAW resonator [7]. Compared to the BVD model, the MBVD model incorporates resistances, which take into account losses in the piezoelectric material

> 1 2 . *<sup>s</sup>*

*Lm Cm* (19)

**3. Design of bulk acoustic wave filters** 

**3.1. Implementation** 

and in the electrodes.

**Figure 15.** Electrical impedance of shunt and series resonators for providing a band-pass filter with a ladder network

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 335

 *A*, the surface of the top electrode (*A* = *L*.*W* where *L* and *W* are the length and the width of the top electrode respectively, the aspect ratio *L*/*W* remaining generally the same for

 *d* and *l*: the thickness of the piezoelectric layer and the thickness of the loading layer respectively, which control the series resonant frequencies (*fs*) of series and parallel

The lumped elements of the MBVD model are related to these parameters through the

*A*

2 1 *<sup>s</sup>*

2 2 <sup>4</sup> <sup>1</sup> *s*

*t*

*k* 

 <sup>2</sup> 1

*C fs* 

2 *<sup>m</sup>*

0 1 2 *<sup>o</sup> p*

> []. *<sup>s</sup> <sup>L</sup> R R*

One can note that Lm, Cm and Co can also be approximated directly by polynomial

The thicknesses of the piezoelectric and loading layers and the area of each resonator are optimized with respect to the specifications. This optimization can be driven by the

expressions of the layer thicknesses and resonator area as explained in section 2.1.3.

minimization of a cost function defined by the filtering pattern [8].

*Q C fp* 

*L fs <sup>R</sup> Q* 

*s*

*p*

*f* 

*<sup>d</sup>* (24)

(27)

(28)

(29)

(30)

(25)

(26)

C ee 0 0r

*m o*

*L*

*p*

*<sup>f</sup> <sup>f</sup>*

 

<sup>2</sup> *<sup>m</sup> m*

*m*

*R*

*<sup>f</sup> C C*

*kt*

*<sup>2</sup>*, the electromechanical coupling coefficient *Qs* , the quality factor at series resonant frequency *Qp* , the quality factor at parallel resonant frequency

 *εr* , the piezoelectric material permittivity *R[]* , the square resistance of electrodes

The remaining variable parameters are:

all electrodes)

resonators.

following expressions:

With

**Figure 16.** Electrical impedance of shunt and series resonators for providing a band-pass filter with a lattice network

The ladder configuration presents a high rejection close to the filter passband but a poor out of band rejection. On the other hand, the lattice filter exhibits higher out of band rejection but a poor rejection close to the filter passband. Combining these two configurations, one can obtain a mixed ladder-lattice filter with very good properties, as shown in Figure 17. S21 parameter is the forward transmission coefficient of the filter.

**Figure 17.** Behavior of the mixed ladder lattice filter.

#### **3.2. Synthesis**

Regarding the synthesis, each resonator is characterized by several fixed technological parameters obtained by electromechanical modeling or process characterization:


The remaining variable parameters are:


The lumped elements of the MBVD model are related to these parameters through the following expressions:

$$\mathbf{C}\_0 = \mathbf{e}\_0 \mathbf{e}\_r \frac{A}{d} \tag{24}$$

$$\mathbf{C}\_{m} = \mathbf{C}\_{o} \left[ \left( \frac{f\_{s}}{f\_{p}} \right)^{2} - \mathbf{1} \right] \tag{25}$$

With

334 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

parameter is the forward transmission coefficient of the filter.

**Figure 17.** Behavior of the mixed ladder lattice filter.

**3.2. Synthesis** 

lattice network

**Figure 16.** Electrical impedance of shunt and series resonators for providing a band-pass filter with a

The ladder configuration presents a high rejection close to the filter passband but a poor out of band rejection. On the other hand, the lattice filter exhibits higher out of band rejection but a poor rejection close to the filter passband. Combining these two configurations, one can obtain a mixed ladder-lattice filter with very good properties, as shown in Figure 17. S21

Regarding the synthesis, each resonator is characterized by several fixed technological

parameters obtained by electromechanical modeling or process characterization:

$$f\_p = \frac{f\_s}{1 - \frac{4}{\pi^2}k\_t^2} \tag{26}$$

$$L\_m = \frac{1}{\mathbb{C}\_m \left(2\pi f s\right)^2} \tag{27}$$

$$R\_m = \frac{2\pi L\_m f s}{Q\_s} \tag{28}$$

$$R\_o = \frac{1}{Q\_p C\_0 \Delta \pi f p} \tag{29}$$

$$R\_s = R\_{\text{ll}} \frac{L}{\rho \nu} \tag{30}$$

One can note that Lm, Cm and Co can also be approximated directly by polynomial expressions of the layer thicknesses and resonator area as explained in section 2.1.3.

The thicknesses of the piezoelectric and loading layers and the area of each resonator are optimized with respect to the specifications. This optimization can be driven by the minimization of a cost function defined by the filtering pattern [8].

## **3.3. Electromagnetic co-simulation**

The previous synthesis relies on MBVD models of BAW resonators, which do not take into account metallic losses or couplings due to interconnections and access ports. Since metallic lines used for connecting resonators have irregular geometries depending on the arrangement of resonators, models for such elements cannot be implemented in a synthesis tool. Nevertheless a simulation is possible a posteriori with the layout of the filter in order to estimate additional losses or to check eventual couplings due to metallic lines.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 337

Figure 18 shows the co-simulation of a mixed ladder lattice filter in Agilent ADS/Momentum environment, including the layout and the motional parts directly derived

Several BAW devices have been designed using the method proposed previously. Molybden is chosen as electrode material and the sputtering method is used. Therefore, AlN films have been deposited in (002) direction with the c-axis perpendicular to the substrate surface [10-13]. The optimized resonators and filters with the proposed method have been fabricated by CEA-Leti [14]. Figures 19 and 20 show a Solidly Mounted Resonator (SMR) resonator and a comparison between the simulated result and the measurement data respectively. The apodization of the top electrode was used to avoid parasite modes. One can observe a good agreement between simulations and measurements. S21 and S11 parameters are

**4. Fabrication and characterization of BAW resonators and filters** 

respectively the forward transmission and the reflection coefficients of the filter.

**Figure 20.** Comparison between the simulated result and the measurement data for SMR resonator

from the previous synthesis.

**Figure 19.** Structure of SMR resonator

**4.1. Single resonator** 

The layout of the filter to be realized can be drawn with an electromagnetic (EM) software, e.g. Momentum included in Agilent ADS [9]. Using such a EM software, all electrostatic and electromagnetic phenomena are characterized considering the geometry and the physical characteristics of stacked layers.

The electrostatic part of MBVD resonators (*Rs*, *Ro* and *Co*) is directly taken into account in the distributed model (related particularly to the area of resonators). However, the motional part (*Rm*, *Cm* and *Lm*) is modeled by lumped elements connected through internal ports as shown in the example given in Figure 18.

**Figure 18.** UMTS filter co-simulation

Figure 18 shows the co-simulation of a mixed ladder lattice filter in Agilent ADS/Momentum environment, including the layout and the motional parts directly derived from the previous synthesis.

## **4. Fabrication and characterization of BAW resonators and filters**

## **4.1. Single resonator**

336 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

estimate additional losses or to check eventual couplings due to metallic lines.

The previous synthesis relies on MBVD models of BAW resonators, which do not take into account metallic losses or couplings due to interconnections and access ports. Since metallic lines used for connecting resonators have irregular geometries depending on the arrangement of resonators, models for such elements cannot be implemented in a synthesis tool. Nevertheless a simulation is possible a posteriori with the layout of the filter in order to

The layout of the filter to be realized can be drawn with an electromagnetic (EM) software, e.g. Momentum included in Agilent ADS [9]. Using such a EM software, all electrostatic and electromagnetic phenomena are characterized considering the geometry and the physical

The electrostatic part of MBVD resonators (*Rs*, *Ro* and *Co*) is directly taken into account in the distributed model (related particularly to the area of resonators). However, the motional part (*Rm*, *Cm* and *Lm*) is modeled by lumped elements connected through internal ports as

**3.3. Electromagnetic co-simulation** 

characteristics of stacked layers.

**Figure 18.** UMTS filter co-simulation

shown in the example given in Figure 18.

Several BAW devices have been designed using the method proposed previously. Molybden is chosen as electrode material and the sputtering method is used. Therefore, AlN films have been deposited in (002) direction with the c-axis perpendicular to the substrate surface [10-13]. The optimized resonators and filters with the proposed method have been fabricated by CEA-Leti [14]. Figures 19 and 20 show a Solidly Mounted Resonator (SMR) resonator and a comparison between the simulated result and the measurement data respectively. The apodization of the top electrode was used to avoid parasite modes. One can observe a good agreement between simulations and measurements. S21 and S11 parameters are respectively the forward transmission and the reflection coefficients of the filter.

**Figure 19.** Structure of SMR resonator

**Figure 20.** Comparison between the simulated result and the measurement data for SMR resonator

The resonant frequency is 2.212 GHz and anti-resonant frequency is 2.152 GHz, which leads to a coupling coefficient kt² equal to 6.69%. The quality factor of this resonator is around 200.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 339

**Figure 23.** Measured and computed responses (model with losses) of the 3-pole filter

Bandpass filters have been synthesised with differential structures (100input/output impedance). These filters have been fabricated by CEA-LETI and UPM (Universidad Politécnica de Madrid) [15]. Each resonator is deposited on a Bragg mirror (SiN/SiOC) and is built with an Aluminium Nitride (AlN) piezoelectric layer and two Iridium (Ir) electrodes. With this technology, top electrode thickness is defined in order to act like a loading layer [16, 17] and to reach desired resonant frequency. Ir electrodes are utilized in order to enhance electromechanical coupling [18]. Iridium is a metal that presents a high-density [19] which leads to a high acoustic impedance, a low electric resistivity, and a specific crystal structure that promotes the growth of AlN films of excellent piezoelectric activity [20].

**4.3. Differential ladder and lattice filters** 

**Figure 24.** 1.5-stage ladder filter compared to UMTS standard

## **4.2. 2-pole and 3-pole ladder filters**

2-pole and 3-pole filters shown in Figure 21 were fabricated by CEA-Leti. Measurements and simulations for each filter are presented respectively in Figures 22 and 23. One can observe a good agreement between the measured responses and the simulated ones. The measured response of the 2-pole filter presents a passband of 55 MHz centered at 2.13 GHz (fractional bandwidth of 2.58 %) and the insertion losses are about 1.5 dB. Similarly, the 3 pole filter shows a 54 MHz passband at 2.13 GHz (FBW: 2.53 %) and insertion losses of approximately 2 dB.

**Figure 21.** (a) Fabricated 2-pole filter. (b) Fabricated 3-pole filter (top view).

**Figure 22.** Measured and computed responses (model with losses) of the 2-pole filter

**Figure 23.** Measured and computed responses (model with losses) of the 3-pole filter

#### **4.3. Differential ladder and lattice filters**

338 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

**Figure 21.** (a) Fabricated 2-pole filter. (b) Fabricated 3-pole filter (top view).

**Figure 22.** Measured and computed responses (model with losses) of the 2-pole filter

**4.2. 2-pole and 3-pole ladder filters** 

approximately 2 dB.

The resonant frequency is 2.212 GHz and anti-resonant frequency is 2.152 GHz, which leads to a coupling coefficient kt² equal to 6.69%. The quality factor of this resonator is around 200.

2-pole and 3-pole filters shown in Figure 21 were fabricated by CEA-Leti. Measurements and simulations for each filter are presented respectively in Figures 22 and 23. One can observe a good agreement between the measured responses and the simulated ones. The measured response of the 2-pole filter presents a passband of 55 MHz centered at 2.13 GHz (fractional bandwidth of 2.58 %) and the insertion losses are about 1.5 dB. Similarly, the 3 pole filter shows a 54 MHz passband at 2.13 GHz (FBW: 2.53 %) and insertion losses of

> Bandpass filters have been synthesised with differential structures (100input/output impedance). These filters have been fabricated by CEA-LETI and UPM (Universidad Politécnica de Madrid) [15]. Each resonator is deposited on a Bragg mirror (SiN/SiOC) and is built with an Aluminium Nitride (AlN) piezoelectric layer and two Iridium (Ir) electrodes. With this technology, top electrode thickness is defined in order to act like a loading layer [16, 17] and to reach desired resonant frequency. Ir electrodes are utilized in order to enhance electromechanical coupling [18]. Iridium is a metal that presents a high-density [19] which leads to a high acoustic impedance, a low electric resistivity, and a specific crystal structure that promotes the growth of AlN films of excellent piezoelectric activity [20].

**Figure 24.** 1.5-stage ladder filter compared to UMTS standard

One can notice that there is a good agreement between simulated and measured responses in Figures 24 and 25. As electrical losses of Iridium were not characterized before simulation, they were not taken into account. Measured insertion losses are about 3dB for ladder structure and 6dB for lattice structure. A good out of band rejection is obtained, only limited by measurement noise.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 341

The architecture presented in Figure 26 is clearly aimed at future software defined terminals by pushing the band or standard specific components as close as possible to the antenna. The targeted standard is UMTS, one of the main mobile communication standards in Europe using 1920 – 1980 MHz frequency band for TX and 2110-2170 MHz frequency band for RX. The architecture could easily be extended to additional standards such as DCS or PCS at the

Two oversampled low-pass stages representing I and Q channels, work synchronously to generate a high sample rate 1-bit output stream [21] that feeds a digital image-reject mixer [22]. This latter stage produces a high frequency sampled two-level RF signal which exhibits very good in-band performances and a quantization noise shaping due to modulation [21]. A first stage of filtering precedes the power amplifier whose matching network converts a nominal single ended load impedance of 50 to the optimal impedance for the active stage. A duplexer is inserted in order to isolate the receiver from the transmitter while

The UMTS transmit filter is designed on a ladder-lattice topology with 50 differential input - 100 differential output ports to benefit from advantages of both structures. This filter aims at meeting the standard emission mask by lowering the out-of-band quantization noise. Figure 27 shows the output spectrum of the digital RF signal generator and indeed justifies that the transmitter requires a large amount of out-of-band filtering at the output of the modulators. The most stringent specification for UMTS is the required rejection (50 dB) in the RX band (2110 MHz - 2170 MHz). It is very difficult to reach such a high rejection at a very close bandwidth from the center frequency band. A typical lattice BAW based filter exhibits a typical attenuation of roughly 40-45dB. Consequently, the insertion of a duplexer in the UMTS transmission chain is required to achieve this high attenuation in the RX band as well as its isolation between the TX and RX paths. The design and measurement results of these BAW-based structures are presented in the following

**5.1. RF signal generator** 

*5.1.1. Third-order delta-sigma modulator* 

cost of extra BAW filters for the appropriate frequency bands.

**Figure 26.** Modules definitions for first single standard UMTS transmitter

enabling them to share a common antenna.

sections.

**Figure 25.** 2-stage lattice filter compared to UMTS standard

## **5. Integration of BAW components in a transmitter module**

This section deals with the development of a BAW-SMR technology for application to mobile multi-standard communication terminals. UMTS and DCS1800 were chosen as targeted standards to validate a bi-standard BAW-based RF architecture, presenting both reconfigurability of active parts and integration of BAW technology.

This section presents the first demonstration of a complete UMTS transmitter including a CMOS 90nm digital RF signal generator connected to a BAW filter, a Power Amplifier (PA) stage and a BAW duplexer. Although the active part was originally designed for multi-mode purposes, UMTS standard was chosen to illustrate the proof of concept. The purpose is to emphasize the use of BAW-based structure for wireless mobile communications.

In the first section the proposed multi-standard transmission chain will be detailed, and a focus on the transmit BAW filter will be discussed. The design and measurement results of the stand-alone filter will then be presented before talking about the advantages of this filter with the RF signal generator. The BAW duplexer will be described and the measurement results of this filter will be detailed. Finally, the full transmitter test bench will be highlighted and the measurement results commented on.

## **5.1. RF signal generator**

340 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

limited by measurement noise.

communications.

**Figure 25.** 2-stage lattice filter compared to UMTS standard

**5. Integration of BAW components in a transmitter module** 

reconfigurability of active parts and integration of BAW technology.

highlighted and the measurement results commented on.

This section deals with the development of a BAW-SMR technology for application to mobile multi-standard communication terminals. UMTS and DCS1800 were chosen as targeted standards to validate a bi-standard BAW-based RF architecture, presenting both

This section presents the first demonstration of a complete UMTS transmitter including a CMOS 90nm digital RF signal generator connected to a BAW filter, a Power Amplifier (PA) stage and a BAW duplexer. Although the active part was originally designed for multi-mode purposes, UMTS standard was chosen to illustrate the proof of concept. The purpose is to emphasize the use of BAW-based structure for wireless mobile

In the first section the proposed multi-standard transmission chain will be detailed, and a focus on the transmit BAW filter will be discussed. The design and measurement results of the stand-alone filter will then be presented before talking about the advantages of this filter with the RF signal generator. The BAW duplexer will be described and the measurement results of this filter will be detailed. Finally, the full transmitter test bench will be

One can notice that there is a good agreement between simulated and measured responses in Figures 24 and 25. As electrical losses of Iridium were not characterized before simulation, they were not taken into account. Measured insertion losses are about 3dB for ladder structure and 6dB for lattice structure. A good out of band rejection is obtained, only

## *5.1.1. Third-order delta-sigma modulator*

The architecture presented in Figure 26 is clearly aimed at future software defined terminals by pushing the band or standard specific components as close as possible to the antenna. The targeted standard is UMTS, one of the main mobile communication standards in Europe using 1920 – 1980 MHz frequency band for TX and 2110-2170 MHz frequency band for RX. The architecture could easily be extended to additional standards such as DCS or PCS at the cost of extra BAW filters for the appropriate frequency bands.

**Figure 26.** Modules definitions for first single standard UMTS transmitter

Two oversampled low-pass stages representing I and Q channels, work synchronously to generate a high sample rate 1-bit output stream [21] that feeds a digital image-reject mixer [22]. This latter stage produces a high frequency sampled two-level RF signal which exhibits very good in-band performances and a quantization noise shaping due to modulation [21]. A first stage of filtering precedes the power amplifier whose matching network converts a nominal single ended load impedance of 50 to the optimal impedance for the active stage. A duplexer is inserted in order to isolate the receiver from the transmitter while enabling them to share a common antenna.

The UMTS transmit filter is designed on a ladder-lattice topology with 50 differential input - 100 differential output ports to benefit from advantages of both structures. This filter aims at meeting the standard emission mask by lowering the out-of-band quantization noise. Figure 27 shows the output spectrum of the digital RF signal generator and indeed justifies that the transmitter requires a large amount of out-of-band filtering at the output of the modulators. The most stringent specification for UMTS is the required rejection (50 dB) in the RX band (2110 MHz - 2170 MHz). It is very difficult to reach such a high rejection at a very close bandwidth from the center frequency band. A typical lattice BAW based filter exhibits a typical attenuation of roughly 40-45dB. Consequently, the insertion of a duplexer in the UMTS transmission chain is required to achieve this high attenuation in the RX band as well as its isolation between the TX and RX paths. The design and measurement results of these BAW-based structures are presented in the following sections.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 343

*fs* **(MHz)** *fp* **(MHz)** *K²eff* **(%)** *Qs Qp*

**X1 X2 X3 X4 X5 X6** 

**Unloaded** 1954 2004 6.2 500 300 **Loaded** 1898 1946 6.1 500 300

*Rm* () 4.58 5.16 2.82 2.92 1.68 2.12 *Lm* (nH) 186.6 216.5 118.3 119 70.4 86.3 *Cm* (fF) 35 32 59 56 100 77 *R0* () 0.39 0.43 0.23 0.25 0.14 0.18 *Co* (pF) 0.68 0.63 1.16 1.07 1.94 1.47 *Rs* () 0.19 0.17 0.19 0.20 0.23 0.22

**Figure 29.** Scattering parameters of the UMTS filter: (a) wide band performance, (b) in band

performance, compared to UMTS standard

**Table 1.** Unloaded and loaded resonators given technological data

**Table 2.** MBVD elements values of optimised UMTS filter

**Elements values** 

**Figure 27.** Spectrum of a modulated signal, up-converted around the RF carrier.

## *5.1.2. BAW filter*

#### *5.1.2.1. Filter synthesis*

The synthesis of a UMTS filter has been performed [23], where a passband is needed for the 1920 – 1980 MHz range. Important constraints of the UMTS standard [24] are a high rejection and a high selectivity. To achieve these performances, a differential mixed ladderlattice filter topology [25] with 100differential impedance has been proposed. As explained in section 3, a mixed ladder-lattice filter as shown in Figure 28, provides sharp band edges and a good rejection in the stopband.

**Figure 28.** UMTS filter topology

After synthesis, optimal technological data for unloaded and loaded resonators are found as listed in Table 1. Related MBVD elements of each resonator are listed in Table 2 and the scattering parameters are given in Figure 29.


**Table 1.** Unloaded and loaded resonators given technological data

**Figure 27.** Spectrum of a modulated signal, up-converted around the RF carrier.

The synthesis of a UMTS filter has been performed [23], where a passband is needed for the 1920 – 1980 MHz range. Important constraints of the UMTS standard [24] are a high rejection and a high selectivity. To achieve these performances, a differential mixed ladderlattice filter topology [25] with 100differential impedance has been proposed. As explained in section 3, a mixed ladder-lattice filter as shown in Figure 28, provides sharp

After synthesis, optimal technological data for unloaded and loaded resonators are found as listed in Table 1. Related MBVD elements of each resonator are listed in Table 2 and the

*5.1.2. BAW filter* 

*5.1.2.1. Filter synthesis* 

**Figure 28.** UMTS filter topology

scattering parameters are given in Figure 29.

band edges and a good rejection in the stopband.


**Table 2.** MBVD elements values of optimised UMTS filter

**Figure 29.** Scattering parameters of the UMTS filter: (a) wide band performance, (b) in band performance, compared to UMTS standard

One can observe, in Figure 30, that scattering parameters completely fulfill the specifications. Moreover, the layout of the filter has been co-simulated with a 2.5D electromagnetic software in order to estimate additional losses and to characterize eventual couplings due to metallic lines. As shown in Figure 30, the co-simulation is still in line with the specifications.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 345

**Figure 32.** Measurement of the UMTS filter, compared to UMTS standard

The all digital RF signal generator architecture is presented in Figure 33. A first stage oversamples the baseband I and Q signals to reach the modulator frequency. The principle of ΔΣ modulation and digital mixing has been explained in section 5.1 and one can find further explanations in [21]. The generator output signal is a very high speed 1-bit signal with bandpass shaped quantization noise centered on the standard transmit band. The differential output buffer is able to drive a differential 50Ω load with a 1V power supply, at very low output impedance of 0.6Ω. As the BAW filter needs a 50Ω differential input impedance match, 25Ω resistors need to be inserted between the signal generator and the BAW filter. This very simple passive and unfortunately unmatched matching network degrades the filter transfer function, introducing ripple in the pass band and lowering out-of-band attenuation. Both the RF signal generator and the BAW filter are molded with a resist on a module which is then

soldered on a larger PCB. The output of this subset circuit is 100 Ω differential [26].

outputs an RF modulated signal at 1.95GHz using the first image-band.

The baseband input signals are generated by an external Arbitrary Waveform Generator (Tektronix AWG 420), in which a Matlab sequence has been programmed to generate a WCDMA modulated signal on I and Q channels. The external clock reference comes from a synthesizer delivering a clock signal at 2.6GHz with an output level of –5dBm. This clock is the nominal frequency clock of the modulators and of the digital image-reject mixer and fully determines the center frequency of the RF signal. Using a 2.6GHz, the ΔΣ generator

The measurement results (Figure 34) consist of a spectrum analysis of output signals to evaluate the benefits of using BAW filtering in a modulation approach. It clearly shows

**5.2. Complete UMTS transmitter** 

*5.2.1. RF Signal generator* 

**Figure 30.** EM co-simulation of the UMTS filter, compared to UMTS standard

#### *5.1.2.2. Fabrication and measurements*

The filter has been fabricated by CEA-Leti. Each resonator is deposited on a Bragg mirror (SiN/SiOC), using an Aluminium Nitride (AlN) piezoelectric layer, two Molybdenum (Mo) electrodes and a Silicon Oxide (SiO2) loading layer. A photo of the filter is presented in Figure 31. The measured unpackaged quality factor at resonance frequency is close to 800 for the series resonators and close to 300 for the parallel ones. The measured response of the filter is presented in Figure 32. Insertion losses are around 3 dB and the required rejection and selectivity are fulfilled. The bandwidth is slightly reduced due to a lower resonant frequency for series resonators during fabrication

**Figure 31.** Fabricated filter with differential accesses.

**Figure 32.** Measurement of the UMTS filter, compared to UMTS standard

## **5.2. Complete UMTS transmitter**

#### *5.2.1. RF Signal generator*

344 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

**Figure 30.** EM co-simulation of the UMTS filter, compared to UMTS standard

The filter has been fabricated by CEA-Leti. Each resonator is deposited on a Bragg mirror (SiN/SiOC), using an Aluminium Nitride (AlN) piezoelectric layer, two Molybdenum (Mo) electrodes and a Silicon Oxide (SiO2) loading layer. A photo of the filter is presented in Figure 31. The measured unpackaged quality factor at resonance frequency is close to 800 for the series resonators and close to 300 for the parallel ones. The measured response of the filter is presented in Figure 32. Insertion losses are around 3 dB and the required rejection and selectivity are fulfilled. The bandwidth is slightly reduced due to a lower resonant

*5.1.2.2. Fabrication and measurements* 

frequency for series resonators during fabrication

**Figure 31.** Fabricated filter with differential accesses.

the specifications.

One can observe, in Figure 30, that scattering parameters completely fulfill the specifications. Moreover, the layout of the filter has been co-simulated with a 2.5D electromagnetic software in order to estimate additional losses and to characterize eventual couplings due to metallic lines. As shown in Figure 30, the co-simulation is still in line with

> The all digital RF signal generator architecture is presented in Figure 33. A first stage oversamples the baseband I and Q signals to reach the modulator frequency. The principle of ΔΣ modulation and digital mixing has been explained in section 5.1 and one can find further explanations in [21]. The generator output signal is a very high speed 1-bit signal with bandpass shaped quantization noise centered on the standard transmit band. The differential output buffer is able to drive a differential 50Ω load with a 1V power supply, at very low output impedance of 0.6Ω. As the BAW filter needs a 50Ω differential input impedance match, 25Ω resistors need to be inserted between the signal generator and the BAW filter. This very simple passive and unfortunately unmatched matching network degrades the filter transfer function, introducing ripple in the pass band and lowering out-of-band attenuation. Both the RF signal generator and the BAW filter are molded with a resist on a module which is then soldered on a larger PCB. The output of this subset circuit is 100 Ω differential [26].

> The baseband input signals are generated by an external Arbitrary Waveform Generator (Tektronix AWG 420), in which a Matlab sequence has been programmed to generate a WCDMA modulated signal on I and Q channels. The external clock reference comes from a synthesizer delivering a clock signal at 2.6GHz with an output level of –5dBm. This clock is the nominal frequency clock of the modulators and of the digital image-reject mixer and fully determines the center frequency of the RF signal. Using a 2.6GHz, the ΔΣ generator outputs an RF modulated signal at 1.95GHz using the first image-band.

> The measurement results (Figure 34) consist of a spectrum analysis of output signals to evaluate the benefits of using BAW filtering in a modulation approach. It clearly shows

that the out-of band quantization noise nearby the central bandwidth has been reduced below the noise level of the spectrum analyzer in this setup, due to the near-band high rejection of the ladder BAW filter configuration. Emission specifications in DCS are satisfied whereas more than 20dB of attenuation is still needed in UMTS RX band. Far-band filtering (thanks to lattice configuration) is also efficient, resulting in more than 35dB attenuation at low frequencies. In the first image band, the measured ACLR are 43 and 42dB, respectively at 5MHz and 10MHz offsets. The EVM is 3.7% and the measured channel power is -27dBm. This low value is due to the use of the first image and the loss in the series resistors.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 347

The BAW duplexer has been designed using a mix of the exposed BAW filter methodology and co-simulation with electromagnetic tools [27], [28], [29]. The topology of this duplexer is

The BAW duplexer was made with two BAW filters and a glass substrate containing high quality passive elements (IPD from ST Microelectronics [30]). Each BAW filter occupies 1mm². The TX and RX filters were flip-chipped on the 3.9x3.9mm² substrate, as shown in

**Figure 36.** Left: Photograph of bumps and micropackaged TX filter. Center and right: On probe

*5.2.2. BAW duplexer* 

shown in Figure 35.

**Figure 35.** BAW duplexer topology

measurement of the BAW duplexer

Figure 36.

The strongest constraint in UMTS transmission architecture is the very high level of rejection needed in the UMTS reception band which is very close to the transmission band. The measurement results show that a ladder-lattice BAW filter is not sufficient to fulfill this specification. A BAW-based duplexer, providing further isolation between both signal paths is consequently mandatory and will be described in the following section.

**Figure 33.** Signal generator and BAW filter architecture (left), photograph of the DS modulator and BAW assembly

**Figure 34.** Delta-sigma digital generator output spectrum before (red) and after (black) BAW filtering

## *5.2.2. BAW duplexer*

346 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

that the out-of band quantization noise nearby the central bandwidth has been reduced below the noise level of the spectrum analyzer in this setup, due to the near-band high rejection of the ladder BAW filter configuration. Emission specifications in DCS are satisfied whereas more than 20dB of attenuation is still needed in UMTS RX band. Far-band filtering (thanks to lattice configuration) is also efficient, resulting in more than 35dB attenuation at low frequencies. In the first image band, the measured ACLR are 43 and 42dB, respectively at 5MHz and 10MHz offsets. The EVM is 3.7% and the measured channel power is -27dBm.

The strongest constraint in UMTS transmission architecture is the very high level of rejection needed in the UMTS reception band which is very close to the transmission band. The measurement results show that a ladder-lattice BAW filter is not sufficient to fulfill this specification. A BAW-based duplexer, providing further isolation between both signal paths

**Figure 33.** Signal generator and BAW filter architecture (left), photograph of the DS modulator and

**Figure 34.** Delta-sigma digital generator output spectrum before (red) and after (black) BAW filtering

BAW assembly

This low value is due to the use of the first image and the loss in the series resistors.

is consequently mandatory and will be described in the following section.

The BAW duplexer has been designed using a mix of the exposed BAW filter methodology and co-simulation with electromagnetic tools [27], [28], [29]. The topology of this duplexer is shown in Figure 35.

**Figure 35.** BAW duplexer topology

The BAW duplexer was made with two BAW filters and a glass substrate containing high quality passive elements (IPD from ST Microelectronics [30]). Each BAW filter occupies 1mm². The TX and RX filters were flip-chipped on the 3.9x3.9mm² substrate, as shown in Figure 36.

**Figure 36.** Left: Photograph of bumps and micropackaged TX filter. Center and right: On probe measurement of the BAW duplexer

The on-probe measurement of the duplexer is performed in two steps, for TX and RX paths respectively. This is due to calibration restrictions: two ports are 50 single ended and the third one is a 100 differential port, and on probe calibration of the vectorial analyzer is not guaranteed in these conditions. Figure 37 shows the comparison between probe measurement and backward simulation (taking into account of shifted components compared to expected elements in initial simulation). It shows globally a good fitting. The Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 349

only strong difference is the TX isolation at the RX frequency, which is due to grounding effects; it disappears when the ground is bounded. It should be noticed that because of the one-path measurement conditions, the other path is open, leading to a strong mismatch in the adjacent band. Moreover, the unexpected low coupling factor of the resonators and the

When mounted on board as shown in Figure 38, the BAW duplexer exhibits approximately the same performances as on probe, with a better out-of-band rejection. The RX rejection in the TX path is better than 40dB, the TX rejection in the RX path is better than 45dB, the TX insertion loss evolves from 2.6dB to 4.6dB in the upper border of the band, and the RX

especially balun variations explain the relatively low performances of this duplexer.

insertion loss falls from 4.5 to 6dB mainly due to the balun mismatch.

**Figure 38.** On board mounting of the BAW duplexer.

the BAW duplexer to get the complete transmitter.

output of the PA module.

*5.2.3. Complete transmitter test bench and measurement results.* 

Figures 39 and 40 present the measurement setup. Two test benches have been implemented: one with the RF signal generator feeding the transmit BAW filter and the PA module, and the second one is made of the latter test bench which has been completed by

At the differential output of the RF signal generator board, the BiCMOS7RF differential input – single ended output PA is connected. Isolators are placed on each signal path between the two boards. The PA exhibits a peak power gain of 13dB at 1.7GHz and almost 500MHz of -3dB bandwidth with an output compression point of 27.5dBm showing that this stage will not contribute to non-linearity with output power below 20dBm. The power gain at 1.95GHz is 10.3dB. Figure 41 presents the measured output spectrum for the signal at the

**Figure 37.** On probe measurement (blue) and simulation (BAW) – (a) TX path, (b): RX Path

only strong difference is the TX isolation at the RX frequency, which is due to grounding effects; it disappears when the ground is bounded. It should be noticed that because of the one-path measurement conditions, the other path is open, leading to a strong mismatch in the adjacent band. Moreover, the unexpected low coupling factor of the resonators and the especially balun variations explain the relatively low performances of this duplexer.

When mounted on board as shown in Figure 38, the BAW duplexer exhibits approximately the same performances as on probe, with a better out-of-band rejection. The RX rejection in the TX path is better than 40dB, the TX rejection in the RX path is better than 45dB, the TX insertion loss evolves from 2.6dB to 4.6dB in the upper border of the band, and the RX insertion loss falls from 4.5 to 6dB mainly due to the balun mismatch.

**Figure 38.** On board mounting of the BAW duplexer.

348 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

The on-probe measurement of the duplexer is performed in two steps, for TX and RX paths respectively. This is due to calibration restrictions: two ports are 50 single ended and the third one is a 100 differential port, and on probe calibration of the vectorial analyzer is not guaranteed in these conditions. Figure 37 shows the comparison between probe measurement and backward simulation (taking into account of shifted components compared to expected elements in initial simulation). It shows globally a good fitting. The

**Figure 37.** On probe measurement (blue) and simulation (BAW) – (a) TX path, (b): RX Path

## *5.2.3. Complete transmitter test bench and measurement results.*

Figures 39 and 40 present the measurement setup. Two test benches have been implemented: one with the RF signal generator feeding the transmit BAW filter and the PA module, and the second one is made of the latter test bench which has been completed by the BAW duplexer to get the complete transmitter.

At the differential output of the RF signal generator board, the BiCMOS7RF differential input – single ended output PA is connected. Isolators are placed on each signal path between the two boards. The PA exhibits a peak power gain of 13dB at 1.7GHz and almost 500MHz of -3dB bandwidth with an output compression point of 27.5dBm showing that this stage will not contribute to non-linearity with output power below 20dBm. The power gain at 1.95GHz is 10.3dB. Figure 41 presents the measured output spectrum for the signal at the output of the PA module.

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 351

One can observe that as the signal level at the input of the PA is low, there is no "visible" distortion on the signal (when looking in-band), and the ACLR level is preserved. When investigating a wider frequency window, we see the Sigma-Delta like lower side skirts regrowing in the band due to the wide-band amplification of the B7RF PA. Finally, a 42dB ACLR at 5 and 10MHz from the carrier for a -17.3dBm integrated power is obtained. EVM

At the single-ended output of the B7RF power amplifier, an isolator and the BAW duplexer are connected on the Tx input. The output chain measurements are done on the antenna pin of the duplexer demo-board, while the differential output toward the Rx path is shunted to 50 Ω loads. In figure 42, the filtering role of the duplexer Tx path on the overall transmitted signal is clearly shown. The ACLR constraints are generally preserved through the complete transmission chain. Moreover, the low-side spectrum re-growth observed in figure 42 is swept over by the BAW duplexer Tx filtering path. Finally, a 41dB and 41.7dB ACLR respectively at 5 and 10MHz from the carrier for a -20.8dBm integrated power is obtained.

**Figure 42.** Measured wide-band output spectrum and EVM feature for the full transmitter (filtered ΔΣ

The complete transmitter is in line with the ACLR and EVM specifications. The transmitter was not able to fulfill the transmit power specifications due to some discrepancies on the distributed gain over the chain. Nevertheless, spurious emission requirements are almost fulfilled, thanks to the BAW high out-of-band rejection. The goal of the presented results was to demonstrate a novel type of Software Defined Radio transmitter architecture for W-CDMA (and DCS) standard(s) with BAW based filter and W-CDMA duplexer. Extra gain should be inserted into the whole transmission chain at the price of an increase in EVM and

has been measured to be 4%.

EVM has been measured to be 5%.

+ B7RF PA + BAW duplexer)

non-linearity degradation (lowering the ACLR).

**Figure 39.** Test fixture for the full transmitter

**Figure 40.** Measurement setup for filtered ΔΣ+ B7RF PA + BAW duplexer (full transmitter)

**Figure 41.** Output spectrum of the power gain stage, zoomed on the bandwidth (left) and full span (right)

One can observe that as the signal level at the input of the PA is low, there is no "visible" distortion on the signal (when looking in-band), and the ACLR level is preserved. When investigating a wider frequency window, we see the Sigma-Delta like lower side skirts regrowing in the band due to the wide-band amplification of the B7RF PA. Finally, a 42dB ACLR at 5 and 10MHz from the carrier for a -17.3dBm integrated power is obtained. EVM has been measured to be 4%.

350 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

**Figure 40.** Measurement setup for filtered ΔΣ+ B7RF PA + BAW duplexer (full transmitter)

**Figure 41.** Output spectrum of the power gain stage, zoomed on the bandwidth (left) and full span

**Figure 39.** Test fixture for the full transmitter

(right)

At the single-ended output of the B7RF power amplifier, an isolator and the BAW duplexer are connected on the Tx input. The output chain measurements are done on the antenna pin of the duplexer demo-board, while the differential output toward the Rx path is shunted to 50 Ω loads. In figure 42, the filtering role of the duplexer Tx path on the overall transmitted signal is clearly shown. The ACLR constraints are generally preserved through the complete transmission chain. Moreover, the low-side spectrum re-growth observed in figure 42 is swept over by the BAW duplexer Tx filtering path. Finally, a 41dB and 41.7dB ACLR respectively at 5 and 10MHz from the carrier for a -20.8dBm integrated power is obtained. EVM has been measured to be 5%.

**Figure 42.** Measured wide-band output spectrum and EVM feature for the full transmitter (filtered ΔΣ + B7RF PA + BAW duplexer)

The complete transmitter is in line with the ACLR and EVM specifications. The transmitter was not able to fulfill the transmit power specifications due to some discrepancies on the distributed gain over the chain. Nevertheless, spurious emission requirements are almost fulfilled, thanks to the BAW high out-of-band rejection. The goal of the presented results was to demonstrate a novel type of Software Defined Radio transmitter architecture for W-CDMA (and DCS) standard(s) with BAW based filter and W-CDMA duplexer. Extra gain should be inserted into the whole transmission chain at the price of an increase in EVM and non-linearity degradation (lowering the ACLR).

## **6. Conclusion**

In this chapter several methods for BAW devices simulation have been presented and investigated. Regarding the efficiency and the rapidity of the needed computation, designers could choose the 1D, the co-simulation piezoelectric equations and 2.5 D simulations or the full 3D Finite Element Method. The co-simulation equations and 2.5 D seems to be the best compromise between simulation time and quality of results. The detailed methods have been validated on several types of filters (Ladder, Lattice and mixed Ladder-Lattice). Optimization method and tools for filters have also been proposed to compute high quality filters (low insertion losses, high rejection, high pole number…).

Modeling and Design of BAW Resonators and Filters for Integration in a UMTS Transmitter 353

[4] Lakin, K. M., Kline, G. R., and Mccarron, K. T., "Development of Miniature Filters for Wireless Application," *IEEE Transaction on Microwave Theory and Techniques*, Vol.43,

[5] K. M. Lakin and K. G. Lakin, « Numerical Analysis of Thin Film BAW Resonators »,

[6] R. Thalhammer, R. Aigner, « Energy loss mechanisms in SMR–type BAW devices »,

[7] J. F. Rosenbaum "Bulk Acoustic Wave Theory and Devices", Boston: Artech House,

[8] L. Catherinot, S. Giraud, M. Chatras, S. Bila, D. Cros, T. Baron, S. Ballandras, P. Monfraix, L. Estagerie "A general procedure for the desing of BAW filters", International Journal of RF and Microwave Computer-Aided Engineering, September

[10] Lanz, R.; Muralt, P. "Solidly mounted BAW filters for 8 GHz based on AlN thin films",

[11] C. Cibert, C. Champeau, M. Chatras, D. Cros and A. Catherinot « Pulsed laser deposition of aluminum nitride thin films for FBAR application » Applied Surface

[12] H.P.Loebl, M.Klee, C.Metzmacher, W.Brand, R.Milson And P.Lok "Piezoelectric Thin AlN Film for Bulk Acoustic Wave (BAW) Resonators" Materials Chemistry and Physics,

[13] Ylilammi, J.Ella, M.Partanen and J.Kaitila "Thin Film Bulk Acoustic Wave Filter" IEEE Transactions on Ultrasonics, Ferroelectrics, and Frequency Control, vol. 49, No. 4, pp.

[16] E. Iborra, M. Clement, J. Olivares, S. Gonzalez-Castilla, J. Sangrador, N. Rimmer, A. Rastogi, B. Ivira, and A. Reinhardt, "BAW resonators based on AlN with Ir electrodes for digital wireless transmissions." 2008 IEEE Ultrason. Symp. Proc., (2008) pp. 2189-

[17] B. Ivira, P. Benech, R. Fillit, F. Ndagijimana, P. Ancey, and G. Parat, "Self-Heating Study of Bulk Acoustic Wave Resonators Under High RF Power" IEEE Trans. Ultrason. Ferr.

[18] A. Reinhardt, F. de Crécy, M. Aïd, S. Giraud, S. Bila, and E. Iborra, "Design of Computer Experiments: A powerful tool for the numerical design of BAW filters" 2008

[19] A. Devos, E. Iborra, J. Olivares, M. Clement, A. Rastogi, and N. Rimmer, "Picosecond Ultrasonics as a Helpful Technique for Introducing a New Electrode Material in BAW Technology: The Iridium Case", 2007 IEEE Ultrason. Symp. Proc., (2007) pp. 1433-1436. [20] J. Olivares, M. Clement, E. Iborra, S. González-Castilla, N. Rimmer, and A. Rastogi, "Assessment of Aluminum Nitride Films Sputtered on Iridium Electrodes ", Ultrasonics

*Microwave Symposium Digest*, 2005 IEEE MTT-S International 12-17 June 2005

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2933-2939, 1995

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Symposium 2007, pp 1401-1404.

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2192.

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[9] http://www.home.agilent.com

1988

BAW based filters and duplexers have then been presented and integrated in a complete UMTS transmitter highlighting the advantages of BAW devices.

Other piezoelectric materials are investigated and appear as promising solutions for realizing wide-band filters or sensors [31], [32].

## **Author details**

Matthieu Chatras, Stéphane Bila, Sylvain Giraud, Lise Catherinot, Ji Fan, Dominique Cros and Michel Aubourg *XLIM, UMR CNRS 7262, University of Limoges, Limoges, France* 

Axel Flament , Antoine Frappé , Bruno Stefanelli and Andreas Kaiser *IEMN, UMR CNRS 8520, Villeneuve d'Ascq, France* 

Andreia Cathelin *STMicroelectronics, TR&D, Crolles, France* 

Jean Baptiste David and Alexandre Reinhardt *CEA-LETI, Grenoble, France* 

Laurent Leyssenne and Eric Kerhervé *IMS, UMR CNRS 5818, Université de Bordeaux, Talence, France* 

## **7. References**


UMTS transmitter highlighting the advantages of BAW devices.

Matthieu Chatras, Stéphane Bila, Sylvain Giraud, Lise Catherinot,

Axel Flament , Antoine Frappé , Bruno Stefanelli and Andreas Kaiser

*XLIM, UMR CNRS 7262, University of Limoges, Limoges, France* 

*IMS, UMR CNRS 5818, Université de Bordeaux, Talence, France* 

IEEE Microwave Symp. Digest, 3, pp 1517-1520

realizing wide-band filters or sensors [31], [32].

Ji Fan, Dominique Cros and Michel Aubourg

*IEMN, UMR CNRS 8520, Villeneuve d'Ascq, France* 

*STMicroelectronics, TR&D, Crolles, France* 

Laurent Leyssenne and Eric Kerhervé

47, Research Signpost, 2005.

Jean Baptiste David and Alexandre Reinhardt

In this chapter several methods for BAW devices simulation have been presented and investigated. Regarding the efficiency and the rapidity of the needed computation, designers could choose the 1D, the co-simulation piezoelectric equations and 2.5 D simulations or the full 3D Finite Element Method. The co-simulation equations and 2.5 D seems to be the best compromise between simulation time and quality of results. The detailed methods have been validated on several types of filters (Ladder, Lattice and mixed Ladder-Lattice). Optimization method and tools for filters have also been proposed to compute high quality filters (low insertion losses, high rejection, high pole number…).

BAW based filters and duplexers have then been presented and integrated in a complete

Other piezoelectric materials are investigated and appear as promising solutions for

[1] K.M.Lakin, G.R. Kline, K.T. Mccarron, "High Acoustic Resonators and Filters", 1993

[2] K. Y. Hashimoto, "RF Bulk Acoustic Wave Filters for Communications", Artech House,

[3] A. Shirakawa, JM. Pham, P. JarrY, E. Kerherve, "FBAR Filters Synthesis Method and Reconfiguration Trends", Chapter 3 of Microwave Filters and Amplifiers book, pp.19-

**6. Conclusion** 

**Author details** 

Andreia Cathelin

**7. References** 

2009.

*CEA-LETI, Grenoble, France* 

	- [21] A. Frappé, A. Flament, B. Stefanelli, A. Kaiser, A. Cathelin, "An all-digital RF signal generator using high-speed ΔΣ modulators", IEEE Journal of Solid-State Circuits, art. No. 15, Vol.44, Oct. 2009

**Chapter 15** 

© 2013 Li et al., licensee InTech. This is an open access chapter distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use,

© 2013 The Author(s). Licensee InTech. This chapter is distributed under the terms of the Creative Commons Attribution License http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution,

distribution, and reproduction in any medium, provided the original work is properly cited.

and reproduction in any medium, provided the original work is properly cited.

© 2013 Li et al., licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

**Surface Acoustic Wave Based Magnetic Sensors** 

Since the radar system was invented in 1922, the development of devices communicating by means of reflected power has experienced a continuously growing interest. In 1948, Harry Stockman published a paper [1] in which he laid the basis for the idea of radio frequency identification (RFID), and the first patent had been filed in 1973 by Charles Walton. After decades of research and commercialization, RFID products became a part of everyday life (e.g. logistics, access control, security). With the growing interest in remote and battery-free devices, researchers are pushing the boundaries of RFID technology to find solutions in new

For many sensors such as those operated in remote or harsh environments, the sensitivity is not the only evaluation criteria. The lifetime, especially of the power source, and the complexity added by wiring often demand wireless and passive operation. Batteries have limited lifetime and also add to the size and mass of the sensors. Alternatively, energy harvesting or an RF-based wireless power supply can be employed [2, 3]. The former method depends on environmental conditions such as solar radiation, temperature change, chemical reagents, vibration etc., which are often not constantly or not sufficiently available. RF power sources, on the other hand, transmit power wirelessly and with full control over

Passive and remote sensors utilizing SAW transponders are devices, which are powered by an RF source. These systems require an interrogation device that requests the sensor signal, a SAW transponder plus a sensing element and two antennas. The basic idea is that an RF signal of certain frequencies generated by the interrogator is received by the SAW transponder, which reflects back a signal modified by the sensing element. This signal contains the environmental information in an amplitude and phase change, which is converted into the physical parameters by the interrogator. In most cases, SAW sensors are coded by having different reflector designs in order to have multiple measurement capabilities from sensors located in the same interrogation area. A great amount of research

Bodong Li, Hommood Al Rowais and Jürgen Kosel

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/55220

fields like sensing applications.

amount and timing.

**1. Introduction** 


## **Surface Acoustic Wave Based Magnetic Sensors**

Bodong Li, Hommood Al Rowais and Jürgen Kosel

Additional information is available at the end of the chapter

http://dx.doi.org/10.5772/55220

## **1. Introduction**

354 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

and optimisation for UMTS application", EuMW 2009, Rome

Circuits, Vol. 38, No. 10, pp. 1635-1642, Oct. 2003.

No. 15, Vol.44, Oct. 2009

http:\\www.3gpp.org\

and Signal Processing,2010.

pp.476

pp683-686.

Francisco, May 2011

New Port(USA), june 2010.

[21] A. Frappé, A. Flament, B. Stefanelli, A. Kaiser, A. Cathelin, "An all-digital RF signal generator using high-speed ΔΣ modulators", IEEE Journal of Solid-State Circuits, art.

[22] Vankka, J. Sommarek, J. Ketola, I. Teikari, M. Kosunen and K. Halonen, "A Digital Quadrature Modulator with on-chip D/A Converter," IEEE Journal of Solid-State

[23] S. Giraud, S. Bila, M. Chatras, D. Cros, M. Aubourg, "Bulk acoustic wave filter synthesis

[24] 3GPP UE Radio Transmission and Reception (FDD) TS 25.101. Available at

[25] A. Shirakawa, P. Jarry, J.-M. Pham, E. Kerhervé, F. Dumont, J.-B. David, A. Cathelin, "Ladder-Lattice Bulk Acoustic Wave Filters: Concepts, Design, and Implementation", International Journal of RF and Microwave Computer-Aided Engineering, 5 June 2008,

[26] A. Flament, S. Giraud, S. Bila, M. Chatras, A. Frappe, B. Stefanelli, A. Kaiser, A. Cathelin, *"Complete BAW filtered CMOS 90nm digital RF signal generator",* Joint IEEE

[27] E. Kerhervé, J.B. David, A. Shirakawa, M. El Hassan, K. Baraka, P. Vincent, A. Cathelin, "SMR-BAW duplexer for W-CDMA application", Journal of Analog Integrated Circuits

[28] A. Shirakawa, P. Jarry, J.M. Pham, E. Kerherve, F. Dumont, J.B. David, A. Cathelin, "Ladder-Lattice BAW Filters: Concepts, Design and Implementation" , International

[29] P.Bradly, R.Ruby, J.Larsoniii, Y.Oshmyansky and D. Figueredo "A Film Bulk Acoustic Resonator (FBAR) Duplexer for USPCS Handset Applications" IEEE International

[30] Calvez, C.; Person, C.; Coupez, J.; Gallée, F.; Gianesello, F.; Gloria, D.; Belot, D.; Ezzeddine, H. "Packaged hybrid Si-IPD anrenna for 60 GHz applications", EuMW 2010,

[31] M. Chatras, L. Catherinot, S. Bila, D. Cros, S. Ballandras, T. Baron, P. Monfraix, L. Estagerie "Large Band Pass BAW Filter for Space Applications"IEEE, IFCS, San

[32] T. Baron, J. Masson, D. Gachon, J.P. Romand, S. ALzuaga, L. Catherinot, M. Chatras, S. Ballandras. "A Pressure Sensor based on HBAR micromachined structure" IEEE IFCS,

North-East Workshop on Circuits and Systems and TAISA Conference, 2009

Journal of RF & Microwave Computer Aided Engineering (RFMiCAE), 2008

Microwave Symposium Digest 2001, Vol.1, P367-370

Since the radar system was invented in 1922, the development of devices communicating by means of reflected power has experienced a continuously growing interest. In 1948, Harry Stockman published a paper [1] in which he laid the basis for the idea of radio frequency identification (RFID), and the first patent had been filed in 1973 by Charles Walton. After decades of research and commercialization, RFID products became a part of everyday life (e.g. logistics, access control, security). With the growing interest in remote and battery-free devices, researchers are pushing the boundaries of RFID technology to find solutions in new fields like sensing applications.

For many sensors such as those operated in remote or harsh environments, the sensitivity is not the only evaluation criteria. The lifetime, especially of the power source, and the complexity added by wiring often demand wireless and passive operation. Batteries have limited lifetime and also add to the size and mass of the sensors. Alternatively, energy harvesting or an RF-based wireless power supply can be employed [2, 3]. The former method depends on environmental conditions such as solar radiation, temperature change, chemical reagents, vibration etc., which are often not constantly or not sufficiently available. RF power sources, on the other hand, transmit power wirelessly and with full control over amount and timing.

Passive and remote sensors utilizing SAW transponders are devices, which are powered by an RF source. These systems require an interrogation device that requests the sensor signal, a SAW transponder plus a sensing element and two antennas. The basic idea is that an RF signal of certain frequencies generated by the interrogator is received by the SAW transponder, which reflects back a signal modified by the sensing element. This signal contains the environmental information in an amplitude and phase change, which is converted into the physical parameters by the interrogator. In most cases, SAW sensors are coded by having different reflector designs in order to have multiple measurement capabilities from sensors located in the same interrogation area. A great amount of research

has been carried out in the past decades in this field, and, as a result, different wireless SAW sensors have been developed to measure a variety of physical and chemical parameters including temperature, stress, torque, pressure, humidity, magnetic field, chemical vapor etc. [4-9]. Several devices are already commercialized [10-12].

Surface Acoustic Wave Based Magnetic Sensors 357

. (1)

�����������������. (2)

structures arranged in an interdigital fashion, whereby the distance between two fingers of a comb defines the periodicity (*p*) (Fig. 1). Upon application of a voltage, charges accumulate at the fingers of the IDT depending on the capacitance of the structure. The resulting electric field generates stress in the substrate due to the piezoelectric effect. If an ac input voltage is applied, the continuously changing polarity of the charges will excite an SAW (Rayleigh wave) traveling through the substrate. At the operating (resonant) frequency of the SAW device, the value of *p* equals the wavelength of the SAW, and the SAW amplitude shows a

**SAW Resonant Frequency:** The resonant motion of an acoustic resonator is caused by the coupling between the transducer (IDT) and the acoustic medium. The resonant frequency, or operating frequency, is determined by the periodicity and the acoustic wave velocity (*v)*

> � = � �

The value of *v* mainly depends on the substrate's material. A typical SAW velocity for piezoelectric materials is several thousand meters per second. Due to the intrinsic anisotropy of piezoelectric materials, *v* is dependent on the direction of propagation. Since different acoustic modes have different wave velocities, a device can resonate at different frequencies. The SAWs are Rayleigh waves, which have a longitudinal and a vertical shear component that can couple with any media in contact with the surface. This coupling strongly affects the amplitude and velocity of the wave allowing SAW sensors to directly

**Electro-Mechanical Coupling Coefficient:** The electro-mechanical coupling coefficient (*κ*) defines the conversion efficiency of the piezoelectric material between the electrical and

��� <sup>=</sup> ������������

maximum value due to constructive superposition.

**Figure 1.** Schematic of a SAW device.

**2.1. Basics of SAW devices** 

sense, e.g., mass loads.

mechanical energies, determined by

SAW-based magnetic sensors have, so far, not been studied in detail. Magnetic sensors are one of the most pervasive kinds of sensors for a large number of applications and are employed in different fields like automotive, biomedical or consumer electronics. Integrating magnetic sensors with SAW transponders enables remote and passive operation, thereby, opens a door for further applications. Early in 1975, a magnetically tuned SAW phase shifter was proposed by Ganguly et al [13]. A thin film of magnetostrictive material was fabricated on the delay line of a SAW device. A phase shift was observed due to the dependence of the wave propagation velocity on the external magnetic field. Recently, a new concept of a magnetic sensor based on a SAW resonator has been published [8]. A magnetostrictive material was used to fabricate the interdigital transducers (IDT) of the SAW device. The resonant frequency of the device changes with an external magnetic field. A different idea was put forth by Hauser and Steindl [14-16] combining a SAW transducer with a giant magnetoimpedance (GMI) microwire sensor. The GMI sensor has a magnetic sensitivity, at least, one order higher than contemporary giant magnetoresistance (GMR) sensors and can be used to measure very low magnetic fields such as those generated by the human heart or muscles. A GMI sensor is operated by an ac current and the impedance changes upon changes of a magnetic field. This makes it a suitable load for a SAW transponder, which converts this impedance change into a magnitude and phase change of the reflected acoustic waves. In order to reduce the size and improve the level of integration of the senor, a new design of an integrated SAW transponder and thin film GMI sensor has been proposed and developed recently by the authors [17, 18]. The SAW transponder and GMI thin film were integrated on the same chip using standard micro-fabrication technology suitable for mass fabrication.

The ideal SAW-based magnetic sensor is small and highly integrated, inexpensive, passive, remotely controlled and have a high magnetic sensitivity together with a large linear range. With regard to these criterions, an SAW-GMI sensor is a very promising candidate.

SAW-based magnetic sensors have been studied for several years. However, this topic has yet not been comprehensively summarized, and the aim of this chapter is to provide a systematic review of the past research as well as the latest results. The performance of the devices crucially depends on different design parameters in a complex fashion. This will be shown by a detailed description and analysis for a device consisting of a SAW transponder and GMI thin film sensor.

## **2. SAW based passive sensors**

A basic SAW device consists of an input interdigital transducer (IDT) and an output (or reflector) IDT, which are fabricated on a piezoelectric substrate. The area between the input IDT and output IDT is called the delay line. The IDT is made of two metallic, comb-like structures arranged in an interdigital fashion, whereby the distance between two fingers of a comb defines the periodicity (*p*) (Fig. 1). Upon application of a voltage, charges accumulate at the fingers of the IDT depending on the capacitance of the structure. The resulting electric field generates stress in the substrate due to the piezoelectric effect. If an ac input voltage is applied, the continuously changing polarity of the charges will excite an SAW (Rayleigh wave) traveling through the substrate. At the operating (resonant) frequency of the SAW device, the value of *p* equals the wavelength of the SAW, and the SAW amplitude shows a maximum value due to constructive superposition.

**Figure 1.** Schematic of a SAW device.

## **2.1. Basics of SAW devices**

356 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

etc. [4-9]. Several devices are already commercialized [10-12].

technology suitable for mass fabrication.

and GMI thin film sensor.

**2. SAW based passive sensors** 

has been carried out in the past decades in this field, and, as a result, different wireless SAW sensors have been developed to measure a variety of physical and chemical parameters including temperature, stress, torque, pressure, humidity, magnetic field, chemical vapor

SAW-based magnetic sensors have, so far, not been studied in detail. Magnetic sensors are one of the most pervasive kinds of sensors for a large number of applications and are employed in different fields like automotive, biomedical or consumer electronics. Integrating magnetic sensors with SAW transponders enables remote and passive operation, thereby, opens a door for further applications. Early in 1975, a magnetically tuned SAW phase shifter was proposed by Ganguly et al [13]. A thin film of magnetostrictive material was fabricated on the delay line of a SAW device. A phase shift was observed due to the dependence of the wave propagation velocity on the external magnetic field. Recently, a new concept of a magnetic sensor based on a SAW resonator has been published [8]. A magnetostrictive material was used to fabricate the interdigital transducers (IDT) of the SAW device. The resonant frequency of the device changes with an external magnetic field. A different idea was put forth by Hauser and Steindl [14-16] combining a SAW transducer with a giant magnetoimpedance (GMI) microwire sensor. The GMI sensor has a magnetic sensitivity, at least, one order higher than contemporary giant magnetoresistance (GMR) sensors and can be used to measure very low magnetic fields such as those generated by the human heart or muscles. A GMI sensor is operated by an ac current and the impedance changes upon changes of a magnetic field. This makes it a suitable load for a SAW transponder, which converts this impedance change into a magnitude and phase change of the reflected acoustic waves. In order to reduce the size and improve the level of integration of the senor, a new design of an integrated SAW transponder and thin film GMI sensor has been proposed and developed recently by the authors [17, 18]. The SAW transponder and GMI thin film were integrated on the same chip using standard micro-fabrication

The ideal SAW-based magnetic sensor is small and highly integrated, inexpensive, passive, remotely controlled and have a high magnetic sensitivity together with a large linear range.

SAW-based magnetic sensors have been studied for several years. However, this topic has yet not been comprehensively summarized, and the aim of this chapter is to provide a systematic review of the past research as well as the latest results. The performance of the devices crucially depends on different design parameters in a complex fashion. This will be shown by a detailed description and analysis for a device consisting of a SAW transponder

A basic SAW device consists of an input interdigital transducer (IDT) and an output (or reflector) IDT, which are fabricated on a piezoelectric substrate. The area between the input IDT and output IDT is called the delay line. The IDT is made of two metallic, comb-like

With regard to these criterions, an SAW-GMI sensor is a very promising candidate.

**SAW Resonant Frequency:** The resonant motion of an acoustic resonator is caused by the coupling between the transducer (IDT) and the acoustic medium. The resonant frequency, or operating frequency, is determined by the periodicity and the acoustic wave velocity (*v)*

$$f = \frac{\nu}{p}.\tag{1}$$

The value of *v* mainly depends on the substrate's material. A typical SAW velocity for piezoelectric materials is several thousand meters per second. Due to the intrinsic anisotropy of piezoelectric materials, *v* is dependent on the direction of propagation. Since different acoustic modes have different wave velocities, a device can resonate at different frequencies. The SAWs are Rayleigh waves, which have a longitudinal and a vertical shear component that can couple with any media in contact with the surface. This coupling strongly affects the amplitude and velocity of the wave allowing SAW sensors to directly sense, e.g., mass loads.

**Electro-Mechanical Coupling Coefficient:** The electro-mechanical coupling coefficient (*κ*) defines the conversion efficiency of the piezoelectric material between the electrical and mechanical energies, determined by

$$
\kappa^{-2} = \frac{input\ energy}{convected\ energy}.\tag{2}
$$

A high coupling coefficient reduces the insertion loss caused by the energy conversion, which results in smaller energy consumption as well as larger effective readout distance of a SAW-based wireless sensor.

**SAW Delay Line:** The SAW delay line refers to the area between the input IDT and output IDT on the substrate (Fig. 1). It creates a time delay between the input signal and the output signal depending on the SAW velocity and the length of the delay line. Due to this feature, SAW devices are widely used in RF electronics. It is also used in sensing applications, where the measurand causes, e.g., a change in the SAW.

**Temperature Coefficient of Delay (TCD):** The TCD reflects the temperature dependence of the time delay and is connected with the thermal expansion coefficient (*α*) and the temperature coefficient of the phase velocity (TCV) by

$$T\!\!\!CD = a - TC\!\!\!V\!\.\,.\tag{3}$$

Surface Acoustic Wave Based Magnetic Sensors 359

Access Paper

None [28]

[25]

[8]

[16,17]

[31]

IDT is connected to a sensor, which changes the IDT's electrical characteristics as a function of the measurand. An example for a load sensor is a pair of conducting rods placed in the earth with a certain distance from each other. As the water level changes, the resistance between the rods changes, which can be detected as a magnitude and phase change of the signal reflected from the load IDT [27]. Another example for a load is a giant magnetoimpedance sensor [16, 17]. A change in the magnetic field yields a change in the

sensor's impedance. Consequently, changes the reflectivity of the output IDT.

Year Intrinsic/ Extrinsic

Pressure Yes 2001 Extrinsic Loaded

Sound No 2005 Extrinsic Loaded

**Table 1.** SAW-based passive sensors.

applicability.

Sensor Type Commerci

alization

Some SAW sensors, their classification and method of detection are presented in Table 1.

Temperature Yes 1990 Intrinsic Resonator Frequency None [23] 2003 Intrinsic Delay line Phase

Transponder

Bio/Chem No/Yes 2006 Extrinsic Resonator Frequency Thin film [29]

Transponder

Transponder

Transponder

Torque Yes 1996 Intrinsic Delay line Phase None [32]

Magnetic sensors are one of the most versatile sensors employed not only for the task of measuring magnetic fields but for a large number of different applications, thereby detecting the measurand indirectly, e.g., via a change of material parameters in construction monitoring or a change of distance in position monitoring. A passive and remote operation of magnetic sensors can be advantageous in many cases and considerably increase their

A SAW-based passive magnetic sensor can be realized either by adding an additional material layer, which is sensitive to magnetic fields, or by loading the output IDT with a magnetic sensor. In the first case, the magnetic layer changes the delay line or the resonant frequency of the SAW device. While in the second case, the sensor changes the reflection

Magnetic No 1975 Extrinsic Delay line Phase Thin film [13]

2001 Extrinsic Loaded

2006/11 Extrinsic Loaded

**3. Passive and remote SAW-based magnetic sensors** 

Design Detection

Method

velocity

2007 Intrinsic Delay line Phase None [24]

2011 Extrinsic Delay line Phase Thin film [26]

2011 Extrinsic Resonator Frequency Magnetostrictive

Amplitude/ Phase

Phase Capacitive

Amplitude GMI wire/ thin

Phase Capacitive

pressure sensor

IDTs

film

pressure sensor

Conducting rods [30]

The temperature dependence of the time delay is the basis of SAW temperature sensors, where higher TCD values yield higher sensitivity. However, for other SAW devices, the influence of the TCD on the time delay is undesirable and has to be minimized or eliminated. For this purpose, temperature compensated cuts of the crystalline substrates are employed, where the TCD is minimized over certain temperature ranges [19-20]. Piezoelectric bi-layers are another concept that has been utilized in order to compensate the TCD in sensing applications [21, 22].

## **2.2. Basic design concepts of passive SAW sensors**

Passive SAW sensors typically operate as resonators, delay lines or loaded transponders. In case of resonators, the reflection of the interrogation signal from the SAW device is a function of the SAW device's resonant frequency, which depends on the measurand. In case of delay lines, the request signal is separated from the response signal by a time difference, whereby this time difference depends on the measurand. Similarly, the request signal and response signal are separated by a time difference in case of a loaded transponder. However, the time difference is constant and the measurand affects the signal amplitude. Intrinsic SAW sensors utilize a change of the substrate's properties. For example, intrinsic temperature sensors were realized by detecting the resonant frequency or phase change of the SAW in materials with large TCD [23]. Intrinsic stress sensors utilize the length change of the delay line caused by mechanical strain applied to the substrate. The stress can be evaluated by measuring the SAW phase shift [24]. Extrinsic SAW sensors can be realized by integrating a SAW device and an additional sensing element. A common extrinsic sensor concept utilizes selective thin films on top of the delay line leading to a change in mass by the measurand [13, 25]. This can be, for example, a thin film with high CO2 solubility and selectivity [26]. As CO2 dissolves into the film, the additional mass load causes a detectable phase shift in the SAW. Another extrinsic concept utilizes a sensitive IDT. For example, in case of a magnetostrictive IDT, a magnetic field applied to the sensor causes a change of the resonant frequency [8]. A loaded transponder is another extrinsic design, where the output IDT is connected to a sensor, which changes the IDT's electrical characteristics as a function of the measurand. An example for a load sensor is a pair of conducting rods placed in the earth with a certain distance from each other. As the water level changes, the resistance between the rods changes, which can be detected as a magnitude and phase change of the signal reflected from the load IDT [27]. Another example for a load is a giant magnetoimpedance sensor [16, 17]. A change in the magnetic field yields a change in the sensor's impedance. Consequently, changes the reflectivity of the output IDT.


Some SAW sensors, their classification and method of detection are presented in Table 1.

**Table 1.** SAW-based passive sensors.

358 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

SAW-based wireless sensor.

the measurand causes, e.g., a change in the SAW.

TCD in sensing applications [21, 22].

temperature coefficient of the phase velocity (TCV) by

**2.2. Basic design concepts of passive SAW sensors** 

A high coupling coefficient reduces the insertion loss caused by the energy conversion, which results in smaller energy consumption as well as larger effective readout distance of a

**SAW Delay Line:** The SAW delay line refers to the area between the input IDT and output IDT on the substrate (Fig. 1). It creates a time delay between the input signal and the output signal depending on the SAW velocity and the length of the delay line. Due to this feature, SAW devices are widely used in RF electronics. It is also used in sensing applications, where

**Temperature Coefficient of Delay (TCD):** The TCD reflects the temperature dependence of the time delay and is connected with the thermal expansion coefficient (*α*) and the

The temperature dependence of the time delay is the basis of SAW temperature sensors, where higher TCD values yield higher sensitivity. However, for other SAW devices, the influence of the TCD on the time delay is undesirable and has to be minimized or eliminated. For this purpose, temperature compensated cuts of the crystalline substrates are employed, where the TCD is minimized over certain temperature ranges [19-20]. Piezoelectric bi-layers are another concept that has been utilized in order to compensate the

Passive SAW sensors typically operate as resonators, delay lines or loaded transponders. In case of resonators, the reflection of the interrogation signal from the SAW device is a function of the SAW device's resonant frequency, which depends on the measurand. In case of delay lines, the request signal is separated from the response signal by a time difference, whereby this time difference depends on the measurand. Similarly, the request signal and response signal are separated by a time difference in case of a loaded transponder. However, the time difference is constant and the measurand affects the signal amplitude. Intrinsic SAW sensors utilize a change of the substrate's properties. For example, intrinsic temperature sensors were realized by detecting the resonant frequency or phase change of the SAW in materials with large TCD [23]. Intrinsic stress sensors utilize the length change of the delay line caused by mechanical strain applied to the substrate. The stress can be evaluated by measuring the SAW phase shift [24]. Extrinsic SAW sensors can be realized by integrating a SAW device and an additional sensing element. A common extrinsic sensor concept utilizes selective thin films on top of the delay line leading to a change in mass by the measurand [13, 25]. This can be, for example, a thin film with high CO2 solubility and selectivity [26]. As CO2 dissolves into the film, the additional mass load causes a detectable phase shift in the SAW. Another extrinsic concept utilizes a sensitive IDT. For example, in case of a magnetostrictive IDT, a magnetic field applied to the sensor causes a change of the resonant frequency [8]. A loaded transponder is another extrinsic design, where the output

(3) . ܸܥܶ െ ߙ ൌ ܦܥܶ

## **3. Passive and remote SAW-based magnetic sensors**

Magnetic sensors are one of the most versatile sensors employed not only for the task of measuring magnetic fields but for a large number of different applications, thereby detecting the measurand indirectly, e.g., via a change of material parameters in construction monitoring or a change of distance in position monitoring. A passive and remote operation of magnetic sensors can be advantageous in many cases and considerably increase their applicability.

A SAW-based passive magnetic sensor can be realized either by adding an additional material layer, which is sensitive to magnetic fields, or by loading the output IDT with a magnetic sensor. In the first case, the magnetic layer changes the delay line or the resonant frequency of the SAW device. While in the second case, the sensor changes the reflection signal of the output IDT. Since SAW devices are operated by RF power, the sensor element has to work at the operation frequency of the SAW device. Among the available magnetic sensors, GMI sensors are the most suitable candidates as they have a high magnetic sensitivity as well as a high operating frequency.

Surface Acoustic Wave Based Magnetic Sensors 361

stress will be induced to the substrate by the IDT change causing a change in the resonant frequency. This sensor showed a frequency change of 200 ppm for a magnetic field of 100

**Figure 3.** Schematic of a magnetic sensor device using a magnetostrictive IDT on a SAW substrate.

A magnetically tuned SAW phase shifter is a one-port SAW structure with a magnetic sensing functionality achieved through a delay line sensitive to magnetic fields. This idea was first introduced by Ganguly et al in 1975 [13]. In their device, the acoustic velocity is varied by an external magnetic field. This functionality is facilitated by a magnetostrictive thin film deposited on top of the delay line (Fig. 4). The propagation velocity of the SAW in the film region depends on the magnetic field. Hence, there is a correlation between the time

Later, research efforts focused on different magnetostrictive materials and measurement

A SAW-based, magnetic and passive sensors comprises a two-port SAW transponder and a magnetic sensor acting as a load at the output IDT. Among the available magnetic field sensors, giant magnetoimpedance (GMI) sensors offer favorable characteristics like high sensitivity to magnetic fields and high operation frequency (compatible with SAW

mT applied perpendicularly to the direction of SAW propagation.

*3.1.2. Passive phase shifter* 

**3.2. SAW-GMI sensors** 

shift of the reflected signal and the magnetic field.

**Figure 4.** Schematic of a magnetically tuned SAW wave phase shifter.

methods [9, 35, 36], and a magnetic sensitivity of 10-4 %/kOe was achieved.

## **3.1. Magnetostrictive SAW devices**

Magnetostriction defines the relationship between the strain and the magnetization states of a material. It is an important property of ferromagnetic materials and was first observed by James Joule in 1842 in nickel samples. For a positive/negative magnetostrictive material, an applied magnetic field causes the material to expand/shrink in the field direction. Inversely, when a stress is applied to the magnetostrictive material, its magnetic anisotropy will change accordingly.

A magnetostrictive-piezoelectric resonator consists of amorphous magnetostrictive material layers as the electrodes sandwiching a piezoelectric core (Fig. 2). An ac signal applied to the electrodes causes the quartz layer to oscillate. The resonant frequency of this oscillation depends on the thickness of the piezoelectric material, the crystal orientation, temperature and mechanical stress, etc.

**Figure 2.** Structure of a composite magnetostrictive/piezoelectric resonator. The magnetic anisotropy is perpendicular to the external magnetic field *Hdc*.

When a magnetic field *Hdc* is applied, the length change induced in the magnetostrictive film exerts stress to the piezoelectric material and, consequently, shifts the resonant frequency of the device. Utilizing this concept, a magnetic sensitivity high enough to detect the terrestrial field has been achieved [33]. In a similar work, a magnetostrictive-piezoelectric tri-layer structure has been embedded in a coil. The dc magnetic field sensitivity was as high as 10-8 T [34].

## *3.1.1. Passive resonator*

A SAW-based, passive resonator for magnetic field detection was developed recently by Kadota et al [8]. Nickel, which is a negative magnetostrictive material, was used to fabricate the sensing IDT on a quartz substrate (Fig. 3). Upon the application of a magnetic field, stress will be induced to the substrate by the IDT change causing a change in the resonant frequency. This sensor showed a frequency change of 200 ppm for a magnetic field of 100 mT applied perpendicularly to the direction of SAW propagation.

**Figure 3.** Schematic of a magnetic sensor device using a magnetostrictive IDT on a SAW substrate.

## *3.1.2. Passive phase shifter*

360 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

sensitivity as well as a high operating frequency.

**3.1. Magnetostrictive SAW devices** 

change accordingly.

and mechanical stress, etc.

[34].

*3.1.1. Passive resonator* 

perpendicular to the external magnetic field *Hdc*.

signal of the output IDT. Since SAW devices are operated by RF power, the sensor element has to work at the operation frequency of the SAW device. Among the available magnetic sensors, GMI sensors are the most suitable candidates as they have a high magnetic

Magnetostriction defines the relationship between the strain and the magnetization states of a material. It is an important property of ferromagnetic materials and was first observed by James Joule in 1842 in nickel samples. For a positive/negative magnetostrictive material, an applied magnetic field causes the material to expand/shrink in the field direction. Inversely, when a stress is applied to the magnetostrictive material, its magnetic anisotropy will

A magnetostrictive-piezoelectric resonator consists of amorphous magnetostrictive material layers as the electrodes sandwiching a piezoelectric core (Fig. 2). An ac signal applied to the electrodes causes the quartz layer to oscillate. The resonant frequency of this oscillation depends on the thickness of the piezoelectric material, the crystal orientation, temperature

**Figure 2.** Structure of a composite magnetostrictive/piezoelectric resonator. The magnetic anisotropy is

When a magnetic field *Hdc* is applied, the length change induced in the magnetostrictive film exerts stress to the piezoelectric material and, consequently, shifts the resonant frequency of the device. Utilizing this concept, a magnetic sensitivity high enough to detect the terrestrial field has been achieved [33]. In a similar work, a magnetostrictive-piezoelectric tri-layer structure has been embedded in a coil. The dc magnetic field sensitivity was as high as 10-8 T

A SAW-based, passive resonator for magnetic field detection was developed recently by Kadota et al [8]. Nickel, which is a negative magnetostrictive material, was used to fabricate the sensing IDT on a quartz substrate (Fig. 3). Upon the application of a magnetic field, A magnetically tuned SAW phase shifter is a one-port SAW structure with a magnetic sensing functionality achieved through a delay line sensitive to magnetic fields. This idea was first introduced by Ganguly et al in 1975 [13]. In their device, the acoustic velocity is varied by an external magnetic field. This functionality is facilitated by a magnetostrictive thin film deposited on top of the delay line (Fig. 4). The propagation velocity of the SAW in the film region depends on the magnetic field. Hence, there is a correlation between the time shift of the reflected signal and the magnetic field.

**Figure 4.** Schematic of a magnetically tuned SAW wave phase shifter.

Later, research efforts focused on different magnetostrictive materials and measurement methods [9, 35, 36], and a magnetic sensitivity of 10-4 %/kOe was achieved.

## **3.2. SAW-GMI sensors**

A SAW-based, magnetic and passive sensors comprises a two-port SAW transponder and a magnetic sensor acting as a load at the output IDT. Among the available magnetic field sensors, giant magnetoimpedance (GMI) sensors offer favorable characteristics like high sensitivity to magnetic fields and high operation frequency (compatible with SAW transponders) making them a very suitable load. SAW-GMI sensors have been fabricated by combining SAW transponders with GMI wire sensors as well as thin film GMI sensors. Both of these methods have shown a higher magnetic sensitivity than direct designs.

#### *3.2.1. Basics of GMI sensors*

The GMI effect was first observed in Co-based amorphous wires by Panina and Mohri in 1994 [37] and has since attracted strong interest due to its sensitivity enabling magnetic field measurement with a nT resolution. The GMI effect is the impedance change of an acpowered ferromagnetic conductor upon the change of a magnetic field. The relative impedance change, also called GMI ratio, is expressed as

$$\text{GMI Ratio (\%)} = 100\% \times \frac{\text{Z} \text{(\%)} - \text{Z} \text{(\%)}}{\text{Z} \text{(\%)}} \text{ or } \text{ GMI Ratio (\%)} = 100\% \times \frac{\text{Z} \text{(\%)} - \text{Z} \text{(\%)} \text{ax}}{\text{Z} \text{(\%)}},\tag{4}$$

where *Z(H0)* is the impedance at zero magnetic field and *Z(Hmax)* is the impedance at saturation field. Both definitions have particular aspects that should be considered. In case of the first expression, *Z(H0)* depends on the remanent state of the magnetic material while, in the second case, *Z(Hmax)* is not always achievable and equipment dependent.

The GMI effect is explained by classical electromagnetism. The change of the complex impedance mainly originates from the skin effect in conjunction with a change of the complex permeability. Analytically, the complex impedance (*Z*) of a conductor is defined by

$$Z = \frac{\upsilon\_{ac}}{\upsilon\_{ac}} = \frac{\int\_{L} \mathbf{\dot{J}}\_{x}(\mathcal{S}) dx}{\iint\_{q} \mathbf{J}\_{x} dq},\tag{5}$$

Surface Acoustic Wave Based Magnetic Sensors 363

���� , (10)

��� (11)

������� , (12)

. (14)

(13)

�� = �� � ���� <sup>−</sup> �

���� =�+�� <sup>+</sup> ��

parameter and *Heff* is the effective magnetic field expressed as [30]

sensors with different geometries [37-40].

permeability change for small magnetic fields.

ribbons, etc. [45-47]

*3.2.2. Wire GMI sensors* 

(Panina et al, 1994) [37]

where

and

field, *H<sup>a</sup>* is the anisotropy field and *A* is the exchange stiffness constant..

where *γ* is the gyromagnetic ratio, *Ms* is the saturation magnetization, *α* is the damping

where *H* is the internal magnetic field that includes the applied field and demagnetizing

By combining Equ. (5) to (11), a theoretical impedance model can be evaluated for GMI

Although the experimentally obtained GMI effect shows a large sensitivity compared to other effects exploited for magnetic sensors, the theoretically estimated values have not been achieved yet. Therefore, a lot of effort has been put into improving the magnetic properties of GMI materials [41-44]. At the same time, GMI sensors of different structures have been developed such as glass-coated wires, thin films, multi layer thin films, meander structures,

As the first discovered GMI sensor structures, GMI wire sensors have been extensively studied. Based on the classical electromagnetism, the theoretical model of the GMI wire is

� =� �����������

�� = ��� ��

�� <sup>=</sup> �

*Rdc* is the dc resistance of the wire, *ζ 0*, *ζ 1* are the Bessel functions, *r* is the radius of the wire, *j* is the imaginary unit, *δm* is the penetration depth, *c* is the speed of light, *f* is the frequency of the ac current, *μø* is the circumferential magnetic anisotropy. The origin of the GMI effect lies in the dependence of *μø* on an axial magnetic field resulting in a change of *δm*. In order to obtain a high GMI ratio, the value of *δm* has to be close to the thickness of the conductor. Hence, the thinner a ferromagnetic conductor and the lower its permeability, the higher the operation frequency required. A well-defined circumferential magnetic anisotropy in combination with a soft magnetic behavior is desirable, since it will provide a large

�������∅

��

����

where *Uac* is the applied ac voltage, *I*ac is the current, *L* is the length and σ the conductivity. *S* and *q* refer to the surface and the cross section of the conductor, respectively. *Jz* is the current density in the longitudinal direction obtained by solving Maxwell's equations. In ferromagnetic materials, by neglecting displacement currents (�� = �), Maxwell's equations can be written as follows:

$$
\nabla \times \mathbf{H} = \mathbf{J},
\tag{6}
$$

$$\nabla \times \mathbf{J} = -\frac{\mu\_0}{\rho\_f} (\dot{\mathbf{H}} + \dot{\mathbf{M}}),\tag{7}$$

$$\nabla \cdot (\mathbf{H} + \mathbf{M}) = \mathbf{0},\tag{8}$$

J is the current density, H is the applied magnetic field, M is the magnetization of the ferromagnetic material, *ρ<sup>f</sup>* is the free charge density and *μ0* is the permeability of vacuum. From Equ. (6) to (8), the expression

$$
\nabla^2 \mathbf{H} - \frac{\mu\_0}{\rho\_f} \mathbf{\dot{H}} = \frac{\mu\_0}{\rho\_f} \mathbf{\dot{M}} - \nabla(\nabla \cdot \mathbf{M}),
\tag{9}
$$

can be derived. Equ. (9) can be solved using the Landau-Lifshitz equation, which relates *M*  and *H*

Surface Acoustic Wave Based Magnetic Sensors 363

$$
\dot{\mathbf{M}} = \mathbf{y}\mathbf{M} \times \mathbf{H}\_{eff} - \frac{a}{\mathbf{M}\_s} \mathbf{M} \times \dot{\mathbf{M}},\tag{10}
$$

where *γ* is the gyromagnetic ratio, *Ms* is the saturation magnetization, *α* is the damping parameter and *Heff* is the effective magnetic field expressed as [30]

$$H\_{eff} = H + H\_{\text{a}} + \frac{2A}{\mu\_0 \mathcal{M}\_{\text{s}}} \nabla^2 \mathbf{M} \tag{11}$$

where *H* is the internal magnetic field that includes the applied field and demagnetizing field, *H<sup>a</sup>* is the anisotropy field and *A* is the exchange stiffness constant..

By combining Equ. (5) to (11), a theoretical impedance model can be evaluated for GMI sensors with different geometries [37-40].

Although the experimentally obtained GMI effect shows a large sensitivity compared to other effects exploited for magnetic sensors, the theoretically estimated values have not been achieved yet. Therefore, a lot of effort has been put into improving the magnetic properties of GMI materials [41-44]. At the same time, GMI sensors of different structures have been developed such as glass-coated wires, thin films, multi layer thin films, meander structures, ribbons, etc. [45-47]

#### *3.2.2. Wire GMI sensors*

As the first discovered GMI sensor structures, GMI wire sensors have been extensively studied. Based on the classical electromagnetism, the theoretical model of the GMI wire is (Panina et al, 1994) [37]

$$Z = \frac{R\_{dc}kr\zeta\_0(kr)}{2\zeta\_1(kr)}\,\tag{12}$$

where

362 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

impedance change, also called GMI ratio, is expressed as

GMI Ratio (%) = 100% × �(�)��(��)

*3.2.1. Basics of GMI sensors* 

can be written as follows:

From Equ. (6) to (8), the expression

and *H*

of these methods have shown a higher magnetic sensitivity than direct designs.

transponders) making them a very suitable load. SAW-GMI sensors have been fabricated by combining SAW transponders with GMI wire sensors as well as thin film GMI sensors. Both

The GMI effect was first observed in Co-based amorphous wires by Panina and Mohri in 1994 [37] and has since attracted strong interest due to its sensitivity enabling magnetic field measurement with a nT resolution. The GMI effect is the impedance change of an acpowered ferromagnetic conductor upon the change of a magnetic field. The relative

where *Z(H0)* is the impedance at zero magnetic field and *Z(Hmax)* is the impedance at saturation field. Both definitions have particular aspects that should be considered. In case of the first expression, *Z(H0)* depends on the remanent state of the magnetic material while,

The GMI effect is explained by classical electromagnetism. The change of the complex impedance mainly originates from the skin effect in conjunction with a change of the complex permeability. Analytically, the complex impedance (*Z*) of a conductor is defined by

> <sup>=</sup> � � ���(�)�� � ∬ ���� �

where *Uac* is the applied ac voltage, *I*ac is the current, *L* is the length and σ the conductivity. *S* and *q* refer to the surface and the cross section of the conductor, respectively. *Jz* is the current density in the longitudinal direction obtained by solving Maxwell's equations. In ferromagnetic materials, by neglecting displacement currents (�� = �), Maxwell's equations

in the second case, *Z(Hmax)* is not always achievable and equipment dependent.

� = ��� ���

��=� ��

��� � �� �� ��

J is the current density, H is the applied magnetic field, M is the magnetization of the ferromagnetic material, *ρ<sup>f</sup>* is the free charge density and *μ0* is the permeability of vacuum.

can be derived. Equ. (9) can be solved using the Landau-Lifshitz equation, which relates *M* 

�� <sup>=</sup> �� ��

�(��) or GMI Ratio (%) = 100% × �(�)��(����)

�(����) , (4)

, (5)

��=�, (6)

� � (���) = 0, (8)

(�� � �� ), (7)

�� � �(���), (9)

$$k = \frac{1+j}{\delta\_m} \tag{13}$$

and

$$
\delta\_m = \frac{c}{\sqrt{4\pi^2 f \sigma \mu\_\oplus}}.\tag{14}
$$

*Rdc* is the dc resistance of the wire, *ζ 0*, *ζ 1* are the Bessel functions, *r* is the radius of the wire, *j* is the imaginary unit, *δm* is the penetration depth, *c* is the speed of light, *f* is the frequency of the ac current, *μø* is the circumferential magnetic anisotropy. The origin of the GMI effect lies in the dependence of *μø* on an axial magnetic field resulting in a change of *δm*. In order to obtain a high GMI ratio, the value of *δm* has to be close to the thickness of the conductor. Hence, the thinner a ferromagnetic conductor and the lower its permeability, the higher the operation frequency required. A well-defined circumferential magnetic anisotropy in combination with a soft magnetic behavior is desirable, since it will provide a large permeability change for small magnetic fields.

Different amorphous and ferromagnetic materials were used to fabricate GMI wires [48], and various fabrication methods were developed such as melt spinning, in-rotating water spinning, glass-coated melt spinning etc. [45, 49, 50]. Glass-coated micro-wires (Fig. 5) present outstanding properties in terms of the magnetic anisotropy distribution, which is reinforced by the strong mechanical stress induced by the coating. (CoxFe1-x)72.5Si12.5B15 is one of the most typical materials. By adjusting x from 0 to 1, the magnetostriction of the material changes from positive at high Fe content to negative at high Co content. Negative magnetostrictive compositions in combination with the compressive, radial stress induced by quenching and the glass coating provide the best results, since it supports a strong circumferential anisotropy.

Surface Acoustic Wave Based Magnetic Sensors 365

� ) , (15)

**Figure 7.** (a) Layout of the commercialized GMI sensor from Aichi Steel Co. (b) Noise output of the

Fig. 7 shows a GMI sensor developed by Aichi Steel Co., which has a very high sensitivity of

Magnetic ribbons discussed in this section are planar structures of rectangular shape with a thickness of a few tens of micrometers and a length and width from several millimeters to centimeters. Similar to the micro-wires, magnetostriction is utilized in order to create certain anisotropies during the ribbon's fabrication. Magnetic ribbons that exhibit a strong GMI

���

where *Rdc* is the dc resistance, *a* is the thickness of the ribbon, *k* and *δm* can be obtained from Equ. (14) with the only difference that *μø* represents the transversal permeability instead of

Again, Fe- and Co-based amorphous alloys are preferably used as the magnetic material. The standard fabrication method for the ribbons is melt spinning, where a rotating copper wheel is used to rapidly solidify the liquid alloy. This method produces magnetic ribbons with a thickness of about 25 μm and a width of several mm. With this thickness, ribbon GMI sensors operate at comparably low frequencies of hundred kHz up to a few MHz. A GMI ratio of, e.g., 640% has been obtained with a GMI ribbon made of Fe71Al2Si14B8.5Cu1Nb3.5 at 5

In theory, a single layer magnetic thin film is similar to a magnetic ribbon, and the same analytical expressions are applied for modeling the GMI effect. Practically, the main difference is the fabrication method. Thin film fabrication is a standard micro-fabrication

� ��������

effect have a high permeability as well as a transversal magnetic anisotropy.

� ����� ·

For a planar film of infinite width, the impedance is given by

GMI sensor.

1 V/μT and a noise level of 1 nT [51].

*3.2.3. Ribbon GMI sensors* 

the circumferential one [52].

*3.2.4. Thin film GMI sensors*

MHz [53].

**Figure 5.** SEM image of a glass-coated amorphous micro-wire (Courtesy of M. Vazquez, Inst. Materials Science of Madrid, CSIC).

**Figure 6.** GMI ratio of Co67Fe3.85Ni1.45B11.5Si14.5Mo1.7 glass-coated wires with different geometric ratio *ρ* (the metallic nucleus diameter to the total microwire diameter) at 10 MHz.

Wire-type GMI sensors provide the best performance in terms of the GMI ratio with values as high as 615% (Fig. 6) achieved with optimized glass coated microwires (Zhukova et al, 2002) [43]. The value of the magnetic field at which the maximum GMI ratio is obtained increases as the diameter of the magnetic nucleus decreases compared to the diameter of the glass coating. This is attributed to the different anisotropies induced by the stress from the coating. Due to the high sensitivity provided by GMI wires they have been commercialized despite the facts that fabrication is not silicon based, does not use standard microfabrication methods and, as a consequence, integration with electronics is complex.

**Figure 7.** (a) Layout of the commercialized GMI sensor from Aichi Steel Co. (b) Noise output of the GMI sensor.

Fig. 7 shows a GMI sensor developed by Aichi Steel Co., which has a very high sensitivity of 1 V/μT and a noise level of 1 nT [51].

## *3.2.3. Ribbon GMI sensors*

364 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

circumferential anisotropy.

Science of Madrid, CSIC).

Different amorphous and ferromagnetic materials were used to fabricate GMI wires [48], and various fabrication methods were developed such as melt spinning, in-rotating water spinning, glass-coated melt spinning etc. [45, 49, 50]. Glass-coated micro-wires (Fig. 5) present outstanding properties in terms of the magnetic anisotropy distribution, which is reinforced by the strong mechanical stress induced by the coating. (CoxFe1-x)72.5Si12.5B15 is one of the most typical materials. By adjusting x from 0 to 1, the magnetostriction of the material changes from positive at high Fe content to negative at high Co content. Negative magnetostrictive compositions in combination with the compressive, radial stress induced by quenching and the glass coating provide the best results, since it supports a strong

**Figure 5.** SEM image of a glass-coated amorphous micro-wire (Courtesy of M. Vazquez, Inst. Materials

**Figure 6.** GMI ratio of Co67Fe3.85Ni1.45B11.5Si14.5Mo1.7 glass-coated wires with different geometric ratio *ρ*

Wire-type GMI sensors provide the best performance in terms of the GMI ratio with values as high as 615% (Fig. 6) achieved with optimized glass coated microwires (Zhukova et al, 2002) [43]. The value of the magnetic field at which the maximum GMI ratio is obtained increases as the diameter of the magnetic nucleus decreases compared to the diameter of the glass coating. This is attributed to the different anisotropies induced by the stress from the coating. Due to the high sensitivity provided by GMI wires they have been commercialized despite the facts that fabrication is not silicon based, does not use standard microfabrication

(the metallic nucleus diameter to the total microwire diameter) at 10 MHz.

methods and, as a consequence, integration with electronics is complex.

Magnetic ribbons discussed in this section are planar structures of rectangular shape with a thickness of a few tens of micrometers and a length and width from several millimeters to centimeters. Similar to the micro-wires, magnetostriction is utilized in order to create certain anisotropies during the ribbon's fabrication. Magnetic ribbons that exhibit a strong GMI effect have a high permeability as well as a transversal magnetic anisotropy.

For a planar film of infinite width, the impedance is given by

� ����� · ��� � �������� � ) , (15)

where *Rdc* is the dc resistance, *a* is the thickness of the ribbon, *k* and *δm* can be obtained from Equ. (14) with the only difference that *μø* represents the transversal permeability instead of the circumferential one [52].

Again, Fe- and Co-based amorphous alloys are preferably used as the magnetic material. The standard fabrication method for the ribbons is melt spinning, where a rotating copper wheel is used to rapidly solidify the liquid alloy. This method produces magnetic ribbons with a thickness of about 25 μm and a width of several mm. With this thickness, ribbon GMI sensors operate at comparably low frequencies of hundred kHz up to a few MHz. A GMI ratio of, e.g., 640% has been obtained with a GMI ribbon made of Fe71Al2Si14B8.5Cu1Nb3.5 at 5 MHz [53].

## *3.2.4. Thin film GMI sensors*

In theory, a single layer magnetic thin film is similar to a magnetic ribbon, and the same analytical expressions are applied for modeling the GMI effect. Practically, the main difference is the fabrication method. Thin film fabrication is a standard micro-fabrication

technology producing a film thickness of some nanometers up to a few micrometers. Thin film GMI sensors are of great interest due to the advantages arising from the fabrication in terms of the flexibility in design and integration. They can easily be fabricated on the same substrate as the electronic circuit and other devices. In the context of passive and remote sensors, this is particularly relevant, since the GMI element can be easily integrated with an SAW device. For this reason, GMI thin film sensors will be discussed in more detail and our recent results will be presented.

Surface Acoustic Wave Based Magnetic Sensors 367

� ) , (16)

54]. An alternating current *Iac* mainly flowing through the conductor generates a transversal flux *Btran*, which magnetizes the magnetic layers. Upon the application of an external field *Hext* in longitudinal direction, the magnetization caused by *Iac* will be changed. This is equivalent to a change of the transversal permeability of the magnetic layers and is reflected

The analytical model of the impedance for a magnetic/conducting/magnetic tri-layer

� ������� � ��� ����

where *Rdc* is the dc resistance of the inner conductor, 2*d1* is the thickness of the conductor, *d2* is the thickness of the magnetic layers as shown in Fig. 8 (c) and *δc* is the penetration depth

Analytical solutions for the impedance of thin film GMI sensors can only be found for rather simple structures. In order to calculate the impedance of more complicated geometries, for example, a sandwich structure with isolation layers between the conductor and the magnetic layers [41], a meander structure multilayer [46] or to take into account edge effects,

Fig. 9 shows the comparison of the GMI ratios simulated for a single magnetic layer, a trilayer structure made of a magnetic/conducting/magnetic stack and a five-layer structure with isolation layers between the conducting and magnetic layers using the FEM. The simulated GMI sensors have a width of *w* = 50 μm and length of *l* = 200μm. The magnetic layers have a thickness of *tmag* = 1 μm and the conducting layer has a thickness of *tmet* = 4 μm. The material of the isolation layer is SiO2 with a thickness of 1μm. The conductivity of the ferromagnetic and conducting layers are 7.69×105 S/m ((CoFe)80B20) and 4.56×107 S/m (Gold), respectively. All parameters including *Ms* = 5.6×105 A/m, � = 2.2×105 m/(A·s), *α* = 0.3, *Ha*=

**Figure 9.** Simulated GMI ratios of single layer, sandwiched multilayer and isolated sandwiched

the finite element method (FEM) provides a viable solution [55].

1890 A/m and are taken from literature [56].

multilayer structures.

��

by an impedance change.

of the conducting layer [39].

structure is given by

Compared to wires and ribbons, the results obtained with thin film sensors have not been as good, and the highest GMI ratios reported are around 250% [42]. This may be due to the differences in the magnetic softness as well as the magnetic anisotropy, which is very well established in circumferential and transversal direction in wires and ribbons, respectively, and is difficult to control in thin films. In thin films transverse anisotropy is mainly realized through magnetic field deposition or field annealing, Fig. 8 (a) and (b) show the magnetization curve and domain structure of a Ni80Fe20 thin film (100nm thick) fabricated under a magnetic field of 200 Oe during deposition. A magnetic easy axis and domain structures in transverse direction are observed. Due to the small thickness, thin film GMI sensors normally operate at a higher frequency from hundred MHz to several GHz where the penetration depth is in the range of the film thickness.

**Figure 8.** (a) Magnetization curves obtained by vibrating sample magnetometry of a magnetic thin film (100nm of Ni80Fe20) in transversal and longitudinal directions. (b) Domain pattern of the magnetic layer. (c) Schematic of a typical multi layer GMI structure. The arrows in the ferromagnetic material indicate the magnetization of individual domains (simplified). Upon application of an external magnetic field *Hext*, the magnetization rotates into the direction of *Hext* (dotted arrows).

In general, a GMI sensor with high sensitivity consists of a stack of several material layers. In case of a tri-layer element, one conducting layer is sandwiched between two magnetic layers as shown in Fig. 8 (c). The conducting layer ensures a high conductivity and, in combination with the highly permeable magnetic layers, a large skin effect is obtained [51, 54]. An alternating current *Iac* mainly flowing through the conductor generates a transversal flux *Btran*, which magnetizes the magnetic layers. Upon the application of an external field *Hext* in longitudinal direction, the magnetization caused by *Iac* will be changed. This is equivalent to a change of the transversal permeability of the magnetic layers and is reflected by an impedance change.

366 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

the penetration depth is in the range of the film thickness.

*Hext*, the magnetization rotates into the direction of *Hext* (dotted arrows).

recent results will be presented.

technology producing a film thickness of some nanometers up to a few micrometers. Thin film GMI sensors are of great interest due to the advantages arising from the fabrication in terms of the flexibility in design and integration. They can easily be fabricated on the same substrate as the electronic circuit and other devices. In the context of passive and remote sensors, this is particularly relevant, since the GMI element can be easily integrated with an SAW device. For this reason, GMI thin film sensors will be discussed in more detail and our

Compared to wires and ribbons, the results obtained with thin film sensors have not been as good, and the highest GMI ratios reported are around 250% [42]. This may be due to the differences in the magnetic softness as well as the magnetic anisotropy, which is very well established in circumferential and transversal direction in wires and ribbons, respectively, and is difficult to control in thin films. In thin films transverse anisotropy is mainly realized through magnetic field deposition or field annealing, Fig. 8 (a) and (b) show the magnetization curve and domain structure of a Ni80Fe20 thin film (100nm thick) fabricated under a magnetic field of 200 Oe during deposition. A magnetic easy axis and domain structures in transverse direction are observed. Due to the small thickness, thin film GMI sensors normally operate at a higher frequency from hundred MHz to several GHz where

**Figure 8.** (a) Magnetization curves obtained by vibrating sample magnetometry of a magnetic thin film (100nm of Ni80Fe20) in transversal and longitudinal directions. (b) Domain pattern of the magnetic layer. (c) Schematic of a typical multi layer GMI structure. The arrows in the ferromagnetic material indicate the magnetization of individual domains (simplified). Upon application of an external magnetic field

In general, a GMI sensor with high sensitivity consists of a stack of several material layers. In case of a tri-layer element, one conducting layer is sandwiched between two magnetic layers as shown in Fig. 8 (c). The conducting layer ensures a high conductivity and, in combination with the highly permeable magnetic layers, a large skin effect is obtained [51, The analytical model of the impedance for a magnetic/conducting/magnetic tri-layer structure is given by

$$Z = \, R\_{dc} (1 - 2j\mu \frac{d\_1 d\_2}{\delta\_c^2})\,, \tag{16}$$

where *Rdc* is the dc resistance of the inner conductor, 2*d1* is the thickness of the conductor, *d2* is the thickness of the magnetic layers as shown in Fig. 8 (c) and *δc* is the penetration depth of the conducting layer [39].

Analytical solutions for the impedance of thin film GMI sensors can only be found for rather simple structures. In order to calculate the impedance of more complicated geometries, for example, a sandwich structure with isolation layers between the conductor and the magnetic layers [41], a meander structure multilayer [46] or to take into account edge effects, the finite element method (FEM) provides a viable solution [55].

Fig. 9 shows the comparison of the GMI ratios simulated for a single magnetic layer, a trilayer structure made of a magnetic/conducting/magnetic stack and a five-layer structure with isolation layers between the conducting and magnetic layers using the FEM. The simulated GMI sensors have a width of *w* = 50 μm and length of *l* = 200μm. The magnetic layers have a thickness of *tmag* = 1 μm and the conducting layer has a thickness of *tmet* = 4 μm. The material of the isolation layer is SiO2 with a thickness of 1μm. The conductivity of the ferromagnetic and conducting layers are 7.69×105 S/m ((CoFe)80B20) and 4.56×107 S/m (Gold), respectively. All parameters including *Ms* = 5.6×105 A/m, � = 2.2×105 m/(A·s), *α* = 0.3, *Ha*= 1890 A/m and are taken from literature [56].

**Figure 9.** Simulated GMI ratios of single layer, sandwiched multilayer and isolated sandwiched multilayer structures.

The results clearly show the performance increase achieved with the multilayer structures. Specifically, the isolated sandwich structure has a superior performance, which is due to preventing the current from flowing in the magnetic layer.

Surface Acoustic Wave Based Magnetic Sensors 369

2. Standard photolithography technique guarantees an accurate and reproducible

3. Thin film GMI sensors provide a wide range of working frequencies up to GHz, which

4. Thin film GMI sensor can have a minimized and flexible design as well as large

In this section, a detailed description of our recent work on the design, fabrication and

Fig. 10 shows a schematic of a GMI thin film sensor integrated with a SAW transponder. A wireless signal applied to the source IDT (IDT1) is converted to an SAW and propagates towards the other end of the substrate, where it is reflected from the reference IDT (IDT2) and the load IDT (IDT3). The reflected waves containing the reference and load information are received by IDT1 at different time instants and reconverted to a wireless electrical signal

**Figure 10.** Schematic of an integrated passive and remote magnetic field sensor consisting of a SAW

reflections from other structures on the substrate or the edge of the substrate.

In order to obtain high magnetic field sensitivity, the GMI sensor is matched to the output port (IDT3) at the working frequency of the SAW device. As the impedance of the GMI sensor changes with an applied magnetic field, the matching deteriorates, which causes the amplitude of the signal reflected from IDT3 to change. Since the piezoelectric material is sensitive to environmental changes, e.g. temperature, a reference IDT is used to provide a signal that enables the suppression of such noise by means of signal processing. Two metallic pads next to the input and output IDTs act as mechanical absorbers and suppress

Matching the sensor load to the optimal working point of IDT3 is a crucial aspect in the device design. Therefore, the influence of the load on the signal reflected from IDT3 is simulated. The interaction of a SAW with an IDT can be described by the P-matrix model

matches the high frequency requirement of the SAW transponders.

alignment of the two devices.

magnetic field sensitivity.

sent out via the antenna.

transponder and thin film GMI sensor.

**Design**

testing of an integrated SAW-GMI sensor is presented.

For the fabrication of thin film GMI sensors, Co-based and Fe-based amorphous magnetic alloys were used in earlier studies. Recently, permalloy, which is a NiFe compound, became popular as it provides very high permeability, zero magnetostriction and simple fabrication. Meander shaped multilayers and different stacks of magnetic and conductive layers using permalloy were developed. Some results are summarized in Table 2.


**Table 2.** Recent results on thin film GMI sensors.

GMI thin film sensors not only offer the advantages of standard microfabrication and straight-forward integration with SAW devices, but, as can be seen from Table 2, the operation frequency of GMI thin film sensors is also compatible with the one of SAW devices (usually from hundred MHz to several GHz and can be adjusted within a wide range.

## *3.2.5. Integrated SAW-GMI sensor*

In the first studies, SAW transponders and GMI wire sensors were combined to form remote devices [14-16]. GMI wires were selected for their high sensitivity, and they were bonded to the output IDT of the SAW device, which operated as a reflector, in order to act as load impedance. The strong dependence of the impedance on magnetic fields causes a considerable amplitude dependence of the reflected signal on magnetic fields. Even though these studies provided good results for passive and remote magnetic field sensors, the fabrication method for the GMI wires, which is not compatible with standard microfabrication, is a considerable problem with respect to reproducibility and costs, hence, hindering commercial success of such devices. In order to conquer this problem, a fully integrated SAW-GMI design utilizing standard microfabrication processes is required. The most viable option is a thin film GMI sensors for the following reasons:

1. Thin film GMI sensors can be produced by the same metallization processes as the SAW transponders and on the same substrate.


In this section, a detailed description of our recent work on the design, fabrication and testing of an integrated SAW-GMI sensor is presented.

## **Design**

368 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

permalloy were developed. Some results are summarized in Table 2.

Year Material Frequency GMI Ratio

preventing the current from flowing in the magnetic layer.

**Table 2.** Recent results on thin film GMI sensors.

*3.2.5. Integrated SAW-GMI sensor* 

range.

The results clearly show the performance increase achieved with the multilayer structures. Specifically, the isolated sandwich structure has a superior performance, which is due to

For the fabrication of thin film GMI sensors, Co-based and Fe-based amorphous magnetic alloys were used in earlier studies. Recently, permalloy, which is a NiFe compound, became popular as it provides very high permeability, zero magnetostriction and simple fabrication. Meander shaped multilayers and different stacks of magnetic and conductive layers using

1999 FeNiCrSiB/Cu/FeNiCrSiB 13MHz 77 2.8 [57] 2000 FeSiBCuNb/Cu/FeSiBCuNb 13MHz 80 2.8 [58] 2004 Ni81Fe19/Au/Ni81Fe19 300MHz 150 30 [59] 2004 (Ni81Fe19/Ag)n 1.8GHz 250 9.3 [42] 2005 FeCuNbSiB/SiO2/Cu/SiO2/FeCuNbSiB 5.45MHz 33 1.5 [60] 2011 NiFe/Ag/NiFe 1.8GHz 55 1.2 [61] 2011 NiFe/Cu/NiFe 20MHz 166 8.3 [62]

GMI thin film sensors not only offer the advantages of standard microfabrication and straight-forward integration with SAW devices, but, as can be seen from Table 2, the operation frequency of GMI thin film sensors is also compatible with the one of SAW devices (usually from hundred MHz to several GHz and can be adjusted within a wide

In the first studies, SAW transponders and GMI wire sensors were combined to form remote devices [14-16]. GMI wires were selected for their high sensitivity, and they were bonded to the output IDT of the SAW device, which operated as a reflector, in order to act as load impedance. The strong dependence of the impedance on magnetic fields causes a considerable amplitude dependence of the reflected signal on magnetic fields. Even though these studies provided good results for passive and remote magnetic field sensors, the fabrication method for the GMI wires, which is not compatible with standard microfabrication, is a considerable problem with respect to reproducibility and costs, hence, hindering commercial success of such devices. In order to conquer this problem, a fully integrated SAW-GMI design utilizing standard microfabrication processes is required. The

1. Thin film GMI sensors can be produced by the same metallization processes as the SAW

most viable option is a thin film GMI sensors for the following reasons:

transponders and on the same substrate.

(%)

Sensitivity (%/Oe)

Reference

Fig. 10 shows a schematic of a GMI thin film sensor integrated with a SAW transponder. A wireless signal applied to the source IDT (IDT1) is converted to an SAW and propagates towards the other end of the substrate, where it is reflected from the reference IDT (IDT2) and the load IDT (IDT3). The reflected waves containing the reference and load information are received by IDT1 at different time instants and reconverted to a wireless electrical signal sent out via the antenna.

**Figure 10.** Schematic of an integrated passive and remote magnetic field sensor consisting of a SAW transponder and thin film GMI sensor.

In order to obtain high magnetic field sensitivity, the GMI sensor is matched to the output port (IDT3) at the working frequency of the SAW device. As the impedance of the GMI sensor changes with an applied magnetic field, the matching deteriorates, which causes the amplitude of the signal reflected from IDT3 to change. Since the piezoelectric material is sensitive to environmental changes, e.g. temperature, a reference IDT is used to provide a signal that enables the suppression of such noise by means of signal processing. Two metallic pads next to the input and output IDTs act as mechanical absorbers and suppress reflections from other structures on the substrate or the edge of the substrate.

Matching the sensor load to the optimal working point of IDT3 is a crucial aspect in the device design. Therefore, the influence of the load on the signal reflected from IDT3 is simulated. The interaction of a SAW with an IDT can be described by the P-matrix model

introduced by Tobolka [63]. As shown in Fig. 11, *P*11 is the acoustic wave reflection at the output IDT [24]. Specifically, the dependence of *P*11 on the load impedance *Z = Z*(*Hext*) *+ Z*m, where *Z*(*Hext*) is the impedance of the GMI element and *Zm* is the matching impedance, is expressed as

$$P\_{11}(Z) = P\_{11,SC} + \frac{2 \cdot P\_{13}^2}{p\_{33} + \frac{1}{Z}} \, ^\prime \tag{17}$$

Surface Acoustic Wave Based Magnetic Sensors 371

Gold

6.25μm

30

Fig. 12 (a) shows the simulation result of the IDT's reflectivity as a function of the load. The slope of this plot corresponds to the magnetic field sensitivity. Therefore, the optimum matching capacitance can be determined. Fig. 12 (b) presents the rate of change of *P*11 for 1nH inductance changes (corresponding to a field change of approximately 50A/m). The result shows that with the optimum matching capacitance a maximum reflectivity change rate of 0.3dB/nH can be achieved. As the fabricated GMI sensor has an inductance change

**Figure 12.** (a) Magnitude *P*11 as a function of the matching capacitance and sensor inductance. (b) Rate

The piezoelectric substrate chosen for this application is LiNbO3 as it provides a strong electromechanical coupling corresponding to a high value of *P*13. The detailed design parameters of the SAW transponder are shown in Tab. 3. The working frequency of the device is 80MHz, resulting in a periodicity *p* of 50μm (Equ. (1). The value of *p* determines the electrode width and gap. The distances between the IDTs yield a 1.25μs delay between

> Electrode material

> > width

4 IDT segment number

The GMI sensor consists of a tri-layer structure with two ferromagnetic layers of 100nm in thickness made of Ni80Fe20 and a conducting copper layer with a thickness of 200nm. The

80MHz Aperture 30λ

Design parameter Design parameter

LiNbO3 (128 deg. Y-X cut)

Periodicity 50μm Electrode/gap

from 5nH to 15nH, a reflectivity change of 3dB can be expected.

of change (absolute value) of *P*11 for 1nH load inductance change.

IDT1 and IDT2 and a 0.625μs delay between IDT2 and IDT3.

Substrate material

Center frequency

Electrodes per segment

**Table 3.** Design parameters for the SAW device.

where *P*11,sc is the short circuit reflection coefficient, *P*13 is the electro-acoustic transfer coefficient and *P*33 is the input admittance of the transducer. In order to have a large change of *P*11, which is equivalent to the sensitivity of the SAW device loaded by an impedance sensor, the influence of Z in equation (17) needs to be large. Therefore, a SAW transducer with a small *P*11,sc and large *P*13 will provide a large sensitivity. *P*11,sc can be minimized by using a double electrode IDT design as shown in Fig. 10, which provides cancelation of the internal mechanical reflections of the IDT.

**Figure 11.** Electric and acoustic ports of the SAW sensor

The electro-acoustic transfer coefficient *P*13 and input admittance *P*33 can be obtained by,

$$P\_{13} = \frac{1}{2\sqrt{2Z\_a}}r\_m(1 - e^{-j\varphi\_m})\tag{18}$$

$$P\_{33} = j\omega \mathcal{C}\_{IDT} + \frac{r\_m^2}{z\_a} (1 - e^{-j\varphi\_m}) \tag{19}$$

Where *r*m is the ratio of the electrical to acoustical transformer, CIDT is the capacitance of the IDT, Za is the acoustic impedance and �� is the transit angle [63].

Since the GMI sensor is an inductive element, matching is accomplished by a series capacitance resulting a load impedance

$$\mathbf{Z} = \mathbf{1}/\text{j}\boldsymbol{\alpha}\mathbf{C}\_{\text{m}} + \mathbf{R} + \text{j}\boldsymbol{\alpha}\mathbf{L}(\mathbf{F}\_{\text{ext}}),\tag{20}$$

where *Cm* is the matching capacitance, *R* is the average resistance (over the considered magnetic field range) of the GMI sensor and *L*(*H*ext) the inductance of the GMI sensor.

Fig. 12 (a) shows the simulation result of the IDT's reflectivity as a function of the load. The slope of this plot corresponds to the magnetic field sensitivity. Therefore, the optimum matching capacitance can be determined. Fig. 12 (b) presents the rate of change of *P*11 for 1nH inductance changes (corresponding to a field change of approximately 50A/m). The result shows that with the optimum matching capacitance a maximum reflectivity change rate of 0.3dB/nH can be achieved. As the fabricated GMI sensor has an inductance change from 5nH to 15nH, a reflectivity change of 3dB can be expected.

**Figure 12.** (a) Magnitude *P*11 as a function of the matching capacitance and sensor inductance. (b) Rate of change (absolute value) of *P*11 for 1nH load inductance change.

The piezoelectric substrate chosen for this application is LiNbO3 as it provides a strong electromechanical coupling corresponding to a high value of *P*13. The detailed design parameters of the SAW transponder are shown in Tab. 3. The working frequency of the device is 80MHz, resulting in a periodicity *p* of 50μm (Equ. (1). The value of *p* determines the electrode width and gap. The distances between the IDTs yield a 1.25μs delay between IDT1 and IDT2 and a 0.625μs delay between IDT2 and IDT3.


**Table 3.** Design parameters for the SAW device.

370 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

expressed as

internal mechanical reflections of the IDT.

**Figure 11.** Electric and acoustic ports of the SAW sensor

capacitance resulting a load impedance

introduced by Tobolka [63]. As shown in Fig. 11, *P*11 is the acoustic wave reflection at the output IDT [24]. Specifically, the dependence of *P*11 on the load impedance *Z = Z*(*Hext*) *+ Z*m, where *Z*(*Hext*) is the impedance of the GMI element and *Zm* is the matching impedance, is

���(�) = ������ <sup>+</sup> �����

where *P*11,sc is the short circuit reflection coefficient, *P*13 is the electro-acoustic transfer coefficient and *P*33 is the input admittance of the transducer. In order to have a large change of *P*11, which is equivalent to the sensitivity of the SAW device loaded by an impedance sensor, the influence of Z in equation (17) needs to be large. Therefore, a SAW transducer with a small *P*11,sc and large *P*13 will provide a large sensitivity. *P*11,sc can be minimized by using a double electrode IDT design as shown in Fig. 10, which provides cancelation of the

The electro-acoustic transfer coefficient *P*13 and input admittance *P*33 can be obtained by,

� ��

Where *r*m is the ratio of the electrical to acoustical transformer, CIDT is the capacitance of the

Since the GMI sensor is an inductive element, matching is accomplished by a series

where *Cm* is the matching capacitance, *R* is the average resistance (over the considered

magnetic field range) of the GMI sensor and *L*(*H*ext) the inductance of the GMI sensor.

��� <sup>=</sup> � �����

IDT, Za is the acoustic impedance and �� is the transit angle [63].

��� = ������ <sup>+</sup> ��

� ����� �

, (17)

��(� � �����) (18)

Z = 1/jωCm + R +jωL(Hext), (20)

(� � �����) (19)

The GMI sensor consists of a tri-layer structure with two ferromagnetic layers of 100nm in thickness made of Ni80Fe20 and a conducting copper layer with a thickness of 200nm. The

sensor has a rectangular geometry of 100 μm× 4000 μm. The conducting layer is connected to the IDT3 [18].

Surface Acoustic Wave Based Magnetic Sensors 373

**Figure 14.** (a) Time domain measurement of the SAW-GMI device. Inset: Frequency domain measurement. (b) Comparison of the simulated and experimental device response together with the

Magnetic field sensors are one of the most widely used sensors and employed for many different applications. Current commercial magnetic sensors are wire connected to a circuit providing power and readout. These wire connections prevent the sensors from being used for certain applications. In addition, as the complexity and the number of devices, increases in modern systems such as automobiles, wire connections are becoming an increasing problem due to limited space. For those and other reasons, wireless solutions are being

As pointed out in the previous sections, SAW-based sensors have been developed for different applications, and this technology also provides a platform for realizing wireless and passive magnetic sensors. They can provide solutions for various applications, for example, where the sensors have to withstand harsh environmental conditions or are

Out of the countless applications for SAW-based passive and remote magnetic field sensors,

Nanotechnology and miniaturized systems are becoming increasingly popular in the biomedical field. Technologies based on magnetic effects are of particular interest since they can be controlled remotely via magnetic fields. For example, NVE Corporation recently developed a battery operated magnetic sensor to be used as a magnetic switch for implantable devices. When a magnetic field is applied, the sensor turns on triggering a specified action. It turns off when the field is removed. The sensor works at a stable operating point of 15 Oe [64]. Magnetic beads have been extensively used in many biomedical applications. These magnetic beads are made of ferromagnetic material ranging in size between 5 nm to 500 um. A new application of such particles promises benefits in

measured GMI ratio curve.

much sought after.

**4. Potential applications** 

required to have a long lifetime without maintenance.

a few will be highlighted in the following.

## **Fabrication**

The fabrication of the combined device is accomplished in several steps as shown in Fig. 13. On a LiNbO3 wafer, a 40 nm Ti adhesion layer and 200 nm gold layer are sputter deposited and patterned by ion milling into individual SAW devices. The leads and SMD footprints are designed together with the SAW device to facilitate an on-chip impedance matching circuit, which was accomplished by a 150pF capacitor connected in series with the GMI element. The GMI element comprises a tri-layer structure (Ni80Fe20(100nm) / Cu(200nm) / Ni80Fe20(100nm)) deposited at room temperature with a uniaxial magnetic field of 200 Oe applied in the transversal direction.

**Figure 13.** Fabrication flow chart of the integrated SAW-GMI device.

## **Results**

A network analyzer (Agilent E8363C) is used to apply an RF signal to IDT1 and measure the electric reflection coefficient (*S*11) of the input IDT, which is related to the admittance matrix of the whole device and *P*11. The time domain signal of *S*11 is converted from the frequency domain using fast Fourier transform. As shown in Fig. 14 (a), two reflection peaks at 2.45μs and 3.55μs are observed indicating the reflections from the reference IDT and the load IDT accordingly. The magnetic response of the integrated device is determined by applying a variable magnetic field in longitudinal direction to the device. A 2.4dB amplitude change of the reflection signal can be observed. A comparison of the simulated and experimental results together with the measured GMI ratio curve is shown in Fig. 14 (b).

**Figure 14.** (a) Time domain measurement of the SAW-GMI device. Inset: Frequency domain measurement. (b) Comparison of the simulated and experimental device response together with the measured GMI ratio curve.

## **4. Potential applications**

372 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

**Figure 13.** Fabrication flow chart of the integrated SAW-GMI device.

results together with the measured GMI ratio curve is shown in Fig. 14 (b).

A network analyzer (Agilent E8363C) is used to apply an RF signal to IDT1 and measure the electric reflection coefficient (*S*11) of the input IDT, which is related to the admittance matrix of the whole device and *P*11. The time domain signal of *S*11 is converted from the frequency domain using fast Fourier transform. As shown in Fig. 14 (a), two reflection peaks at 2.45μs and 3.55μs are observed indicating the reflections from the reference IDT and the load IDT accordingly. The magnetic response of the integrated device is determined by applying a variable magnetic field in longitudinal direction to the device. A 2.4dB amplitude change of the reflection signal can be observed. A comparison of the simulated and experimental

to the IDT3 [18].

applied in the transversal direction.

**Fabrication** 

**Results** 

sensor has a rectangular geometry of 100 μm× 4000 μm. The conducting layer is connected

The fabrication of the combined device is accomplished in several steps as shown in Fig. 13. On a LiNbO3 wafer, a 40 nm Ti adhesion layer and 200 nm gold layer are sputter deposited and patterned by ion milling into individual SAW devices. The leads and SMD footprints are designed together with the SAW device to facilitate an on-chip impedance matching circuit, which was accomplished by a 150pF capacitor connected in series with the GMI element. The GMI element comprises a tri-layer structure (Ni80Fe20(100nm) / Cu(200nm) / Ni80Fe20(100nm)) deposited at room temperature with a uniaxial magnetic field of 200 Oe

> Magnetic field sensors are one of the most widely used sensors and employed for many different applications. Current commercial magnetic sensors are wire connected to a circuit providing power and readout. These wire connections prevent the sensors from being used for certain applications. In addition, as the complexity and the number of devices, increases in modern systems such as automobiles, wire connections are becoming an increasing problem due to limited space. For those and other reasons, wireless solutions are being much sought after.

> As pointed out in the previous sections, SAW-based sensors have been developed for different applications, and this technology also provides a platform for realizing wireless and passive magnetic sensors. They can provide solutions for various applications, for example, where the sensors have to withstand harsh environmental conditions or are required to have a long lifetime without maintenance.

> Out of the countless applications for SAW-based passive and remote magnetic field sensors, a few will be highlighted in the following.

> Nanotechnology and miniaturized systems are becoming increasingly popular in the biomedical field. Technologies based on magnetic effects are of particular interest since they can be controlled remotely via magnetic fields. For example, NVE Corporation recently developed a battery operated magnetic sensor to be used as a magnetic switch for implantable devices. When a magnetic field is applied, the sensor turns on triggering a specified action. It turns off when the field is removed. The sensor works at a stable operating point of 15 Oe [64]. Magnetic beads have been extensively used in many biomedical applications. These magnetic beads are made of ferromagnetic material ranging in size between 5 nm to 500 um. A new application of such particles promises benefits in

cancer therapy by employing the particles either as drug carriers or heat sources (hyperthermia) [65]. In order to have better control of the treatment, magnetic sensors are considered to measure and detect the concentration of these magnetic particles.

Surface Acoustic Wave Based Magnetic Sensors 375

obtaining an optimized performance. A detailed description of the fabrication of an integrated SAW-GMI sensor is provided using standard microfabrication technologies. The GMI ratio of the fabricated sensor is 45 % The SAW-GMI sensor provides a sensitivity of

Magnetic field sensors have countless applications and are widely used in many different fields. The trend towards wireless operation, which is generally observed nowadays, drives the development of passive and remote magnetic field sensors. Several concepts of such sensors employing SAW devices have been presented in this chapter. The most promising one is a SAW-GMI sensor, which has been discussed in detail and which features wireless and battery less operation as well as durability and the ability to withstand harsh environments. This kind of sensor is considerable not only for providing existing applications with a wireless mode; it also largely expands the potential applications of

*Electrical Engineering Department, King Abdullah University of Science and Technology, Thuwal,* 

[1] Stockman H (1948) Communication by means of reflected power, Proceedings of IRE.

[2] Plath F, Schmeckebier O, Rusko M, Vandahl T, Luck H, Moller F, Malocha D C (1994) Remote sensor system using passive SAW sensors. ULTRASON.

[3] Reindl L, Scholl G, Ostertag T, Scherr H, Wolff U, Schmidt F (1998) Theory and application of passive SAW radio transponders as sensors. IEEE Trans. Ultrason.,

[4] Reeder T M and Cullen D E (1976) Surface-acoustic-wave pressure and temperature

[5] Lee K, Wang W, Kim G and Yang S (2006) Surface Acoustic Wave Based Pressure Sensor with Ground Shielding over Cavity on 41° YX LiNbO3. Jpn. J. Appl. Phys. 45:

[6] Pohl A (1997) Wirelessly interrogable surface acoustic wave sensors for vehicular

*Electrical Engineering Department, Georgia Institute of Technology, Atlanta, Georgia, USA* 

3 dB/mT, and its output corresponds well with the simulation results.

magnetic field sensors.

Bodong Li and Jürgen Kosel

**Author details** 

Hommood Al Rowais

36: 1196-1204

1: 585-588

5974–5980

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sensors. Proceedings of the IEEE. 64: 754-756

applications. IEEE T Instrum Meas. 46: 1031-1038

**6. References** 

*Saudi Arabia* 

The automotive industry extensively uses magnetic sensors for different purposes, for example, to measure current in electric vehicles [66] or the rotation speed of gears [67]. Another application employs magnetic sensors to detect passing vehicles using lane markers [68]. Such a system could also be used to detect vehicle speed by measuring the time between two markers of a fixed distance. In yet another application, developed by Stendl et al [69], the wear of a vehicle's tire is detected by measuring the field of magnetic beads embedded in the rubber of tire treads. As the tread size decreases, the magnetic field also decreases. A wireless magnetic sensor is placed just below the threads.

Construction monitoring is an upcoming application for wireless sensor. Long-term monitoring of metallic reinforcements in, e.g., bridges or buildings requires passive and remote sensors, which are capable of detecting changes of the metal. Similarly, the detection of internal defects or corrosion of pipelines is of great interest. Gloria et al [70] developed an Internal Corrosion Sensor (ICS) consisting of a magnet and a Hall sensor. A disturbance in the magnetic field caused by changes of the metal changes the sensor readout. This information is used for both to detect and size the defects.

## **5. Conclusion**

In this chapter we discussed different types of SAW-based, magnetic sensors including resonators, phase shifters and loaded transponders. Sensitivity to magnetic fields can be achieved by either changing the properties of the IDT or delay line utilizing magnetostrictive materials or loading the output IDT with a magnetic field sensor. GMI sensors feature a very high sensitivity and wide range of operating frequencies and, therefore, constitute an especially suitable load. The principle of GMI sensors is described in detail and different GMI structures are discussed. While the highest sensitivity has been obtained with GMI microwires, thin film GMI sensor are advantageous because they can be produced using standard microfabrication methods, and they can be easily integrated with a SAW transponder on the same substrate. These features are crucial with respect to production complexity and costs.

A SAW transponder combined with a GMI element connected to the output IDT is a passive and remote magnetic field sensor, which responds to an interrogation signal with a delayed response signal. The design of such a device needs to take into account different aspects like operation frequency, dimensions of IDTs and delay line or matching the load with the output IDT. In order to obtain a high sensitivity, an impedance change of the GMI element caused by a magnetic field, has to yield a large change in the SAW reflected from the output IDT. A model is presented to simulate the electro-acoustic interaction of the output IDT with the GMI sensor's impedance and the impedance matching capacitance. The simulation results provide information regarding the matching parameters and are invaluable for obtaining an optimized performance. A detailed description of the fabrication of an integrated SAW-GMI sensor is provided using standard microfabrication technologies. The GMI ratio of the fabricated sensor is 45 % The SAW-GMI sensor provides a sensitivity of 3 dB/mT, and its output corresponds well with the simulation results.

Magnetic field sensors have countless applications and are widely used in many different fields. The trend towards wireless operation, which is generally observed nowadays, drives the development of passive and remote magnetic field sensors. Several concepts of such sensors employing SAW devices have been presented in this chapter. The most promising one is a SAW-GMI sensor, which has been discussed in detail and which features wireless and battery less operation as well as durability and the ability to withstand harsh environments. This kind of sensor is considerable not only for providing existing applications with a wireless mode; it also largely expands the potential applications of magnetic field sensors.

## **Author details**

374 Modeling and Measurement Methods for Acoustic Waves and for Acoustic Microdevices

decreases. A wireless magnetic sensor is placed just below the threads.

information is used for both to detect and size the defects.

**5. Conclusion** 

production complexity and costs.

considered to measure and detect the concentration of these magnetic particles.

cancer therapy by employing the particles either as drug carriers or heat sources (hyperthermia) [65]. In order to have better control of the treatment, magnetic sensors are

The automotive industry extensively uses magnetic sensors for different purposes, for example, to measure current in electric vehicles [66] or the rotation speed of gears [67]. Another application employs magnetic sensors to detect passing vehicles using lane markers [68]. Such a system could also be used to detect vehicle speed by measuring the time between two markers of a fixed distance. In yet another application, developed by Stendl et al [69], the wear of a vehicle's tire is detected by measuring the field of magnetic beads embedded in the rubber of tire treads. As the tread size decreases, the magnetic field also

Construction monitoring is an upcoming application for wireless sensor. Long-term monitoring of metallic reinforcements in, e.g., bridges or buildings requires passive and remote sensors, which are capable of detecting changes of the metal. Similarly, the detection of internal defects or corrosion of pipelines is of great interest. Gloria et al [70] developed an Internal Corrosion Sensor (ICS) consisting of a magnet and a Hall sensor. A disturbance in the magnetic field caused by changes of the metal changes the sensor readout. This

In this chapter we discussed different types of SAW-based, magnetic sensors including resonators, phase shifters and loaded transponders. Sensitivity to magnetic fields can be achieved by either changing the properties of the IDT or delay line utilizing magnetostrictive materials or loading the output IDT with a magnetic field sensor. GMI sensors feature a very high sensitivity and wide range of operating frequencies and, therefore, constitute an especially suitable load. The principle of GMI sensors is described in detail and different GMI structures are discussed. While the highest sensitivity has been obtained with GMI microwires, thin film GMI sensor are advantageous because they can be produced using standard microfabrication methods, and they can be easily integrated with a SAW transponder on the same substrate. These features are crucial with respect to

A SAW transponder combined with a GMI element connected to the output IDT is a passive and remote magnetic field sensor, which responds to an interrogation signal with a delayed response signal. The design of such a device needs to take into account different aspects like operation frequency, dimensions of IDTs and delay line or matching the load with the output IDT. In order to obtain a high sensitivity, an impedance change of the GMI element caused by a magnetic field, has to yield a large change in the SAW reflected from the output IDT. A model is presented to simulate the electro-acoustic interaction of the output IDT with the GMI sensor's impedance and the impedance matching capacitance. The simulation results provide information regarding the matching parameters and are invaluable for Bodong Li and Jürgen Kosel

*Electrical Engineering Department, King Abdullah University of Science and Technology, Thuwal, Saudi Arabia* 

Hommood Al Rowais *Electrical Engineering Department, Georgia Institute of Technology, Atlanta, Georgia, USA* 

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**Chapter 16** 

© 2013 Babkin, licensee InTech. This is an open access chapter distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use,

© 2013 The Author(s). Licensee InTech. This chapter is distributed under the terms of the Creative Commons Attribution License http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution,

distribution, and reproduction in any medium, provided the original work is properly cited.

© 2013 Babkin, licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

**Electromagnetic and Acoustic** 

Additional information is available at the end of the chapter

**1.1. Electromagnetic and acoustic transformation** 

Materials in which EMAT is possible to occur:

EMAT may proceed by three basic mechanisms.

will fluctuate under the influence of Lorentz force:

properties, and magnetostriction (e.g. iron, nickel, steels);

magnetostriction (ferrite, amorphous, rare-earth materials).

Sergey E. Babkin

**1. Introduction** 

*EMAT mechanisms* 

http://dx.doi.org/10.5772/56307

**Transformation of Surface Acoustic** 

**Waves and Its Application in Various Tasks** 

Electromagnetic and acoustic transformation (EMAT) is transformation of high-pitched electromagnetic oscillations in the inductive sensor over a specimen into acoustic oscillations in the specimen. For the transformation to be performed the padding constant magnetic field is required. This process is referred to as a direct EMAT. Further oscillations are extended in the specimen in the shape of acoustic waves. Acoustic waves can be deduced outside by means of revertive EMAT when acoustic oscillations in the surface layer of a specimen will be transformed to electromagnetic oscillations in the receiving sensor. The overall process is as follows: a direct EMAT, a distribution of acoustic waves and a revertive EMAT, which is in practice referred to as a double EMAT or just EMAT [1, 2].


1. A vortical current mechanism (Lorentz force mechanism). Electrons in a surface layer

and reproduction in any medium, provided the original work is properly cited.

