**11. Antenna design**

76 Wireless Communications and Networks – Recent Advances

**10. Tapered composite right/left-handed transmission-line (CRLH-TL) leaky-**

Recently composite right/left-handed (CRLH) leaky-wave antennas (LWAs) have been shown as one of the applications of the CRLH transmission line (TL) metamaterials thanks to their advantages of fabrication simplicity and frequency/electrically scanning capability without any complex feeding network. Neverthless the fixed geometrical size of a unit cell of the CRLH-TL Leaky-wave antennas, prevents the possibility to improve the antenna

It is well known as a composite right/left-handed transmission-line (CRLH-TL) metamaterials, used for the leaky-wave antennas (LWAs) allow to obtain a superior frequency scanning ability than its conventional counterpart [26-27]. The leaky-wave antennas possess the advantages of low-profile, easy matching, fabrication simplicity, and

However, the conventional leaky-wave antennas suffer from major limitations in their scanning capabilities. In fact the radiation pattern is restricted to strictly positive for uniform configurations, or to a discontinuous range of negative or positive excluding broadside direction, for periodic configurations. The CRLH LWAs have essentially suppressed these limitations, being able to scans the entire space from = -90° to = +90°

Although actually the designs of CRLH-TL for LWAs available in the literature, are developed as a different number of unit cell with a fixed geometrical size for all the unit

These design prevents the possibility to improve the antenna bandwidth "tapering" the geometrical size of unit cell. In order to obtain an improvement of the antenna bandwidth a novel design of CRLH LWAs was used in our work. The simulation results of the the CRLH

frequency/electrically scanning capability without any complex feeding network.

and thereby paved the way for novel perspectives for leaky-wave antennas.

Fig. 33. Experimental return loss of half bend LWA Type III.

bandwidth "tapering" the geometrical size of unit cell.

**wave antennas (LWAs)** 

cells of the entire antenna.

It has been shown that the leakage rate of the CRLH-TL LWA can be altered by using different sizes of the unit-cell [27] as shown in Fig.34.


Fig. 34. Different size and number of fingers of CRLH-TL unit cell.

In detail the radiation resistance, of the unit-cell having four fingers and the unit-cell of six fingers, both the unit-cells designed to have the phase origin at the same frequency, shows two different bandwidth as mentioned in [27-28].

The radiation resistance of the four finger antenna is always higher than that of the sixfinger one, which implies faster decay of power (more leakage) along the structure for the former.

Moreover it should be noted that increasing the number of fingers the size of the unit-cell has to be reduced in order to have the same centre-frequency for the antenna antennas, otherwise, the centre-frequency for the antenna with unit-cell which have the larger number of fingers will shift down to a lower frequency [29-30].

As shown in [21-22] for a simple microstrip leaky wave antenna the radiation bandwidth is governed by the line width once the substrate is fixed. The bandwidth can be improved by adopting a tapered line structure (Fig.35), where, the radiation of different frequency regions leaks from different parts of the antenna.

Fig. 35. A taper layout of LWA with a different frequency regions leaks from different parts of the antenna.

In fact from the propagation characteristics of the leaky wave antenna, we known that the leakage radiate phenomena, can only be noted above the cutoff frequency of higher order mode, and below the frequency such that, the phase constant is equal at the free space wave number. Decreasing the width of the antenna for a microstrip leaky-wave antenna the cutoff frequency increases shift toward high frequency. This behaviour allows to design a multisection microstrip LWA according [21-22] superimposing different section, in which each section can radiate in a different and subsequence frequency range, obtaining a broadband antennas. In this way each section should be into bound region, radiation region or reactive region, permitting the power, to uniformly radiated at different frequencies.

Following these idea in the our developed procedure we have applied a process to get the dimension of the physical parameters of the unit cell shows in Fig. 36 whit different optimized number of fingers (see Fig. 37). Naturally we have calculated the extraction parameters of every cell of CRLH implementation: LR, CR, LL, and CL using the equation mentioned in [26].

In the case of CRLH transmission line based LWA the amount of radiation by the unit cell can be related to the beam shape required and thus can be used to determine the total size of the structure as mentioned in [31].

Fig. 36. Unit cell equivalent circuit with radiation resistance.

Given the unit cell equivalent circuit in Fig. 36, we have the per unit length series impedance and per unit length shunt admittance as follows [31]:

$$\dot{\mathbf{Z}}^{\cdot} = \overset{\cdot}{R}\_{a}^{\cdot} + \overset{\cdot}{R}\_{l}^{\cdot} + j o \overset{\cdot}{\mathbf{L}}\_{R}^{\cdot} - \frac{\dot{j}}{o \overset{\cdot}{\mathbf{C}}\_{L}^{\cdot}} \tag{21}$$

$$\text{Y}^{\prime} = \text{G}^{\prime} + \text{R}\_{l}^{\prime} + j\alpha \text{C}\_{R}^{\prime} - \frac{\dot{j}}{\alpha \text{L}\_{L}^{\prime}} \tag{22}$$

Where ' *Ra* represents radiation resistance per unit length, ' *Rl* represents the per unit length resistance associated with transmission loss, ' *LR* and ' *CR* denotes per unit length parasitic inductance and capacitance respectively, ' *CL* and ' *LL* denotes times unit cell length left-hand capacitance and inductance respectively.

Propagation constant and characteristics impedance are given by the following relations:

In fact from the propagation characteristics of the leaky wave antenna, we known that the leakage radiate phenomena, can only be noted above the cutoff frequency of higher order mode, and below the frequency such that, the phase constant is equal at the free space wave number. Decreasing the width of the antenna for a microstrip leaky-wave antenna the cutoff frequency increases shift toward high frequency. This behaviour allows to design a multisection microstrip LWA according [21-22] superimposing different section, in which each section can radiate in a different and subsequence frequency range, obtaining a broadband antennas. In this way each section should be into bound region, radiation region or reactive region, permitting the power, to uniformly radiated at different frequencies.

Following these idea in the our developed procedure we have applied a process to get the dimension of the physical parameters of the unit cell shows in Fig. 36 whit different optimized number of fingers (see Fig. 37). Naturally we have calculated the extraction parameters of every cell of CRLH implementation: LR, CR, LL, and CL using the equation

In the case of CRLH transmission line based LWA the amount of radiation by the unit cell can be related to the beam shape required and thus can be used to determine the total size of

Given the unit cell equivalent circuit in Fig. 36, we have the per unit length series impedance

*al R*

*l R*

'

(21)

(22)

*LR* and ' *CR* denotes per unit length parasitic

*LL* denotes times unit cell length left-hand

*L*

'

*L*

''' '

''' '

*<sup>j</sup> Z R R jL <sup>C</sup>* 

*<sup>j</sup> Y G R jC <sup>L</sup>* 

Where ' *Ra* represents radiation resistance per unit length, ' *Rl* represents the per unit length

Propagation constant and characteristics impedance are given by the following relations:

mentioned in [26].

the structure as mentioned in [31].

Fig. 36. Unit cell equivalent circuit with radiation resistance.

and per unit length shunt admittance as follows [31]:

resistance associated with transmission loss, '

capacitance and inductance respectively.

inductance and capacitance respectively, ' *CL* and '

$$
\gamma = \alpha + j\beta = \sqrt{\mathbf{Z}^\prime \mathbf{Y}^\prime} \tag{23}
$$

$$Z\_c = \sqrt{\frac{Z^\cdot}{Y^\cdot}}\tag{24}$$

From the above expression of propagation constant and line impedance we can find the centre frequency of the CRLH LWA [31].

These procedure was applied for subsequency frequency range of interest able to obtain a broadband antenna and a narrow-beam radiation pattern more than the uniform CRLH-TL LWA.

Fig. 37. Layout of a single unit cell of CRLH-TL LWA were p is the length of the unit cell period, lc, is the length of the capacitor *w* and *ws* represent, the overall width of its finger and the width of the stub respectively.

The optimized antenna design as we can see in Fig. 38 was obtained considering the sequence of 16 cells composed respectively by 4 cell with 12 fingers, 4 cells with 10 fingers,

Fig. 38. Layout of the 16 unit-cell CRLH-TL LWA with cells of 10, 8 and 6 fingers.

4 cell with 8 fingers and 4 cell with 6 fingers for the entire length of the antenna of 207.55 mm and width between 2.9 mm (cells with 12 fingers) and 5.9 mm (cells with 6 fingers).

#### **12. Simulation and experimental results**

In the following Fig. 39 and Fig. 40 are showing the simulation data of the return loss obtained with a 3D EM commercial software, of the uniform 16 unit-cell CRLH-TL LWA of 10 fingers compared with the results of tapered 16 unit-cell CRLH-TL LWA. Instead in Fig. 41 and in Fig. 42, are shown the results of the radiation pattern of the uniform CRLH-TL LWA compared with 16 unit-cell of 10 fingers and tapered 16 unit-cell CRLH-TL LWA. It is evident the good performance of the tapered 16 unit-cell CRLH-TL LWA compared with uniform CRLH-TL LWA in term of broadband and narrowbeam.

Fig. 39. Return loss (S11) of the uniform 16 unit-cell CRLH-TL LWA with 10 fingers.

Fig. 40. Return loss (S11) of the 16 unit-cell CRLH-TL LWA with cells of 12, 10, 8 and 6 fingers.

4 cell with 8 fingers and 4 cell with 6 fingers for the entire length of the antenna of 207.55 mm and width between 2.9 mm (cells with 12 fingers) and 5.9 mm (cells with 6 fingers).

In the following Fig. 39 and Fig. 40 are showing the simulation data of the return loss obtained with a 3D EM commercial software, of the uniform 16 unit-cell CRLH-TL LWA of 10 fingers compared with the results of tapered 16 unit-cell CRLH-TL LWA. Instead in Fig. 41 and in Fig. 42, are shown the results of the radiation pattern of the uniform CRLH-TL LWA compared with 16 unit-cell of 10 fingers and tapered 16 unit-cell CRLH-TL LWA. It is evident the good performance of the tapered 16 unit-cell CRLH-TL LWA compared with

Fig. 39. Return loss (S11) of the uniform 16 unit-cell CRLH-TL LWA with 10 fingers.

Fig. 40. Return loss (S11) of the 16 unit-cell CRLH-TL LWA with cells of 12, 10, 8 and 6

fingers.

**12. Simulation and experimental results** 

uniform CRLH-TL LWA in term of broadband and narrowbeam.

Fig. 41. Radiation pattern of the E field of the uniform CRLH-TL LWA for f=1.12 GHz (red line), f=2.30 GHz (brown line), f= 3.20 GHz (blu line).

Fig. 42. Radiation pattern of the E field of the tapered CRLH-TL LWA for f=1.12 GHz (red line), f=2.30 GHz (brown line), f= 3.20 GHz (blu line).

The simulation results were compared with experimental results made on a prototype of CRLH-TL LWA (see Fig. 43) designed with 16 unit-cell on Rogers RT/duroid 5880 substrate with dielectric constant εr = 2.2 and thickness h = 62 mil (loss tangent = 0.0009) showing a quite good agreement with simulated results of tapered 16 unit-cell CRLH-TL LWA as we can see in Fig. 44 and Fig. 45.

Fig. 43. A prototype and its detail of Radiation patter of tapered 16 unit-cell CRLH-TL LWA made on Rogers RT/duroid 5880.

Fig. 44. Experimental return loss (S11) of the 16 unit-cell prototype CRLH-TL LWA with cells of 12, 10, 8 and 6 fingers.

Fig. 45. Experimental E field radiation pattern of tapered prototype CRLH-TL LWA for f=1.12 GHz (red line), f=2.30 GHz (brown line), f= 3.20 GHz (blu line).

## **13. Meander antenna**

Nowadays, miniaturization of electronic devices is the main request that productions have to fulfil. In this process, the reduction of the antenna size is the crucial challenge to face, being its dimensions related to the working frequency.

The rapid developing of the modern society has become to a crescent interest for the wireless communications. Nowadays, everybody wants to be connected everywhere

Fig. 43. A prototype and its detail of Radiation patter of tapered 16 unit-cell CRLH-TL LWA

Fig. 44. Experimental return loss (S11) of the 16 unit-cell prototype CRLH-TL LWA with

Fig. 45. Experimental E field radiation pattern of tapered prototype CRLH-TL LWA for

Nowadays, miniaturization of electronic devices is the main request that productions have to fulfil. In this process, the reduction of the antenna size is the crucial challenge to face,

The rapid developing of the modern society has become to a crescent interest for the wireless communications. Nowadays, everybody wants to be connected everywhere

f=1.12 GHz (red line), f=2.30 GHz (brown line), f= 3.20 GHz (blu line).

being its dimensions related to the working frequency.

made on Rogers RT/duroid 5880.

cells of 12, 10, 8 and 6 fingers.

**13. Meander antenna** 

without the use of cumbersome devices. The current tendency goes towards portable terminals that have to be light and hand pocket. A suitable antenna for the portable terminals should be low cost, low profile, light weight and especially small size.

Printed antennas are commonly used for their simple structure and easy fabrication. As applications are space limited, it is challenging to design an antenna of small size but with simple tunable feature. For microstrip antennas, some techniques, such as making slots in their structure or using high dielectric constant substrates, can be used to reduce the antenna size. However, it results in the narrow bandwidths for its high Q factor and the low radiation efficiency.

In the following sentences is described an antenna printed on a substrate with low dielectric constant in order to get a reliable bandwidth. Moreover, the meander configuration allows us to reduce the antenna size keeping good radiation performance.

Meander dipole antennas have been already designed through numerical techniques that apply either time- or frequency-domain algorithms demanding high computational efforts and long-time processing. The common approach in the design of meander antenna is to draw a meander path with a suited length to the working frequency in commercial software and run simulations. Nevertheless, This empirical approach may lead to several consecutive trials and verifications. In order to decrease the long-time processing and avoid these cut and try methods, it would be convenient to start simulations with a commercial software having an antenna size close to its optimized dimensions. Thus, a good initial configuration can strongly affect the numerical convergence efficiency and the design process would be quicker.

This paper presents a transmission line model that provides an initial geometrical configuration of the antenna that allows us a computational improvement in the design of meander antennas. The dimensions obtained from the model have been used to run a simulation with a commercial software and an antenna resonating very close to that working frequency has been achieved. Finally, a quick optimization has been performed to definitely tune the antenna according to the ISM band.
