**2.1 Dual-polarized slot antenna**

For the purpose to realize dual orthogonal polarizations, slot structure is selected as the main radiator. As shown in Fig. 2, both vertical and horizontal polarizations can exist simultaneously in a rectangular slot. The operating frequency is dictated by the widths of the slot. The slot also has the advantages of wide bandwidth, bi-directional radiation pattern and high efficiency (Lee et al, 2009). However, how to excite these two polarizations is still a question. The traditional method is to feed both polarizations in the same way through two adjacent sides of the slot. Thus, the feeding structure is simple but with large dimension, which isn't able to fulfil our requirement of compact size.

Fig. 2. Polarization mode in slot: (a) vertical polarization, (b) horizontal polarization.

In order to excite dual orthogonal polarizations in a compact structure, we utilized the dual modes of co-planar waveguide (CPW). Fig. 3 shows the geometry of the proposed antenna with CPW feeding structure. The overall dimensions of the antenna are 100x80 mm2. The antenna is made of the substrate of FR4 (εr=4.4, tanδ=0.01), whose thickness is 1 mm. A 52x50 mm2 slot, etched in the front side of light region, serves as the main radiator. In the back side of dark region, an L-shaped microstrip line is fed through port 1. The CPW is fed through port 2 in the front side. As shown in Fig. 4(a), when feeding through port 1, a normal odd mode of CPW is excited to feed the vertical polarization mode. When feeding through port 2, as shown in Fig. 4(b), the mode in the CPW is the even mode as a slot line, which can excite the horizontal polarization mode.

Fig. 3. The geometry of the proposed antenna. (L=100 mm, LS=50 mm, LG=36 mm, L0=15 mm; L1=15 mm, L2=32 mm, L3=12.5 mm, L4=25.5 mm; L5=19 mm, W1=1.9 mm, W2=6 mm, WS=52 mm, W=80 mm, S1=0.35 mm, S2=0.5 mm. Reprinted from (Li et al, 2010) by the permission of IEEE).

Fig. 4. Feeding modes in CPW: (a) odd mode, (b) even mode.

In order to excite dual orthogonal polarizations in a compact structure, we utilized the dual modes of co-planar waveguide (CPW). Fig. 3 shows the geometry of the proposed antenna with CPW feeding structure. The overall dimensions of the antenna are 100x80 mm2. The antenna is made of the substrate of FR4 (εr=4.4, tanδ=0.01), whose thickness is 1 mm. A 52x50 mm2 slot, etched in the front side of light region, serves as the main radiator. In the back side of dark region, an L-shaped microstrip line is fed through port 1. The CPW is fed through port 2 in the front side. As shown in Fig. 4(a), when feeding through port 1, a normal odd mode of CPW is excited to feed the vertical polarization mode. When feeding through port 2, as shown in Fig. 4(b), the mode in the CPW is the even mode as a slot line,

Fig. 3. The geometry of the proposed antenna. (L=100 mm, LS=50 mm, LG=36 mm, L0=15 mm; L1=15 mm, L2=32 mm, L3=12.5 mm, L4=25.5 mm; L5=19 mm, W1=1.9 mm, W2=6 mm, WS=52 mm, W=80 mm, S1=0.35 mm, S2=0.5 mm. Reprinted from (Li et al, 2010) by the

Fig. 4. Feeding modes in CPW: (a) odd mode, (b) even mode.

which can excite the horizontal polarization mode.

permission of IEEE).

The current distributions of both polarizations are shown in Fig. 5 for better explanation. A half wavelength distribution appears on each side of slot. Dimensions of LS and WS determine the resonant frequencies of the vertical mode and horizontal mode respectively. The L3 is the tuning parameter for matching port 1. To match port 2, dimensions of W2, L5 and L6 need to be optimized. Due to the symmetric and anti-symmetric characteristics of the two modes in CPW, high isolation can be achieved between two ports. As a result, the feeding structure can excite both polarization modes simultaneously and independently.

Fig. 5. Current distributions of (a) vertical polarization and (b) horizontal polarization.

To validate the design, the S parameters of the proposed antenna are simulated using Ansoft high frequency structure simulator (HFSS). The antenna has also been fabricated and measured. Fig. 6 shows the measured S parameter of the proposed antenna in solid lines, compared with the simulated ones in dash lines. The centre frequencies of the dual polarizations are both 2.4GHz. The bandwidths of -10dB reflection coefficient are 670MHz (1.96-2.63GHz, 27.9%) and 850MHz (1.93-2.75GHz, 35.4%) for horizontal polarization and vertical polarization, respectively. Throughout the WLAN frequency band (2.4-2.484GHz), the isolation between two ports in the required band is lower than -32.6dB. These results show that the proposed antenna is simpler, more compact than the references (Barba, 2008; Mak et al, 2007; Lee et al, 2009).

Fig. 6. Simulated and measured S parameters of the proposed antenna.

The radiation patterns of the proposed antenna when feeding through port 1 and 2 are shown in Fig. 7 and Fig. 8. For port 1, the vertical polarization case, the 3dB beam widths are

Fig. 7. Measured and simulated radiation patterns when feeding from port 1 at 2.4 GHz: (a) X-Y plane (b) Y-Z plane.

Fig. 8. Measured and simulated radiation patterns when feeding from port 2 at 2.4 GHz: (a) X-Y plane (b) Y-Z plane.

100° and 70° in E-plane (Y-Z plane) and H-plane (X-Y plane). From these results it may be noted that the cross polarization in X-Y plane is worse than what was achieved in earlier designs as values for cross polarization are not lower than -15dB. From the radiation patterns, however, we can observe that the poles of Eφ and Eθ are almost corresponding to the maximum of each other, which means the integration of the two patterns is close to zero. In other words, the signals of co and cross polarizations are almost uncorrelated. In the Y-Z plane, the cross polarization level is sufficiently low to be ignored. For port 2, the horizontal polarization is the dominant polarization. The 3dB beam widths are 60° and 180° in E-plane (X-Y plane) and H-plane (Y-Z plane). From the above discussion, we may conclude that the signals received by the two ports are uncorrelated, so dual-polarization in single antennas can be treated as two independent antennas. The radiation efficiency and gain of the proposed antenna are also measured. In the WLAN band of 2.4-2.484GHz, the efficiency is better than 91.2% and 84.4% for port 1 and 2; and the gain is better than 3.85 dBi and 5.21

The radiation patterns of the proposed antenna when feeding through port 1 and 2 are shown in Fig. 7 and Fig. 8. For port 1, the vertical polarization case, the 3dB beam widths are

Fig. 7. Measured and simulated radiation patterns when feeding from port 1 at 2.4 GHz: (a)

Fig. 8. Measured and simulated radiation patterns when feeding from port 2 at 2.4 GHz: (a)

100° and 70° in E-plane (Y-Z plane) and H-plane (X-Y plane). From these results it may be noted that the cross polarization in X-Y plane is worse than what was achieved in earlier designs as values for cross polarization are not lower than -15dB. From the radiation patterns, however, we can observe that the poles of Eφ and Eθ are almost corresponding to the maximum of each other, which means the integration of the two patterns is close to zero. In other words, the signals of co and cross polarizations are almost uncorrelated. In the Y-Z plane, the cross polarization level is sufficiently low to be ignored. For port 2, the horizontal polarization is the dominant polarization. The 3dB beam widths are 60° and 180° in E-plane (X-Y plane) and H-plane (Y-Z plane). From the above discussion, we may conclude that the signals received by the two ports are uncorrelated, so dual-polarization in single antennas can be treated as two independent antennas. The radiation efficiency and gain of the proposed antenna are also measured. In the WLAN band of 2.4-2.484GHz, the efficiency is better than 91.2% and 84.4% for port 1 and 2; and the gain is better than 3.85 dBi and 5.21

X-Y plane (b) Y-Z plane.

X-Y plane (b) Y-Z plane.

dBi for port 1 and 2. The proposed antenna is a candidate for compact volume dualpolarized antenna application.

#### **2.2 Dual-polarized loop antenna**

The half wavelength resonant structure, such as the patch and the slot, is able to be adopted in dual-polarized antenna design. In order to realize even more compact dimension, we choose the loop antenna, whose circumference is one wavelength. The radiation patterns of the slot and the loop are almost the same. Also, the loop element can support two orthogonal polarizations using the same structure, shown in Fig. 9. Seen from these two modes, the current distribution is 90° rotated from one to another one. Good orthogonality is illustrated with high isolation. The current distribution of its one–wavelength mode is dictated by the feeding position, and feed should not be arranged at the position of the current null. However, the maximum point of one mode is the null of the other mode. It is difficult to feed the dual polarizations in one side of loop.

Fig. 9. Modes in loop antenna: (a) vertical polarization, (b) horizontal polarization.

The feeding method should be considered carefully. In order to excite two orthogonal one– wavelength modes, it is common to arrange two feeds at two orthogonal positions, which will make the overall dimension much larger. A compact size could be realized if such two modes of operation are fed at only one position. The compact CPW feed backed with microstrip line adopted in the last design is an effective solution to feed the dual-mode of loop antenna. Fig. 10 shows the geometry of the loop antenna, which is quite similar as the slot design. This antenna consists of a rectangular loop, a CPW feeding and a microstrip line, and supported by the same FR4 board as last design with the thickness of 1 mm. The loop has width of 4 mm; narrower than the slot design. The loop and CPW are etched on the front side and the microstrip line is printed on the back side.

Fig. 10. Geometry of the proposed loop antenna. (L1=53 mm, L2=33 mm, L3=20 mm, L4=16 mm; L5=5.1 mm, L6=18.5 mm, L7=6.5 mm, W1=40 mm, W2=32 mm, W3=2 mm, W4=1.9 mm, S=0.5 mm. Reprinted from (Li et al, 2011a) by the permission of IEEE).

When the loop fed through port 1, the CPW operates at its typical symmetrical mode. In this mode the vertical polarization is excited. The inner conductor works as a monopole with the vertical polarization. The energy is coupled from monopole to the loop, exciting the vertical polarization mode. It is a good solution to feed the one-wavelength mode at the position of current null. The radiation consists of two modes, the one-wavelength mode of the loop and a monopole mode. When the loop is fed through port 2, the horizontal polarization of the loop antenna is excited. The feed is exactly at the maximum of current, and the horizontal mode is clearly excited in this configuration.

Fig. 11 shows the current distributions of two polarizations, which are totally different from the slot antenna. For the same application of 2.4 GHz WLAN in last design, the rectangular slot is etched in a large ground. The slot's length and width are approximately half wavelength. For a typical slot mode, the width of extended ground is a quarter of wavelength or smaller. If the size of surrounded ground decreases to some level, the slot turns to be a loop mode with the frequency shift. What's more, a loop has four edges with the overall dimension of the loop antenna is 40x53 mm2, including the feeding structure. The slot antenna is with the dimension of 100x80 mm2. It is clear that the area of the proposed antenna is only 26.5% of the slot one. Fig. 12 (a) and (b) show the loop antenna, in front and

microstrip line adopted in the last design is an effective solution to feed the dual-mode of loop antenna. Fig. 10 shows the geometry of the loop antenna, which is quite similar as the slot design. This antenna consists of a rectangular loop, a CPW feeding and a microstrip line, and supported by the same FR4 board as last design with the thickness of 1 mm. The loop has width of 4 mm; narrower than the slot design. The loop and CPW are etched on the

Fig. 10. Geometry of the proposed loop antenna. (L1=53 mm, L2=33 mm, L3=20 mm, L4=16 mm; L5=5.1 mm, L6=18.5 mm, L7=6.5 mm, W1=40 mm, W2=32 mm, W3=2 mm, W4=1.9 mm,

When the loop fed through port 1, the CPW operates at its typical symmetrical mode. In this mode the vertical polarization is excited. The inner conductor works as a monopole with the vertical polarization. The energy is coupled from monopole to the loop, exciting the vertical polarization mode. It is a good solution to feed the one-wavelength mode at the position of current null. The radiation consists of two modes, the one-wavelength mode of the loop and a monopole mode. When the loop is fed through port 2, the horizontal polarization of the loop antenna is excited. The feed is exactly at the maximum of current, and the horizontal

Fig. 11 shows the current distributions of two polarizations, which are totally different from the slot antenna. For the same application of 2.4 GHz WLAN in last design, the rectangular slot is etched in a large ground. The slot's length and width are approximately half wavelength. For a typical slot mode, the width of extended ground is a quarter of wavelength or smaller. If the size of surrounded ground decreases to some level, the slot turns to be a loop mode with the frequency shift. What's more, a loop has four edges with the overall dimension of the loop antenna is 40x53 mm2, including the feeding structure. The slot antenna is with the dimension of 100x80 mm2. It is clear that the area of the proposed antenna is only 26.5% of the slot one. Fig. 12 (a) and (b) show the loop antenna, in front and

S=0.5 mm. Reprinted from (Li et al, 2011a) by the permission of IEEE).

mode is clearly excited in this configuration.

front side and the microstrip line is printed on the back side.

back views, respectively. Fig 12(c) shows the slot antenna design, which also operates in the same band. A significant size reduction is achieved using the loop design.

Fig. 11. Current distributions of (a) vertical polarization and (b) horizontal polarization.

Fig. 12. Photograph of the loop antenna (a) front side, (b) back side and (c) the slot antenna.the total length is one wavelength. Therefore, the dimension of a rectangular loop antenna is much smaller than the slot design with large ground. However, the slot antenna can be adopted in the array design in the same ground for special requirements.

The measured and simulated S parameters are illustrated in Fig. 13. The -10 dB bandwidth of the reflection coefficients are 770 MHz (1.98-2.75 GHz, 32.1%) for the vertical polarization and 730 MHz (1.96-2.69 GHz, 30.4%) for the horizontal polarization, both covering the of 2.4 GHz WLAN band. The isolation in this band is better than -21.3 dB, which is lower than the slot design, as a cost of dimension reduction. The isolation deterioration is mainly contributed to the feeding structure of the vertical polarization. The feeding monopole is located at the current maximum point of the horizontal mode. The energy couples between two modes. But it still fulfils the -15 dB industrial requirement.The radiation pattern s of the loop antenna is quite similar to the slot antenna, but with a lower level of cross polarization. In the 2.4 GHz WLAN band, the measured gains are better than 2.9 dBi and 4.1 dBi. Considering the compact structure of loop, this antenna is suitable for the space-limited systems.

Fig. 13. Simulated and measured S parameters of the loop antenna.

#### **3. Polarization reconfigurable antenna**

As described in the introduction, reconfigurable antenna is an effective solution for the space-limited MIMO systems by adaptive antenna selection. This kind of systems is called adaptive MIMO system. The adaptive MIMO system takes the advantage of varying channel characteristics to make the best use of the improvement of channel capacity (Cetiner et al, 2004). Due to the channel condition, different antenna properties, such as polarizations and radiation patterns, are selected for better transmitting or receiving. Also, different data processing algorithms are used depending on the antenna selection. For this reason, the reconfigurable antenna is very important to the MIMO system, especially for the spacelimited system. In this section, we will introduce the polarization reconfigurable antenna, based on the dual-polarized slot antenna described in the last section. In order to validate the benefit of polarization selecting, the channel capacity of a 2x2 MIMO system using the polarization reconfigurable antenna has been measured in a typical indoor scenario.

#### **3.1 Reconfigurable mechanism**

The geometry of the proposed reconfigurable slot antenna element is shown in Fig. 14, based on the design of (Li et al, 2010). The port 1 and port 2 are combined together and controlled by two PIN diodes. The port 1 is connected the microstrip line on the back side through a via hole, and controlled by PIN 1. The port 2 is connected directly to the microstrip line on the back side, and controlled by PIN 2. When PIN1 is ON and PIN2 is OFF, the antenna is fed through the port 1. The vertical polarization of the slot is excited. When PIN1 is OFF and PIN 2 is ON, the antenna is fed through the port 2, and the horizontal polarization of the slot is excited. Therefore, two ports are fed alternatively and controlled by the PIN diodes. The two PIN diodes need the bias circuit to control. Due to compact feed design, the two PIN diodes share the same bias circuit, saving the space of the antenna system and using less lumped components.

Considering the compact structure of loop, this antenna is suitable for the space-limited

As described in the introduction, reconfigurable antenna is an effective solution for the space-limited MIMO systems by adaptive antenna selection. This kind of systems is called adaptive MIMO system. The adaptive MIMO system takes the advantage of varying channel characteristics to make the best use of the improvement of channel capacity (Cetiner et al, 2004). Due to the channel condition, different antenna properties, such as polarizations and radiation patterns, are selected for better transmitting or receiving. Also, different data processing algorithms are used depending on the antenna selection. For this reason, the reconfigurable antenna is very important to the MIMO system, especially for the spacelimited system. In this section, we will introduce the polarization reconfigurable antenna, based on the dual-polarized slot antenna described in the last section. In order to validate the benefit of polarization selecting, the channel capacity of a 2x2 MIMO system using the

polarization reconfigurable antenna has been measured in a typical indoor scenario.

The geometry of the proposed reconfigurable slot antenna element is shown in Fig. 14, based on the design of (Li et al, 2010). The port 1 and port 2 are combined together and controlled by two PIN diodes. The port 1 is connected the microstrip line on the back side through a via hole, and controlled by PIN 1. The port 2 is connected directly to the microstrip line on the back side, and controlled by PIN 2. When PIN1 is ON and PIN2 is OFF, the antenna is fed through the port 1. The vertical polarization of the slot is excited. When PIN1 is OFF and PIN 2 is ON, the antenna is fed through the port 2, and the horizontal polarization of the slot is excited. Therefore, two ports are fed alternatively and controlled by the PIN diodes. The two PIN diodes need the bias circuit to control. Due to compact feed design, the two PIN diodes share the same bias circuit, saving the space of the

Fig. 13. Simulated and measured S parameters of the loop antenna.

**3. Polarization reconfigurable antenna** 

**3.1 Reconfigurable mechanism** 

antenna system and using less lumped components.

systems.

Fig. 14. Geometry of the proposed loop antenna. (L=120 mm, LS=50 mm, LG=36 mm, L0=16 mm; L1=8.9 mm, L2=33.9 mm, L3=15.3 mm, L4=20.1 mm; L5=30 mm, W1=1.9 mm, WS=53 mm, W=80 mm, S1=0.7 mm, S2=1 mm. Reprinted from (Li et al, 2011b) by the permission of John Wiley & Sons, Inc.).

A prototype of the dual-polarized slot antenna with switching mechanism is fabricated, and shown in Fig. 15. The PIN diodes with bias circuit are on the back side of the antenna. The detailed bias circuits of two PIN diodes (D1 and D2, Philips BAP64-03) are shown in Fig. 15 (c). The 'ON' and 'OFF' states of the two PIN diodes are controlled by a 1-bit single-pole 2 throw (SP2T) switch on the front side. The bias circuit consists of three RF choke inductors (Lb1 Lb2 and Lb3, 12 nH), a DC block capacitor (Cb, 120 pF), three RF shorted capacitors (Cs1

Fig. 15. Photograph of the antenna prototype (a) front side, (b) back side; (c) bias circuit of the PIN diodes.

Cs2 and Cs3, 470 pF) and a bias resistor (R, 46 Ω). The bias resistor is selected depend on the value of VCC and the operating current of the PIN diode. In this application, the VCC is 3 V.

The measured reflection coefficients for both polarizations are shown in Fig. 16. Compared with results of the dual-polarized slot antenna in Fig.13, the difference is mainly contributed from the parasitic parameters of PIN diodes and the bias circuit. The -10dB bandwidths are 700MHz (2.02-2.72 GHz, 29.2%) and 940MHz (1.84-2.78 GHz, 40%) for vertical and horizontal polarizations, both covering the WLAN band (2.4-2.484 GHz). The gain decreases approximately 0.5 dB due to the insertion loss of PIN diodes.

Fig. 16. Simulated and measured S parameters of the reconfigurable antenna.

#### **3.2 Channel capacity measurement**

In this section, we measured the channel capacity of a 2x2 MIMO system in a typical indoor scenario by using the proposed polarization reconfigurable antenna. The measurement setup is shown in Fig.17. The measurement system consists of an Agilent E5071B Vector network analyzer (VNA), which has 4 ports for simultaneous measurement, transmit and receive antennas, a computer and RF cables. Two standard omni-direcitional dipoles are utilized as the transmit antennas (TX), and arranged perpendicular to XY plane along Z axis. Two proposed reconfigurable antennas are used as the receive antennas (RX). The 2x2 antennas are connected to the 4 ports of the VNA. The computer is used to control the measurement procedures and record the measured channel responses. In order to validate the improvement in channel capacity by using reconfigurable antennas, another two reference dipoles are adopted as receive antennas for comparison. The measurement was carried out in a room of the Weiqing building, Tsinghua University, illustrated in Fig. 18. The framework of the room is reinforced concrete, the walls are mainly built by brick, and the ceiling is made with plaster plates with aluminium alloy framework. The heights of desk partition and wood cabinet are 1.4 m and 2.1 m. The transmit antennas are fixed in the middle of room (TX). The receive antennas are arranged in several typical locales which are noted as RX1-5 in Fig. 20. Here, the scenarios when the receive antennas are arranged in RX1 and RX2 are line-of sight (LOS), while that is NLOS when the receive

Cs2 and Cs3, 470 pF) and a bias resistor (R, 46 Ω). The bias resistor is selected depend on the value of VCC and the operating current of the PIN diode. In this application, the VCC is 3 V. The measured reflection coefficients for both polarizations are shown in Fig. 16. Compared with results of the dual-polarized slot antenna in Fig.13, the difference is mainly contributed from the parasitic parameters of PIN diodes and the bias circuit. The -10dB bandwidths are 700MHz (2.02-2.72 GHz, 29.2%) and 940MHz (1.84-2.78 GHz, 40%) for vertical and horizontal polarizations, both covering the WLAN band (2.4-2.484 GHz). The gain decreases

approximately 0.5 dB due to the insertion loss of PIN diodes.

Fig. 16. Simulated and measured S parameters of the reconfigurable antenna.

In this section, we measured the channel capacity of a 2x2 MIMO system in a typical indoor scenario by using the proposed polarization reconfigurable antenna. The measurement setup is shown in Fig.17. The measurement system consists of an Agilent E5071B Vector network analyzer (VNA), which has 4 ports for simultaneous measurement, transmit and receive antennas, a computer and RF cables. Two standard omni-direcitional dipoles are utilized as the transmit antennas (TX), and arranged perpendicular to XY plane along Z axis. Two proposed reconfigurable antennas are used as the receive antennas (RX). The 2x2 antennas are connected to the 4 ports of the VNA. The computer is used to control the measurement procedures and record the measured channel responses. In order to validate the improvement in channel capacity by using reconfigurable antennas, another two reference dipoles are adopted as receive antennas for comparison. The measurement was carried out in a room of the Weiqing building, Tsinghua University, illustrated in Fig. 18. The framework of the room is reinforced concrete, the walls are mainly built by brick, and the ceiling is made with plaster plates with aluminium alloy framework. The heights of desk partition and wood cabinet are 1.4 m and 2.1 m. The transmit antennas are fixed in the middle of room (TX). The receive antennas are arranged in several typical locales which are noted as RX1-5 in Fig. 20. Here, the scenarios when the receive antennas are arranged in RX1 and RX2 are line-of sight (LOS), while that is NLOS when the receive

**3.2 Channel capacity measurement** 

antennas are arranged in RX3, RX4 and RX5. In this measured, the antennas used are fixed at the height of 0.8 m. The space of antenna elements in TX or RX is 0.5λ, with the mutual coupling less than -25dB.

Fig. 17. Experiment setup of the measurement.

Fig. 18. Layout of measurement environment.

The measurement was carried out in the band of 2.2-2.6 GHz, with a step of 2 MHz. Three different orientations (ZZ, YY, and XX) of RX antennas were measured to simulate different operational poses of the mobile terminals. For two horizontal (H) and vertical (V) polarizations reconfigurable antennas, 4 configurations (HH, HV, VH, VV) were switched manually for each channel capacity measurement in a quasi-static environment, and the result with the biggest value was chosen for statistics. Given the small-scale fading effect, 4x4 grid locations for each RX position were measured. Therefore, a total 201x3x16x2=19296 measured channel capacity for LOS condition was obtained, and 201x3x16x3=28944 was the measured results for NLOS condition.

The channel capacity can be calculated through following formula (Foschini & Gans, 1998):

$$C = \log\_2 \det[I\_{N\_r} + \frac{SNR}{N\_t} H\_n H\_n^H] \tag{1}$$

where Nr and Nt are the numbers of RX and TX antennas. *INr* is a Nr x Nr identity matrix, SNR is the signal-to-noise ratio at RX position, *Hn* is the normalized *H*, and *()H* is the Hermitian transpose. *H* is normalized by the received power in the 1x1 reference dipole with identical polarization. We selected the SNR when the average channel capacity is 5 bit/s/Hz in a 1x1 reference dipole system in LOS or NLOS scenario.

The measured Complementary Cumulative Distribution Functions (CCDF) of the channel capacity for the 2x2 MIMO system using polarization reconfigurable antennas in both LOS and NLOS conditions are shown in Fig. 19 and 20. As summarized in Table 1, the average and 95% outage channel capacities are both improved, especially in NLOS scenario. For NLOS, the received signal is mainly contributed from reflection and diffraction, which vary the polarization property of the wave. However, the path loss is higher in NLOS scenario. The transmit power should be enhanced to guarantee the system performance. Considering the insertion loss introduced from non-ideal PIN diodes, better capacity can be obtained by using high quality components. The measurement results prove the benefit by using polarization reconfigurable antennas.

Fig. 19. CCDFs of channel capacity in LOS condition.

manually for each channel capacity measurement in a quasi-static environment, and the result with the biggest value was chosen for statistics. Given the small-scale fading effect, 4x4 grid locations for each RX position were measured. Therefore, a total 201x3x16x2=19296 measured channel capacity for LOS condition was obtained, and 201x3x16x3=28944 was the

The channel capacity can be calculated through following formula (Foschini & Gans, 1998):

*SNR C I HH*

bit/s/Hz in a 1x1 reference dipole system in LOS or NLOS scenario.

<sup>2</sup> log det[ ] *<sup>r</sup>*

where Nr and Nt are the numbers of RX and TX antennas. *INr* is a Nr x Nr identity matrix, SNR is the signal-to-noise ratio at RX position, *Hn* is the normalized *H*, and *()H* is the Hermitian transpose. *H* is normalized by the received power in the 1x1 reference dipole with identical polarization. We selected the SNR when the average channel capacity is 5

The measured Complementary Cumulative Distribution Functions (CCDF) of the channel capacity for the 2x2 MIMO system using polarization reconfigurable antennas in both LOS and NLOS conditions are shown in Fig. 19 and 20. As summarized in Table 1, the average and 95% outage channel capacities are both improved, especially in NLOS scenario. For NLOS, the received signal is mainly contributed from reflection and diffraction, which vary the polarization property of the wave. However, the path loss is higher in NLOS scenario. The transmit power should be enhanced to guarantee the system performance. Considering the insertion loss introduced from non-ideal PIN diodes, better capacity can be obtained by using high quality components. The measurement results prove the benefit by using

*N n n t*

*H*

*<sup>N</sup>* (1)

measured results for NLOS condition.

polarization reconfigurable antennas.

Fig. 19. CCDFs of channel capacity in LOS condition.

Fig. 20. CCDFs of channel capacity in NLOS condition.


Table 1. Average and 95% Outage Channel Capacity (bit/s/Hz).
