**2.3 Module optimization**

After assessment of large-scale prototype measurements, presented in **Figure 5**, the true complexity of Q3 and Q4 transistors driving signals were revealed–although the circuit operation in **Figure 1** may appear simple, it needs a fairly complex excitation. In order to balance out positive and negative half-period amplitudes in terms of


**Table 1.** *PWM duty cycles (%) for Q3/Q4 transistors @ VCC = 5 V.* *Influence of Piezoelectric Actuator Properties on Design of Micropump Driving Modules DOI: http://dx.doi.org/10.5772/intechopen.103789*


#### **Table 2.**

*PWM duty cycles (%) for Q3/Q4 transistors @ VCC = 12 V.*

DC offset, a single, switched PWM generator, with a switching frequency in order of 30 kHz with at least 7-bit PWM resolution had to be implemented as a cost-effective solution. Micropump output voltage should be monitored and included in closed-loop regulation with adjustment of duty cycle to minimize the driving signal DC offset.

The above-listed requirements would preferably have to be implemented in software, using an 8-bit microcontroller. In our cost-effective implementation, depicted in **Figure 6**, a Microchip ATtiny10 [8], was selected for its price and availability in a 6-pin SOT-23 package. In order to extend the module power supply operating range to 18 V, the microcontroller is connected to a non-inverting, 2-channel MOSFET driver TC4427, which also acts as a voltage level translator for driving signals NPN\_DRIVE and PNP\_DRIVE (refer to **Figure 1**). Driver inputs are 5 V compatible, therefore the microcontroller can operate at a 3.3 V power supply, thus keeping total current consumption at a minimum, which is a prerequisite for

**Figure 6.** *Miniaturized driver based on microcontroller ATtiny10.*

autonomous (e.g. battery-powered) application. Low voltage 3.3 V microcontroller power supply was generated using LM1117–3.3 circuit.

In order to provide driving signals for Q3/Q4 transistors, an 8-bit PWM unit zero in ATtiny10 with two output compare channels (A and B) was used in phase non-aligned mode. Compare value of PWM channel A and B was set using dedicated output compare registers OC0A/OC0B, respectively. Interrupt, which can be triggered upon output compare match with corresponding compare register (OC0A/OC0B), was not used. Instead, each PWM channel toggles between a switching (PWM) state and an inactive (fully open) state. Switching between both states, previously achieved with a '257 multiplexer, can be implemented without additional components, by switching the corresponding pin mode from PWM state (i.e. OC0x mode) to normal I/O state (i.e. PORTB I/O mode).

Achieving switching frequencies in the range of 30 kHz with 8-bit resolution requires the internal oscillator to be configured at 8 MHz. Using such an internal clock signal with no prescaling on timer TMR0 synthesizes a 31.25 kHz PWM clock. Resulting PWM unit features an 8-bit resolution.

Micropump excitation frequency was determined by counting TMR0 timer overflows. As the number of overflows reaches a predefined value, the Q3/Q4 transistors excitation roles are reversed by toggling the PWM pin mode and resetting the overflow counter. All this was accommodated inside interrupt routines, without the need for a complex program. In order to compensate for supply voltage variations, the microcontroller also features an analogue-to-digital (A/D) converter, which is used for module power supply measurement. Using A/D readout, PWM driving frequency and both duty cycles were adjusted according to the values, predefined in a software table, derived from **Tables 1** and **2**. This table also encompasses overand under-power supply voltage detection, which causes the module to turn both Q3 and Q4 outputs off. Presented approach implements an open-loop control, which adjusts the duty cycle value according to the power supply voltage.

## **2.4 Electrical and fluidic characterization**

Previously presented optimized module was tested for current consumption and output signal accuracy with different capacitive loads connected to the output. A multimeter on the power supply line was used to measure current consumption and an oscilloscope was used to measure output signal parameters (waveform shape, amplitude symmetry, frequency). High voltage variable capacitor with values from 10 nF to 100 nF and several piezoelectric micropumps in parallel were used to vary the capacitance of output load. A constant micropump operation frequency was set at 190 Hz, while the power supply voltage was set to 9 V.

Though the resulting capacitance driving capability in **Figure 7** exhibits practically linear dependency, it must be emphasized, that the signal shape deteriorates severely and the positive signal slew rate drops from 140 V/ms range to 80 V/ms.

Measured limiting value of load capacitance, which maintains 140 V/ms slew rate was found at 57 nF with a driving voltage of 172 Vpp. Larger capacitive loads (> 57 nF) cause the micropump driver current to saturate, which deteriorates the output signal rise- and fall-time and correspondingly the slew-rate. This reduction consequentially results in a drastic deterioration of micropump flow and backpressure characteristics. Such output signal deterioration was attributed primarily to limited current passing through the inductor L1.

*Influence of Piezoelectric Actuator Properties on Design of Micropump Driving Modules DOI: http://dx.doi.org/10.5772/intechopen.103789*

**Figure 7.** *Micropump driving voltage vs. variable load capacitance.*

In order to achieve better signal integrity, peak L1 inductor current should be increased–which is in contradiction with low-power design aspect. Due to unavailability of higher-quality inductors, all our designs used a Bourns SRR0603-102KL inductor [].

**Figure 8** represents measured results of output signal with 12 nF capacitive loading, achieving 236 VPP drive amplitude, with positive and negative slew rates 142 V/ms and 163 V/ms, respectively. Positive cycle amplitude was 122 V, while the negative was −114 V.

**Figure 8.** *Micropump driving voltage at 87 Hz.*

Afterwards, the module current consumption was measured after performing a sweep of power supply voltages.

Driver current consumption vs. power supply voltage is depicted in **Figure 9**. Module current consumption increases linearly with respect to increasing load and signal frequency up to 55 mA at 12 V power supply, which was the upper design interval

**Figure 9.** *Driver current consumption.*

**Figure 10.** *Measurement setup for micropump fluidic characterization.*

## *Influence of Piezoelectric Actuator Properties on Design of Micropump Driving Modules DOI: http://dx.doi.org/10.5772/intechopen.103789*

limit and also the power dissipation limit of the driver TC4427. Microcontroller current consumption remains negligible throughout entire measurement range. Measured driver current consumption was found between 22 mA (5 VCC) and 55 mA (12 VCC).

A dedicated computer-controlled system for characterization of piezoelectric micropumps was set up after electrical evaluation of the optimized micropump driver. The measurement system, presented in **Figure 10**, enables frequency scans and allows simultaneous pressure and flow measurements.

Analyzed micropump driver was connected to a micropump, developed in our laboratory [7]. A reservoir, filled with DI water, was connected to the micropump input and pressure/flow evaluation equipment on its output.

Micropump flow was measured indirectly by weighing the mass of pumped media over a known period of time using a Kern ABJ 120−4M precision scale. Obtained results were corrected to account for the evaporation of medium during the measurement, which was determined in separate experiments.

Flow measurements are presented in **Figure 11**. Results show frequency scan with comparison to Bartels mp-x Controller [1], which was preset with the same amplitude and frequency conditions. The only difference in driving parameters was the signal shape, where the Bartels mp-x Controller was set to square-wave signal, and the tested driver provided RC like shape, presented in **Figure 8**. The same micropump was driven on a frequency interval from 50 Hz to 400 Hz and both duty cycles were set according to values in **Tables 1** and **2** to achieve minimal DC offset of drive voltage.

**Figure 12** summarizes obtained both positive (SR<sup>+</sup> ) and negative (SR− ) slew-rates as well as micropump driving voltage amplitude (not peak-to-peak) with respect to frequency.

For lower frequency range (i.e. under 80 Hz), designed driver actually supersedes the flow performance of Bartels mp-x controller, achieving 1.2 ml/min in comparison to 0.8 ml/min obtained with mp-x module. However, as the driving frequency increases, the slew rate and drive amplitude decrease, resulting in a steady exponential decay of flow performance. As can be observed from **Figures 11** and **12**, best flow

**Figure 11.** *Micropump DI water flow rate vs. frequency.*

**Figure 12.** *Slew rate/drive amplitude vs. frequency.*

performance of the designed driver is achieved at low frequencies (< 80 Hz), where it is primarily limited by driving voltage amplitude VPUMP and both slew-rates SR+ and SR− , depicted in **Figure 12**.
